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Full text of "Basic Television ( parts 1-3 complete )"

BASIC TELEVISION 

Part I 




c«mmqn -co*l 



A Basic Training Manual developed by 

H. A. COLE, C.Eng., M.I.E.R.E., 

working in conjunction with 

the Editorial and Art Staff of the Publishers. 




LONDON 

THE TECHNICAL PRESS LTD 



NEW YORK 
THE BROLET PRESS 



First published 1967 

Copyright © 1967 by 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 

New York, U.S.A. 

All rights reserved 

First published 1967 

Reprinted 1970 

Reprinted with metric amendments 1972 



The words "COMMON-CORE", with device and without device, 
are trade-marks of the Copyright owners 



SBN 291.39572.4 



Made and printed by offset in Great Britain by 
William Clowes & Sons, Limited, London, Beccles and Colchester 



PREFACE 

Ahe aim of this Series on BASIC TELEVISION is to explain in simple language the 
physical principles which make television possible and the way in which a typical 
television system works — from the generation of the signal in the TV camera to the 
final presentation of the picture on the screen by your own fireside. The Series is based 
on the two TV systems working in Great Britain today — the very-high frequency (VHF) 
one working on 405 lines per picture and the ultra-high frequency (UHF) one working 
on 625 lines per picture. The receiver considered in Parts 2 and 3 is the British Dual- 
Standard Receiver which is capable, on operation of the " Standard Selection" control, 
of receiving programmes on either of these two considerably different systems. 

Two decisions of particular importance had to be made in planning the Series. 
The first was to describe the working of the TV receiver almost wholly in terms of 
valves, even though in many of the latest single-standard and colour receivers the 
thermionic valve is being progressively replaced by semiconductor devices. This 
decision was made on two grounds. The first was that a large majority of the 
millions of receivers operational in Britain in the second half of 1971 are wholly or 
mainly valve-operated rather than transistorized and that, for technical and economic 
reasons which are more fully discussed in the final Section of Part 3 "trends in tv 
receiver design", the valve will in all probability continue to play an important part 
in TV receivers, especially in those built on the Dual-Standard principle, for a signi- 
ficant number of years to come. The second reason was that, since the COMMON- 
CORE Series as it exists at present is planned on the basis of explaining the working 
of electronic devices in terms of current flow through a valve, it was desirable to 
keep this account of the basic principles on which television works compatible with 
the foundation COMMON-CORE volumes in their present form. 

The other major decision in planning BASIC TELEVISION was to cover black-and- 
white ("monochrome") transmission and reception Only, in the interest of keeping 
the descriptions of the various stages in the studio camera, the transmitter and the 
receiver relatively simple and relatively short. With the basic principles involved thus 
established (it is hoped) in the reader's mind, a further Series on Basic Colour TV, 
fully transistorized to reflect modern progress, is currently planned. 

Most of the measurements given in the Series have been expressed (or in Part 1, 
which was first published in 1967, re-expressed) in SI Metric units. In particular, 
"Hertz" and "MHz" have been used in place of "cycles per second" and "Mc/s" 
throughout. But certain measurements either familiar to the viewer (e.g., the sizes 
of picture tube) or else representative of orders of magnitude rather than of precise 
distances have been left in inches, miles, etc., as being more likely in that form to give 
the ordinary reader a clear picture of the point being made. 

The Series has been written and illustrated to take its place in the growing 
COMMON-CORE Series of Illustrated Training Manuals on subjects connected with 
electricity and electronics. Originated in the United States by the distinguished 
New York firm of technical education consultants and graphiological engineers, 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 



the twenty-one Manuals of which the COMMON-CORE Series now consists have 
already sold over 1,500,000 copies in their British and Commonwealth editions. Six 
of the Manuals have been wholly conceived, written and illustrated in the United 
Kingdom; while all the remainder have been extensively rewritten to conform with 
British terminology and notation. 

The BASIC TELEVISION Manuals presuppose in the reader a working know- 
ledge of the contents of the foundation volumes of the COMMON-CORE Series, 
principally the five Parts of BASIC ELECTRICITY and the six Parts of BASIC 
ELECTRONICS. Prior acquaintance with the two-part series BASIC ELEC- 
TRONIC CIRCUITS will also prove useful when the operation of the TV receiver 
is studied in Parts 2 and 3. 

The BASIC TELEVISION Series has been written, in conjunction with the editorial 
staff of the Publishers, by Mr. H. A. Cole, a Senior Scientific Officer in the Elec- 
tronics and Applied Physics Division of the Atomic Energy Research Establishment 
at Harwell. Mr. Cole is a Chartered Engineer, and a Member of the Institution 
of Electronic and Radio Engineers. All illustrations of a technical nature have been 
drawn by Mr. Cole himself, with the Art Department of the technical press 
responsible for their "decoration" and captioning. 



TABLE OF CONTENTS 



Section 

1 The Role of Television 

2 The Basic Principles 

3 Scanning 

4 The Picture Signal 

5 The Video Signal 

6 The Television Studio 

7 The Transmitter and Aerial 

8 Signal Bandwidth 



Page 
1.1 

1.5 

1.21 

1.37 

1.68 

1.81 

1.99 

1.131 



The COMMON 




CORE Series 



of Basic Training Manuals 
embraces so far the following titles: 

BASIC ELECTRICITY 

BASIC ELECTRONICS 

BASIC SYNCHROS AND SERYOMECHANISMS 

BASIC ELECTRONIC CIRCUITS 

BASIC RADAR 

BASIC INDUSTRIAL ELECTRICITY 

BASIC TELEVISION 



Foreword on International 
TV Systems 



The television set round which this Series has been written is the so-called British 
Dual-Standard Set, which is capable of receiving signals on two distinct line-systems — 
the 405-line and the British 625-line systems. 

If you wonder at the emphasis placed on the word "British" in that phrase, "the 
British 625-line system", the reason for it is that it has regrettably not yet been possible 
to secure international agreement on all the technical details of any standard 625-line 
system. 

For some time past, it has been the aim of the C C I R (the Comiti Consultatif 
International des Radio, or International Radio Consultative Committee) to persuade 
all the countries of the world to adopt a common TV system, on the grounds that it 
would be of great benefit to everyone from the point of view of convenience, ease of 
programme exchange, and manufacturing economy. Although complete agreement 
is still a long way off, progress has certainly been made over the past few years. 

There are at present seven major TV systems in the world : the American 525-line, 
the French 625-line, the French 819-line, the West European 625-line, the East 
European 625-line, the British 405-line, and the British 625-line systems. The 
British 405-line system is due to be gradually discontinued over the next few years 
and will eventually be replaced by a 625-line system. 

Unfortunately, not all European countries — even the Western ones — agree on the 
technical details of a standard 625-line system. It is true that they agree on such 
important features as aspect ratio, scanning sequence, method of interlacing and a 
few others; but differences still exist over (for example) the choice of vision bandwidth, 
channel spacing, sound-to-vision carrier spacing, and the degree of modulation which 
shall correspond to black level. These differences, though not very great, can some- 
times prevent satisfactory exchange of two 625-line programmes. For example, the 
625-line system employed by Belgium and France uses amplitude modulation for the 
sound carrier, whereas all other European countries use frequency modulation. 
Similar differences exist elsewhere in Europe over the relative spacing of the sound 
and vision carriers. 

The Western European and Eastern European systems differ mainly in the values 
chosen for channel width and vision bandwidth. The Western European system uses 
a 5 MHz vision bandwidth and 7 MHz channel spacing, whereas the Eastern European 
system uses a 6 MHz vision bandwidth and 8 MHz channel spacing. 

The British 625-line system differs from both European systems in that it uses 
a 5-5 MHz vision bandwidth and 8 MHz channel spacing. Other differences concern 
the width of the vestigial sideband and the setting of the black level. 



§ I : THE ROLE OF TELEVISION 



1.1 



The world's first transmission of a regular televison programme was inaugurated 
by the British Broadcasting Corporation on the evening of November 2nd, 1936. 
On that opening night of the new service, the number of receivers in private hands 
which were capable of taking the programme radiated was very small. But the 
demand for sets soon started rising fast, and by the time the service had to be closed 
down in September 1939, Britain easily led the world with at least 20,000 sets in 
regular use. 

Today, despite the seven-year interruption caused by the War, the number of sets 
in private households is some 12 million; and nearly 99% of the population could 
receive television transmissions in their own homes. 

Elsewhere, the growth of TV has been comparable. In Europe, where the various 
national television services could not get under way again until well after the end of 
hostilities, the transmission of international programmes by Eurovision is now a 
regular event. In the United States (where a UNESCO report has estimated that 
there were no more than 5,000 receivers in use as recently as May 1941), Britain's 
figure of 12 million sets has long ago been left far behind. Other countries now 
transmitting television programmes designed, in varying degrees, for public informa- 
tion and entertainment include Australia, Russia, Japan, and many of the Republics 
of South America and of the more recently independent countries of Asia and Africa. 

In Britain, since the transmission of television programmes was resumed by the 
BBC in 1946, the social impact has been enormous. Several independent companies 
relying for their revenue on televised advertisements began broadcasting their pro- 
grammes from 1955 onwards; and a third national Television Channel (using, as 
you will see, somewhat different techniques) began to broadcast BBC 2 programmes 
in 1964. 

Today, few British homes lack a television aerial sprouting from their roof-top; 
and "The Telly" has become a household word. 




1.2 [§ I 

TV as Entertainment — and as a Cultural Force 

The sheer entertainment value which this new electronic marvel can provide would 
have seemed, only a quarter of a century ago, almost fabulous. At the turn of a 
switch, artists and performers of every kind — men and women commanding the 
highest salaries in the world of entertainment— can be brought almost physically 
into the comfort of your own sitting-room, there to give what amounts to a private 
Command Performance in every home in the land! 




Events taking place many hundreds of miles away can be similarly enjoyed, often 
at the very moment they are taking place— Lord's, Wembley, Wimbledon and St. 
Andrew's, Ascot, Aintree or Oulton Park, Trooping the Colour or the consecration 
of a new Cathedral. Add the vast range of music and drama presented, of spectacle 
and music-hall, of serious talks and not-so-serious quiz-games, of news broadcasts, 
symphony concerts and dancing competitions, of politicians, pop-singers, scientists, 
clergymen and visiting celebrities. . . . Not bad value there for a TV licence fee of 
a few pounds a year, if you care to think of things that way! 

Together with the power to entertain, however, goes also the power to influence. 
It has been claimed that the result of a modern General Election can be swayed by 
the performance of a dozen politicians on TV; while no one who witnessed them on 
the television screen can ever forget the impact made by the living scenes of the 
Queen's Coronation in Westminster Abbey in 1953, or of the funeral of Sir Winston 
Churchill passing through the streets and up the River of London on that grey 
January morning in 1965. 

The magic power of Television carries with it, therefore, a heavy responsibility. 
For even while it entertains, it cannot help but broaden the cultural interests of 
millions of its viewers. People who would never have considered paying to see a 
serious play, or listening to a Wagner opera, or visiting a Gallery to look at a master- 
piece of modern art, have been enabled to discover in themselves interests which they 
never knew they possessed, and to derive from these interests a satisfaction of the 
mind quite different (in many cases) from anything they had ever experienced before. 

It is estimated, by way of a single example, that more people watched the Laurence 
Olivier film Richard III, when it was televised recently in the United States, than had 
watched all the stage productions of all Shakespeare's plays put together, in all the 
350-odd years since the plays were written. . . . 



§ I] 13 

TV in Education, in Medicine and in Industry 

In Education, in Medicine and in Industry, television already has important 
applications; but far bigger possibilities lie ahead. 

Lessons on specialist subjects are already, on a small scale, broadcast to schools. 
But Britain, it is said, faces a shortage of 70,000 teachers by the end of the 1960's. 
There could be an enormous saving of brain-power (and possibly an improvement in 
overall teaching performance as well) if whole courses could be taught to thousands 
of students by a single expert appearing at stated hours on TV. 

In Medicine, many of our teaching hospitals today use TV cameras to take a 
continuous close-up picture of an experienced surgeon's hands at work on an opera- 
tion. The picture is then relayed to a class of medical students in another room, where 
the twin problems of crowding and infection cannot arise. 

In Industry, television is already much used, and will be more so. With its sound 
and picture impulses distributed in a "closed circuit" by land-line rather than through 
the air, widely separated executives in a large business can hold face-to-face conferences 
without loss of secrecy, or of time spent in travel. 

Television can watch at close range things happening at critical points in an 
industrial process which are either inaccessible to a human being, or else too dangerous 
for him to approach (as when a TV set is used to monitor the remote handling of 
highly radioactive materials). 



9. ©I 

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Other industrial uses of television, too numerous even to guess at today, un- 
doubtedly lie ahead. 



1.4 



TV in Research into the Unknown 

Most of the land surface of the Earth 
has now been fairly closely investigated; 
but exploration of the depths of the Ocean, 
or of the immensities of Space which sur- 
round our tiny planet, has scarcely begun, 
[n both adventures, television will clearly 
play a major part. 




The depth to which a human diver can descend is normally limited to a 
few hundred feet. Yet the greatest depths of the Ocean are much farther 
below sea-level than the summit of Mount Everest is above it! Suitably 
armoured, and equipped with proper lighting to pierce the utter blackness 
of the deep, TV cameras let down on cables from a ship can send back 
pictures of immense value — whether the quest be for a sunken ship or 
disabled submarine, or for a biological "snapshot" of one of the gruesome 
creatures that blindly hunt their sightless prey thousands of feet below the 
surface of the waves. 

In Space, the Russians have already sent back televised pictures of photo- 
graphs of the far side of the Moon, and the Americans (over a range of some 
50 million miles) of the surface of the planet Mars. Circling the Earth as this 
is written are a number of American satellites whose function is to study the 
cloud formations that form and re-form over the Earth, and to send back 
pictures which may be of great value in the accurate forecasting of world- 
wide weather conditions. 



EARLY BIRD, launched privately in the United States, is already trans- 
mitting TV pictures across the Atlantic; and planned for the near future are 
joint European-American launchings of satellites to be placed in orbit at 
intervals round the Earth. Their function is to receive, amplify and re- 
radiate radio-frequency waves modulated to carry TV pictures, and so to 
defeat the limitations on world-wide TV coverage imposed by the curvature 
of the Earth. Constantly recharged by thousands of solar cells set into the 
outer skin of their carriers, the batteries in these space-craft are planned to 



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§2: THE BASIC PRINCIPLES 



1.5 



The major problems which a practical TV system has to solve can be stated as 
follows. A pretty girl is singing away in a television studio in London. How is it 
possible to transmit to a televison receiver some 50 miles away — 

An instantaneous, moving picture of her appearance; and, simultaneously, 
A faithful reproduction of the sounds she is making? 
The block diagram below shows, in broadest outline, how it is done. 

Separate Sound & Vision Signals 
Transmitting L 



Aerial 



Receiving Aerial 



Camera 




Vision 
Signal - 



A Microphone V 

1 
Audio Signal 


Vision 
Transmitter 




Sound 
Transmitter 






Sound Signal 



Picture Tube 



The sound waves actuated by the singer's voice are converted by a microphone into 
electrical impulses, which are amplified and conveyed (either by land-line or by SHF 
microwave link) to a radio-frequency transmitter, and then radiated from an aerial 
as in a normal radio broadcast. This is the sound signal. 

Simultaneously, the girl's appearance is "scanned" by a television camera which is 
capable of detecting changes in the intensity of the light reflected from her person, 
and of converting these changes into electrical impulses. To these impulses (the 
picture signal) are added synchronizing and blanking pulses by a control unit in the 
studio, and the resulting video signal is fed to a separate transmitter, where it is im- 
posed on another r.f. carrier wave. The resulting vision signal is taken to the same aerial 
as that used for the sound signal; and the two modulated carriers are radiated together. 

In Part 2 of this Series you will see how the two separate and distinct signals are 
collected by the receiving aerial and passed to the TV receiver. This remarkable 
"Magic Box" converts the two signals back into sound emitted from a loudspeaker 
and into a picture displayed on a special screen. 

A TV system can therefore be broken down into six main tasks: 

1. The Conversion of Sound and Light into electrical impulses; 

2. The Combination of these signals with separate high-frequency carrier waves at a 
Transmitting Station; 

3. The Radiation of the now-modulated carriers by the Aerial System; 

4. The Reception of the sound and vision signals by the Receiving Aerial; 

5. The Separation of the two signals in the Receiver; and 

6. The Presentation of the picture signal on the Picture Screen, and the Emission of the 
original audio signal from the Loudspeaker. 



1.6 



[§1 



The Sound Signal 

The conversion of sound waves into a series of electrical impulses is a process 
already familiar to you from your reading of Basic Electricity and Basic Electronics. 
You learnt, for instance, in the very first Part of Basic Electricity that one of the six 
basic methods of producing electricity was pressure; and in the second Part of Basic 
Electronics that sound was no more than the movement of pressure waves through 
the air. 

When these sound waves reach a microphone, the variations in air pressure they 
produce actuate a diaphragm in the microphone, and cause it to produce an a.c. 
voltage. The frequency of this voltage is identical to the frequency of the sound wave 
which produced it, and its amplitude is proportional to the loudness of the sound itself. 



Sound Looks Like This 




The sound signal thus produced is amplified, and passed to the modulator stage 
of the sound transmitter. Here it is used to modulate the amplitude (or, in the case 
of an FM system, the frequency) of an r.f. carrier wave. After modulation, this 
carrier is further amplified, and is then taken through special feeder lines to a trans- 
mitting aerial — all just as you learnt in Basic Electronics. 

At the receiving end of the link, a single aerial (usually mounted on a chimney) 
detects the r.f. carriers with their audio and video modulations, and feeds them to the 
receiver. The signals are amplified, and then converted into i.f. signals by normal 
superheterodyne action. The sound i.f. signal is separated from the vision i.f. signal 
by means of special filter circuits, and passed to the sound detector. Here the 
modulation is removed from the carrier, amplified once more, and applied to the 
loudspeaker. 

In this way, the actual sounds detected by the microphone in the studio are re- 
produced in your own home. 



§2] 



1.7 



The Vision Signal— What is Light? 

Your next job is to find out how the same thing is done for Light as you already 
know how to do for Sound. Here you will find yourself on ground that is wholly 
new to you; so you must begin by learning something of what Light itself is. 

In your study of Basic Electronics, you investigated certain properties of radio 
waves. You know, for example, that a normal medium-wave sound broadcast 
makes use of radio waves having frequencies lying between 300 kHz and 3 MHz per 
second. As frequencies get higher, you pass through the short-wave sound broad- 
cast band, then through the frequencies used in television, until you reach the 
frequencies at which radio waves can be used in radar. If you have read Part 1 of 
the companion Series to this book called Basic Radar, you will know that bands 
used in modern radar range from about 225 MHz to something of the order of 
300,000 MHz. 

What happens when frequencies get higher still? Well, a good many things hap- 
pen; but one in particular is that as the frequency of the radiation approaches the 
very high figure of 400 mega-mega Hertz (4X10 14 Hz), it begins to take a form 
which the human eye is able to pick up as Light 

Light reaches your eye, as you will see on the next page, in a number of different 
forms. But the point to grasp now is that your eye can detect light waves only over 
a comparatively narrow, strongly-defined band of frequencies — from about 4 x Vfi* 
to about 7-5 X 10 14 Hertz. This range is known as the waveband of visible light. 
You can see the position occupied by this narrow waveband in the full Electro- 
magnetic Spectrum (as it is called) illustrated in the chart below. 






ttl* 




T 

10" 10" 

FREQUENCY (Hz) 

You will see from the above that the electromagnetic radiations which your eye 
picks up, and which your brain recognizes as Light, are essentially the same pheno- 
menon as radio waves, but on a different scale. It is not surprising, therefore, that 
they obey the same physical laws in their behaviour. 

You would not, in fact, be very far wrong if you looked on the human eye as a 
specialized kind of radio receiver — one which is capable of receiving radiations of a 
certain (ultra-short) wavelength and of converting them into signals which it passes on 
to the brain. 



1.8 [ §2 

What is Colour? 

You have just seen that the waveband of visible light extends over a narrow range 
of frequencies. Within this waveband, different sub-ranges of frequencies convey 
to your eye differing sensations — and these differing sensations we know as Colour. 

Light of the lowest frequency visible to the human eye appears to us as red. 
(Radiations having frequencies just below that of "red light" transmit to us the 
sensation we call heat. You have heard of the "infra-red cooker". Infra is Latin 
for below; so if you have equipped your wife with a cooker of this kind, you may be 
interested to know that your mid-day meal next Sunday will have been cooked by 
radiations having a wavelength just below those which enable your eye to detect an 
ordinary G.P.O. letter-box outside the window!) 

In order of increasing frequencies above Red, Light presents itself to the eye in a 
number of gradually differing hues. Some people can detect as many as a hundred 
of them; but the seven principal ones, called the spectral colours, are (still in ascending 
order of frequencies) : Red, Orange, Yellow, Green, Blue, Indigo and Violet. These 
seven are the main constituents of what is called the spectrum of visible light. You 
can inspect this spectrum if you shine a strong white light through a triangular glass 
prism on to a piece of white paper. Or you can look carefully the next time you 
see a rainbow. . . . 

Beyond violet, radiations of still higher frequencies, though you cannot see them, 
can nevertheless damage your eyes. That is why people receiving therapy (or merely 
"cooking" their skins to a becoming shade of brown!) by means of "ultra-violet ray" 
lamps need to wear special spectacles to exclude the damaging radiations with their 
"beyond-violet" frequencies. 




So would anybody who risked exposing their eyes to the so-called "X-rays", which 
are merely radiations having higher frequencies still. 

To sum up briefly, then, so far. Radio and radar waves, heat, light in its various 
colours, and X-rays — all these are alike in that they consist of electromagnetic 
radiations of different wavelengths, but otherwise of the same basic type. Once you 
have grasped this cardinal fact, the problem of converting light into an electrical 
impulse begins to look a good deal less formidable that it once did. 



§2] 1.9 

How Light is Produced 

You will remember that the atoms of every element consist of a nucleus, and of a 
number of electrons circling in orbit round it — the number of electrons (and therefore 
the balancing positive charge on the nucleus) being different with every one of the 
92 natural elements, and with all the eleven synthetic elements which Man has since 
added to that number. 

These electrons circle the nucleus of their atom in fixed orbit paths, or "shells", 
each at a varying distance from the nucleus and each containing (in conditions of 
stability) a fixed number of electrons. Two electrons complete the innermost shell of 
all atoms, for instance, and eight (in atoms which have 10 or more electrons) complete 
the next one out. 

All atoms fill their innermost orbits first, and hate leaving these inner orbits incom- 
plete. They therefore betray a strong desire to "pull in" outer electrons to fill the 
place of any they may lose from an inner orbit. 

You learnt in Part 2 of Basic Electricity that the outermost electrons of some atoms 
are fairly easily stripped away from their atom altogether, and become free. (This 
is particularly liable to happen if they are travelling in an orbit which is incomplete.) 
But a very much greater "kick" of energy is needed to knock away from an atom 
one of its inner electrons — either into an orbit more remote from the nucleus, or even 
away from the atom altogether. 

When such a kick arrives, an outer electron instantly drops into the place left 
empty in the inner shell. As it does so, it has to adjust itself quickly to the conditions 
ruling in its new orbit. In particular, being now in a narrower orbital path round the 
nucleus, it has to slow down. One of the ways in which a moving body can slow 
down is to give up energy. Your car slows down by giving up energy to its brakes in 
the form of heat ; 

m Electrons slipping into an Inner Orbit | 
jj give up Energy in the form of High- J 
■ Frequency Radiation. K 

The great German physicist, Max Planck, showed in the early years of this century 
that the greater the amount of energy surrendered by an electron slowing down, the 
higher the frequency of the resulting radiation would be. 

Now different atoms, as you know, have different numbers of electrons circling in 
orbit round their nucleus. When one of these electrons moves into an inner orbit, 
it surrenders a quantity of energy which varies with the proximity to the nucleus of the 
orbit into which it moves. Generally speaking, the closer in to the nucleus the 
movement takes place, the greater the quantity of energy surrendered — and therefore 
the higher the frequency of the radiation given off. 

You know that heat, light in its different colours, and X-rays are all radiations of 
different frequencies. All are produced by the movement of electrons into vacancies 
caused in the inner orbits of different atoms by a kick of energy coming from outside. 



1.10 [§2 

How Light is Produced (continued) 

Take as an example of the process described on the last page radiations on two 
frequencies within the waveband of visible light. It has been found that agitation of 
the electrons circling the nucleus of the element sodium will produce a wave of light 
having a frequency in the range of yellow/orange; while mercury produces light of a 
steely blue colour. You can see the difference if you look at sodium-filled street 
lamps, and compare their light with that given off by mercury-filled ones. 

But whatever the colour of the light to which agitation of the atoms of any material 
gives rise, the strong "kick" of energy needed to dislodge electrons from the inner 
orbits of these atoms has got to come from somewhere. 

By far the most important source of this extraneous energy available on Earth is 
the Sun. Electromagnetic waves generated in the Sun by its own tremendous internal 
heat travel the 93-odd million miles to Earth, and still have enough energy left to 
agitate the atoms of elements on which they fall, and to make them give out radiations 
of their own on their own particular wavelengths. This is what you really mean 
when you say that you perceive objects "by daylight". You see them because your 
eye detects radiations knocked out of their surface atoms by the sharp kicks of energy 
they receive from the light of the Sun. 

Where does Pure White Light Come From ? 

When you were reading, a page or two back, the list of the spectral colours which 
make up the spectrum of visible light, you may have thought it odd that white did 
not figure on it. White is not in fact a spectral colour at all. Oddly enough, it is a 
combination of a great many other colours ; and true white light is produced only by a 
combination of every single frequency within the entire waveband of visible light. 

The principal source of pure white light known to us is the Sun, and (if you ignore 
minor imperfections in the purity of its light caused by its own atmosphere) you can 
say that 



The Pure White Light of the Sun 

Contains the Frequencies ef 

Bvery Co/our in the Visible Spectrum 



§2] 



1.11 



How Light is Produced (continued) 

You saw on the last page that the light-waves from the Sun are capable of agitating 
all the atoms whose resulting radiations produce in your eye the sensation of seeing 
"every colour in the rainbow". But what happens when the Sun is not there? On 
a pitch-dark night, for instance, there is no radiation from. any source capable of 
agitating the surface atoms of any object around you— and you will see nothing at all. 

For thousands of years past, Man has unconsciously solved the problem of supply- 
ing the initial kick of energy needed to produce light by the same basic means as is 
used in the Sun— namely, by the application of heat. At first, probably, by using a 
blazing torch ; later by setting fire to some sort of wick set in some sort of oil or 
grease; more recently still, by passing an electric current through a piece of conductor 
wire and so heating it. 

But although Man has got fairly close to rivalling the perfect whiteness of the 
Sun's light with some of his recent inventions, he has not done so yet; and no artificial 
light yet produced will enable your eye to see all the colours of the visible spectrum in 
their exactly correct proportions. 

Look, for instance, at a red pillar-box by the light of a mercury-filled street-lamp. 
It will hardly look red at all. For a pillar-box appears to us as "red" only when the 
light shining on it contains the frequencies which are capable of agitating the atoms of 
those elements on its surface which in turn are capable of giving off red light. The 
pillar-box, in other words, "reflects" only the "red frequencies", and "absorbs"— 
i.e. fails to reflect— all the rest. If the light striking the pillar-box contained no red 
at all (and if the spectrum of visible light were more sharply defined than in fact it is), 
you would barely be able to post your letters until the pillar-box was painted some 
other colour; for you would have great difficulty in finding it. 

That is why things appear to change in colour when the character of the light falling 
on them changes. You know (or, if you don't, your wife will) how different the 
colours in a frock or a piece of material can look under different conditions of light 
— and how important it is to inspect them, before buying, in those conditions of 
light in which they will most often be seen. 




What the woman in the picture is actually doing (though she probably doesn't 
realize it) is trying to get the light-waves falling on the bit of stuff she is holding to 
include all those of the correct frequency ! 



1.12 [§2 

How Light Becomes a Signal 

You have seen how one form of energy — Heat — is used to create another form of 
energy — Light. You must now see how the energy of light is converted into a third 
form of energy — Electricity. For you know that the Vision Signal you are interested 
in is itself an electrical impulse. 

There exist in Nature certain elements which behave in rather an odd way when 
rays of light fall on them. They are, in other words, sensitive to light — or photo- 
sensitive, as the effect is called. 

The two kinds of photo-sensitivity with which you will be concerned in this Series 
are photo-electricity and photo-conductivity, for both phenomena are used in modern 
TV camera tubes. (The third main kind of photo-sensitivity, by the way — it is called 
the photo-voltaic effect — is put to use in the solar cells which continuously recharge 
the batteries of many of the satellites which are now orbiting the Earth.) 

Photo-electricity 

When rays of light fall on a surface coated with certain elements, the energy of the 
rays knock electrons right out of the coating material, and sets them free. Examples 
of such photo-electric (or photo-emissive) elements are zinc, potassium, sodium and — 
especially — caesium. One of the most sensitive photo-electric materials yet developed 
is a strip of metal coated with silver oxide, on top of which a thin film of caesium has 
been deposited. 

An important application to which a strip of metal treated in this way has been 
put is in the photo-electric cell (or photocell, as it is generally abbreviated), about 
which you must now learn. The illustration on the right below shows what an 
operational photocell looks like. On the left appears the symbol by which a photo- 
cell is conventionally represented in electrical diagrams. As you will see on the 
next page, it is essentially a representation of what the photocell looks like when it is 
viewed from on top. 



-Cathode^ 




totocdt 







§ 2] 1.13 

The Photocell 

You will remember that in the ordinary thermionic radio valve the cathode, when 
heated, emits electrons which are attracted to and collected by the relatively positive 
anode, and that an electric current results. 

In the photocell, a strip of metal coated on its inside surface with a film of photo- 
electric material is made into the shape of a semi-cylindrical plate. It is then placed 
close to a rod-type anode; and the whole assembly is enclosed in a glass envelope 
from which two leads are taken to the outside. 



The anode is, as before, given positive polarity; while the semi-cylindrical plate 
(which is called in this application the photo-cathode) is energized by the focusing on 
to its interior surface of rays of light. When this happens, electrons are emitted. 
They are collected by the anode, and an anode current flows. 

Within certain limits, the value of this current is linearly related to the degree of 
illumination falling on the photo-cathode— the brighter the light, the greater the current. 
Operating currents in a photocell are, however, extremely small, being typically of the 
order of 1.5 to 10 micro-amperes. 



If a load be connected in series with 
the anode of the photocell, a voltage 
will be developed across this load 
whenever the photo-cathode is illumi- 
nated. The greater the degree of 
illumination, the more current will 
flow, and the greater will be the voltage 
developed across the load. 

You will see at once that you have 
here a device capable of converting 
light-energy into electrical signals; and 
the photo-emissive principle is in 
fact used in most (but not all) modern 
TV cameras. 



Photo-cathode 



Photo-electrons 



Anode 



Direction ' "- 
of Light Rays 








tsteJJ 







Other uses for the photocell include a special type of burglar alarm. When the 
thief moves across a beam of light, which need not necessarily be within the visible 
spectrum, he momentarily shields the photo-cathode, cuts off the current and with it 
the voltage across the anode load, and so actuates the alarm mechanism. 

Another use is in a set of garage doors which open themselves when a car's head- 
lights are shone on to a conveniently situated photocell. The voltage developed 
across the anode load of the photocell operates a relay switch, which in turn starts 
up an electric motor opening the garage doors. 



1.14 [§2 

Photo-conductivity 

The other main type of photo-sensitivity of interest to you in the conversion of 
light into an electrical impulse is photo-conductivity. 

When certain materials — of which selenium is a leading example — are exposed to 
light, their resistance to the passage of electrical current is diminished. In other 
words, their conductivity improves. 

You will recall that, in the photo-electric effect, the rays of light striking the photo- 
sensitive material caused electrons to be given off from the surface of the material. 
In the photo-conductive effect, on the other hand, the rays cause electrons to be 
liberated within the material itself. More free electrons within a piece of material, 
as you know, increase its conductivity. 

The resistance of a piece of clean selenium is considerably changed when it is 
exposed to light. The amount of the change even varies with the colour of the light 
itself— a fact which should not surprise you after what you have learnt of the different 
frequencies which give rise to different colours being observed by your eye. 

Once darkness is restored, the electrical resistance of selenium reverts to normal. 



?} Selenium 
aSheet 




;-'Ohms,; 



Ohms 



The PHOTO-CONDUCTm Effect 

For TV purposes, however, there is one important disadvantage in the response of 
selenium (and of a good many other materials having similar properties) to the 
application and removal of light. 

This disadvantage is that conductivity does not change to its new value at once. 
In other words, there is a time-lag. When light is applied to it, the conductivity of 
selenium rises only gradually to its new value. When the light is removed, it reverts only 
gradually to its darkness level — dropping rapidly at first, then increasingly more slowly. 

The full time-lag between removal of the light and restoration of the old level of 
conductivity was at one time as long as a whole second. It has been greatly reduced 
by modern manufacturing techniques; but cameras using the photo-conductive effect 
are still inferior to those using the photo-emissive effect when scenes of fast movement 
are being televised. 



§2] 



1.15 



How a Camera "Sees" a Scene 

You have just seen that a photocell can convert light into electrical signals, and — 
more important still — that it can vary the strength of these signals according to the 
intensity of the light it receives. It performs this "conversion act", and the variations 
upon it, very quickly indeed — so quickly as to make it almost instantaneous. 

Fine, you may say — but how do "variations in the intensity of light received by a 
photocell" help to televise the finish of The Derby . . . ? 

Consider closely what an ordinary black-and-white "still" picture printed in a 
newspaper really consists of. Say it is a photograph of an old woman's face. 

If you look at it through a magnifying glass, you will see that the picture is really 
made up of thousands and thousands of tiny individual areas of black and white. 
The white dots present always the colour of the paper itself; the black ones are all of 
exactly the same degree of blackness. What varies is their relative density of packing. 

In the process used, which is known as half-tone photo-engraving, the black dots are 
packed so tightly together on those areas of the printed image which represent the 
darker areas of the old woman's face that they seem to coalesce. In other words, 
the black dots appear to vary in size according to the tonal content of the different 
areas of the picture. 




MAGNIFICATIOH 



HALF-TOHE 




1.16 



[§2 



How a Camera "Sees" a Scene {continued) 

Every individual black dot in the process described on the last page is known as a 
picture element. It represents the smallest detail of the picture which is capable of 
being reproduced. In the magnification illustrated, it is the smallest black dot you 
can see in the lightest area of the picture. 

Most newspaper photographs are made up of comparatively large picture elements, 
generally big enough to be visible to the naked eye. Greater clarity of detail would 
be possible if all the black dots were kept much smaller, and if their density was 
increased wherever darker areas had to be represented. But paper of better quality 
than ordinary newsprint would be necessary for this method to be successful. (Com- 
pare, for example, the quality of the photographs reproduced in The Times with those 
in a mass-circulation journal using less good paper.) 

Whatever the details of the method, the essential point of the half-tone process is 
that all the intermediate tones between black and white — that is to say, all the various 
shades of grey — can be effectively presented by a suitable arrangement of the picture 
elements, which are all pure black, on a white background of paper. 

Now go back to your Derby finish. Assume that you are photographing it in 
black and white only. The photographic print method which your camera will be 
using differs considerably from the half-tone process you have just been examining; 
but the principle on which it works is identical. Intermediate tones are represented 
by varying concentrations of minute areas of black and silver, which here does the 
job of white. 

The next step is to take a series of such photographs in very rapid succession. Each 
of them will differ slightly from its predecessor as the horses and their jockeys move 
on. Present this succession of slightly different photographs one after the other to 
the human eye — and you will have been to the cinema sufficiently often to know that 
if you do it fast enough, the illusion of smooth and continuous movement will be 
created. 

But this extra refinement in no way alters the fact that what you are looking at is 
essentially no more than a constantly varying arrangement of black and white dots. 



*J ■■/.&;■ iarei 








ir>y 


*S* 


1/ 


^SMM 


jfeSia 



The television camera uses no film on which to print what it sees. Instead, it 
translates the scene it is viewing into a pattern of electrical charges analogous to the 
black dots used in the half-tone process. The smallest of these charges represents a 
picture element, or elemental area of the scene. The job of the TV camera is to con- 
vert these elemental areas into electrical signals whose amplitudes vary with the amount 
of charge on the element. 

The ability of a photographic image-reproducing process to present the finer detail 
of a scene is known as its resolving power, and the picture is said to be "of good (or 
poor) resolution". In TV language, however, the word definition is always preferred. 



§2] 



1.17 



How your Eye Sees a Scene 

Compared with any mechanical/electronic "seeing system" yet invented, the human 
eye is a marvellously efficient optical instrument; and you should have some idea of 
how it works. 

Reduced to its basic essentials, your eye is a crystalline lens system which collects 
light of all frequencies in the visible spectrum, and focuses it on to a light-sensitive 
screen (called the retina) situated in a concave arc round the back of the eye. 



*** ****** ty e 




Optic Nerve 
?^. IIS iff 'Lens ^tV 

Cornea 

/ \^ A»^ ^dter/ ^""^s?*^^ To Brain 

Ciliary 

Muscle 

The retina itself is made up of millions of nerve fibres, each sensitive to light of a 
different frequency, which are connected in groups to the optic nerve. This nerve, 
in turn, is connected to the brain. When an image is focused on the retina, signals 
from those particular groups of nerve-fibres which are sensitive to the frequencies 
involved are instantaneously transmitted via the optic nerve to the brain; and the 
sensation called sight is created. 

The focusing of the image on to the retina is done by the lens system of the eye. 
The ciliary muscle controls the degree of curvature of the crystalline lens, and therefore 
its focal length. 

Since the nerve fibres in the retina are very delicate and could be damaged by 
sudden large changes in the quantity of light falling on them, there is situated between 
the lens itself and the outer cornea (whose job it is actually to collect the light) an 
adjustable curtain called the iris. This curtain varies in colour from person to person. 
When you say that a girl has blue, or brown, or green eyes, it is her iris that you are 
really talking about ! 

In the centre of the iris is a circular aperture called the pupil of the eye. The 
muscle which controls the iris relaxes and fully opens the pupil when the eye is 
observing a dark scene; but when strong light is present, the muscle reduces the pupil 
to little more than the size of a large pin-head. In this way the light collected by the 
cornea is controlled in amount before it reaches the lens and is focused by it on to 
the sensitive retina. 



1.18 [§2 

Persistence of Vision 

By means of the optical mechanism outlined on the last page, the eye is able to 
present to the brain, directly and instantaneously, a complete picture of the object viewed 
— its three dimensions of height and width and depth, its tonal content, and every 
shade (wavelength) of colour within the visible spectrum which the surface of the 
object has the power to reflect. 

But the eye has also another attribute of the highest importance. It is called 
persistence of vision, and it can be denned as follows : 

oooooooooooooooooooooooooooooooooooooooooo 



o 

o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 



The Human Eye possesses the ability to Retain an 
Impression of the Shape, the Colour and the 
Brightness of an Image for a Fraction of a 
Second after Light from the Image has Ceased 
to be Received. 



o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 
o 



oooooooooooooooooooooooooooooooooooooooooo 

This property of the eye plays an essential role in the reception of both cinematic 
and television images ; for it means that the illusion of a continuous picture can be 
obtained from a series of individual images, each differing slightly from its predeces- 
sor, presented to the eye in very rapid succession. 

You can easily demonstrate to yourself the principle of persistence of vision by 
marking out near the rim of a stiff paper disk, some four or five inches in diameter, 
a series of heavy black dots, and then rotating the disk rapidly about its centre. As 
good a way as any of doing this is to "play" the disk on the turn-table of a record- 
player. You will see that, above a certain speed of revolution, the individual dots 
lose their separate identities, and appear to merge into a continuous grey circle. 





fJSSSSSsJsBs! 






DISK AT REST 



DISK IN ROTATION 






§2] 

REVIEW of the Basic Principles of TV 



1.19 



Visible Light is an electromagnetic radiation of the same general nature as a radio 
wave, and obeys the same physical laws. It is, however, of much higher frequency. 

The electromagnetic radiation which gives rise to Visible Light is occasioned when a 
"kick" of energy (generally heat) from an outside source displaces an electron from 
an inner shell of an atom. An electron from an outer shell promptly drops into the 
vacant space, and in doing so surrenders energy in the form of a radiation having a 
wavelength lying within the Waveband of Visible Light. 



The Waveband of Visible Light is a 
narrow band of frequencies higher than 
those which give rise to the phenomenon of 
heat, but lower than the frequencies of the 
ultra-violet and X-rays. It lies between 
about 4 x 10" and 7*5 x 10" Hertz. 



i i 

S ! 

I t 



hi! 






r»« 




The Spectral Colours within the Waveband of Visible Light are (in order of increasing 
frequency) Red, Orange, Yellow, Green, Blue, Indigo and Violet. Pure white light is 
a combination of every frequency in the visible spectrum. 

Objects appear to the human eye to be of a certain colour only when the light falling 
on them contains the frequencies which give rise to that colour. 



The Photo-electric Effect occurs when rays of light strike a surface coated with 
certain elements— notably caesium— and the energy of the rays knocks electrons right 
out of the coating material and sets them free. 

The Photocell is a device for turning 
Light into electrical signals by making use 
of the Photo-electric Effect. A rod-type 
^ "^8 anode is enclosed in a glass envelope with 

^iotoctlU ^^^^ a semi-circular cylindrical plate (called the 

photo-cathode), which is coated on its inside surface with a film of photo-electric material 
and is placed close to the anode. This anode is connected to a battery so as to be made 
positive with respect to the cathode. When light is focused on to the interior surface of 
the photo-cathode, electrons are set free and are attracted to the positive anode. An 
anode current then flows. 




The magnitude of this current depends on the intensity of the light striking the coated 
surface of the photo-cathode, and varies almost instantaneously with it. 



1.20 

REVIEW of the Basic Principles of TV (continued) 

The Photo-Conductive Effect. When 
certain materials, notably selenium, are 
exposed to light, their resistance to the 
passage of electrical current is diminished 
and their conductivity improves. When 
the light is removed, the conductivity of the 
material returns to its former level. The 
effect thereby provides another means of 
converting light of varying intensity into 
an electrical signal of varying amplitude. 



[§2 




Tie MI*TO-CMMCTt¥e tlftct 



The changes in conductivity of the photo-conductive materials do not, however, take 
place instantaneously; and the resulting time-lag is a disadvantage in TV cameras using 
this effect when they are used to record fast-moving scenes. 



The Picture Element is the smallest area in any picture which it is possible to present 
in any process of image reproduction. In the familiar photographic print, the picture 
elements are minute grains of light-blackened silver, clustered either thickly or less 
thickly on the darker and lighter areas of the picture respectively. In the newspaper- 
type half-tone, they are black dots. In the TV camera, they are minute electrical 
charges. 



In all systems, it is the size of the picture elements which determines the amount of 
detail in the scene which the system is capable of resolving. The smaller the picture 
elements, the greater the definition of the system. 



Definition. The ability of any photographic image-reproducing process to present 
the finer details of a scene is known as its resolving power or definition. 




Persistence of Vision. The human eye 
possesses the ability to retain an impression 
of the shape, the colour and the brightness 
of an image for a fraction of a second after 
light from the image has ceased to be 
received. 



§3: SCANNING 



1.21 



If a television system is to operate effectively, it must be able to produce at some 
distant point an apparently instantaneous and continuous record of all the essential 
details comprising the scene being televised. Ideally it should reproduce an image 
identical in every respect to that observed by the eye. 

But you have just seen that the human eye is capable of examining simultaneously 
all the structural content of the image focused on the retina, and of conveying this 
information without delay to the brain. If a television system is to be able to emulate 
the eye, it too must perform a simultaneous and complete examination of the entire 
scene, and reproduce it at the receiving end of the link. 

Although such a feat is theoretically possible, it is worth spending a moment or 
two looking at the type of equipment that would be necessary to perform it, to see 
why something more practical has to be devised. 

Take an exceedingly simple scene — merely a letter 
H painted in shiny white paint on a black back- 
ground. It is indicated by the figure {) in the illus- 
tration across pages 1.22 and 1.23, and is strongly 
illuminated by direct light from the lamp Q . 
This light is reflected Q from the scene, much 
more strongly from the bright, smooth surface of 
the H than from the black, matt surface of its 
background. 

Directly facing the scene is a panel Q holding enough photocells (only 25 are 
actually shown) to be able to cover every square inch of the scene. Each photocell 
is so recessed in an open-ended tube that it is shielded from all light other than that 
reaching it from the area of the scene it is "watching". 

The light accepted by each photocell is converted by it into a separate electrical 
signal — the more intense the light, the larger the amplitude of the signal; and these 
signals are conveyed, each by a separate wire Q , to a bank of 25 amplifiers Q . 
These amplifiers are needed to "step up" the signals to a strength adequate for feeding 
into an array of 25 more wires A — and very long ones this time — connected to the 
distant receiver. 

Note that it is at the point where the 25 separate signals call for 25 separate ampli- 
fiers that the bulk and cost of this theoretical arrangement begins to become heavy. 

Note also that the 25 long wires could in theory be replaced by 25 separate trans- 
mitters which would radiate the signals as radio-frequency waves. This would 
supply the signal to many receivers simultaneously ; but it would call for an enormous 
frequency bandwidth for the 25 simultaneous transmissions to be possible without 
serious mutual interference. 




1.22 



[§3 



Why "Simultaneous TV" is Impracticable 

When the signals of this theoretical TV system reach the receiver, they will again 
need amplification. You have, therefore, a second bank of 25 amplifiers (j), 
connected to a panel Q on which are mounted 25 lamps, each focused on to a 
small area of the viewing screen {Jj) . The larger the amplitude of the signal leaving 
each amplifier, the brighter will shine the lamp it controls, and the stronger will be 
the beam which that lamp throws on to the viewing screen. 

Every lamp at the receiver is carefully connected to that photocell in the transmitter 
which occupies the same position on its panel as the lamp in the receiver does on its 
panel; so that the more strongly a photocell sends a signal on its way, the brighter 
will glow the corresponding lamp on the panel in the receiver. 

In this way, a letter H — of a sort — will be thrown on to the viewing screen ; and 
the system as a whole can be said to work. 

But you will note that the H at the viewing end is formed of eleven rather ill-defined 
"areas of illumination" only, and is consequently very inferior in definition to the 
original H in the scene. This inferiority would have been even more marked if the 
scene had been more complex. It would, for instance, have been difficult to re- 
produce a convincing letter C by this process — while as for the finish of the Derby . . . ! 



THBORBTICAL SMULUNBOUS 
o 



Lamp 



transmitter 



25 Amplifiers 




Reflected 
Light 



25 Connecting Wires 
(Not all shown) 



25 Connecting Wires 
(Not all shown) 



§3] 



1.23 



Why "Simultaneous TV" is Impracticable (continued) 

This inability to resolve fine detail is, of course, principally due to the compara- 
tively small numbers of lamps and photocells used in the "system" illustrated. If 
the number of both were substantially increased, the resolution of the system would 
be somewhat improved. 

A "revised letter H" could, for example, be made up of (say) 100 areas of illumina- 
tion, instead of only 25. This would imply a battery of something of the order of 
500 photocells to "watch" the whole of the scene simultaneously, plus two banks of 
500-odd amplifiers apiece, plus all the associated wiring, plus a battery of 500 lamps 
to throw their varying amounts of illumination on to that part of the viewing screen 
at which they were aimed. Quite an undertaking already . . .! 

Yet even the improved letter H produced with the aid of all this paraphernalia 
would not approach in definition the coarsest of half-tones. To do even as much 
as that, many thousands of photocells, lamps, and their separate amplifiers and 
ancillary wiring would be required — and the bulk and cost of the resulting apparatus 
would be quite prohibitive. 

For a TV system to be at all practical, something better is needed. You are now 
ready to learn about the methods which have been devised. 

IMAGB'R£PROdUCIHG SySTiM 



Viewing Screen 



reo 




O Lamps 



U 25 Amplifiers 



1.24 



t§3 



Sequential Scanning 

The disadvantages of the primitive image-reproducing system you have just been 
looking at can be greatly reduced by making use of that peculiar property of the eye, 
persistence of vision. For if signals from the photocells in the camera, instead of 
being all transmitted simultaneously, can be transmitted one after the other in very 
rapid succession indeed, persistence of vision in an eye at the receiving end will create 
the illusion that the image there observed is really made up of a large number of 
simultaneously-produced areas of illumination. 

The method which has been devised of transmitting the photocell signals in this 
very rapid succession is called sequential scanning. It is one of the basic principles 
on which all television systems work. 

In the illustration below and opposite, every one of the 25 photocells in the trans- 
mitter is connected to a fast-moving scanning switch which picks up, one after the 
other, signals corresponding to the brightness level of the area of the scene which 
each individual photocell is "watching", and feeds it to a common amplifier. In the 
diagram, the switch is shown as collecting the signal corresponding to the light-output 
of Photocell No. 4, but it must be thought of as moving round very fast indeed. 

Connections to the switch are arranged so that signals are collected from the photo- 
cells in sequence — top line first, from left to right; then down to the second line from 
left to right; and so on from top to bottom. 







25 Photocells 



Transmitting 
Aerial 



25-way Scanning 
Switch 
(Only 3 Connecting 
Wires Shown) 



Amplifier 



Radio 
Waves 



§3] 



1.25 



Sequential Scanning (continued) 

When the switch reaches the connection to Photocell No. 25 in the bottom right- 
hand corner, it flies back to No. 1 and begins the cycle all over again. 

The output from the common amplifier is fed to an aerial, and transmitted as a 
modulated r.f. signal. 

At the receiver, the signals picked up are amplified; and then connected to another 
switch, synchronized with that in the transmitter, which is in turn connected to a bank 
of 25 lamps arranged in the same geometrical pattern as the photocells. The brilliance 
of Lamp No. 4 is thus controlled by the signal produced by Photocell No. 4 and so on ; 
and the area of the viewing screen which is illuminated by every lamp corresponds to 
the corresponding area of the transmitted scene. 

One obvious advantage of this system over the more primitive one you have seen 
is that the equipment needed is very much reduced. Another is that normal radio- 
communication methods can be applied to the single link between transmitter and 
receiver. A third is that a large number of receivers can be served without serious 
problems of frequency bandwidth. 

The key to the system is, of course, perfect synchronization of the scanning switch 
in the transmitter with the lamp selector switch in the receiver; for without this, much 
distortion of the reproduced image will occur. You will learn how this problem of 
synchronization is solved later on. 

Viewing 
Screen 






Receiving 
Aerial 



° o ° x_ 

0®0°?o 

00 i 



-qOOOOo 




25 Lamps 
Brilliance Controlled 
by Signal Received 



o 

o 



25 
25-way Scanning 

Switch 
(Only 3 Connecting 
Wires Shown) 



1.26 [§3 

Flicker 

Provided that the rate of sequential scanning is high enough, the eye can be success- 
fully "tricked", by reason of the persistence of its vision, into believing that a very 
rapidly renewed image on the viewing screen has in fact been there all the time. 
At the level of picture brilliance regarded as acceptable by the average TV viewer, 
the persistence of vision of the normal eye has been found to be about one-fiftieth 
of a second; so that the optical illusion necessary for any television system to work 
can be achieved if every individual area of the viewing screen can be illuminated by 
its appropriate lamp not less than 50 times per second. 

At any rate of scanning lower than this, a phenomenon known as flicker develops. 
For flicker to be avoided in the primitive image-reproducing system, the entire bank 
of photocells would have to be completely scanned in time for the first (top left-hand) 
area to be re-scanned before the image of it had "died" in the viewing eye. If this 
rate of scan were not achieved, every succeeding image would impinge on an eye 
which was receiving no light at all; and such a succession of "light/no-light/light/no- 
light" would obviously produce in the viewing eye a tiresome flickering effect. 

The achievement of a rate of scan high enough to overcome nicker was one of the 
major problems which faced the pioneers of TV. The earliest attempts involved the 
mounting of a number of differently-angled mirrors round a rotating shaft. One 
of the very earliest — the so-called "Mirror Drum Apparatus", designed in 1882, can 
be seen to this day in London's famous Science Museum in South Kensington. 



Simplified arrangement of scanning-disk 

...MThe Transmitters 




Photocell 



Transmitting 
J Aerial 



3] 



1.27 



The Nipkow Disk 

Another mechanical scanning device much used by the television pioneer John 
Logie Baird was the so-called Nipkow Disk, named after the Russian scientist, Paul 
Nipkow, who designed it in 1884 and patented it in Berlin. 

Nipkow's device consisted of an opaque disk, towards the periphery of which a 
number of small holes were drilled in the form of a single turn of a spiral. The 
scene to be televised was focused on to a small area on the circumference of the disk 
exactly wide enough to cover every hole in the spiral. Behind this narrow band was 
placed a photocell. 

As the disk rotated, the outermost hole scanned a narrow strip of the right-hand 
side of the scene, doing so from bottom to top; and light reflected from scene elements 
in this strip reached the photocell. As soon as this hole had scanned to the top of 
the scene, the next hole took over in the bottom right-hand corner again, and scanned 
upwards on a line slightly to the left of the first. 

This procedure was repeated with every hole in turn — the fact that each was slightly 
inset from its neighbour ensuring that the entire scene was scanned in every revolution 
of the disk. 

Light striking the photocell was converted, in the normal way, into a series of 
electrical signals, each proportional to the tonal value of the scene element which 
produced it. These signals were then amplified and transmitted, one after the other, 
to the receiver. There they were arranged to control the brilliance of a lamp placed 
behind a second disk, identical to and synchronized with the first, which revolved so 
as to allow light from the lamp to pass through its holes on to the viewing screen. 
In this way, an image of the entire scene was built up. 

method of television communication 







The Receiven 



PI 

ft* :i 



Viewing Aperture 




Scanning Disk 

(Synchronised with Transmitter Disk) 



1.28 [§ j 

Electronic Scanning 

All the mechanical scanning systems just mentioned had three great disadvantages. 

The first was that they made very uneconomical use of the amount of light illuminat- 
ing the scene; for only a small fraction of the total light reflected from it ever reached 
the photocell. The result was that, for an acceptable level of light to get through to 
the photocell, extremely powerful lights had to be used. This often made things very 
unpleasant for the artists, whose make-up would often melt in the heat. 

The second disadvantage was that, in order to get a picture of any reasonable 
definition, a great many peripheral holes were needed. This meant that the Nipkow 
disk had to be made very large ; and then had to be made to rotate very fast to avoid 
flicker. 

The third difficulty lay in achieving synchronization of the disk at the receiver end 
with that at the transmitter. Men still alive tell of how synchronization was achieved 
in the early days by the varying pressure of a thumb against the surface of a spinning 
disk! 

As you will learn in the next few pages, one of the principal factors in determining 
the degree of definition of a televised picture is the number of lines in the scanning 
period. In early BBC transmissions, which had to make use of mechanical scanning 
systems for want of anything better, it was seldom more than 30. In a 30-line system, 
the number of elemental areas which it is possible to scan is about 1,000. This is not 
enough to provide an acceptable picture. Some totally different system was needed. 
It was found by using the cathode ray tube (CRT) which you learnt about in Section 
1 1 of Basic Electronics, Part 5. 




The CRT 



BEAM OF 
ELECTRONS 



COATING OF FLUORESCENT 
MATERIAL 



SPOT FORMED BY 

ELECTRONS 

STRIKING 

FLUORESCENT 

COATING 



You know that the mass of an electron is almost infinitesimally small. It weighs, 
in fact, about 9 x 10 ~ 28 of a gramme — a figure which may mean more to you if you 
write it down as a decimal point followed by 27 noughts and then the figure 9 ! A body 
as small as that acquires almost no momentum (which is the product of its mass times 
its velocity) even when it is travelling exceedingly fast. It can therefore be made to 
move and to change direction at a speed immeasurably greater than is possible for 
any mechanical moving part — -at a speed, in fact, comparable to that of light itself. 



§3] 



1.29 



Electronic Scanning (continued) 

In the electronic scanning system, the scene is observed by the TV camera, and 
projected on to a light-sensitive target contained in a special type of cathode-ray tube 
called a camera tube. You will learn in the next Section that this target is the effective 
equivalent of a large number of photocells set very close together. 

A beam of electrons is then made to sweep across the target, very fast indeed, in a 
series of horizontal lines running from side to side, and from top to bottom. As 
the beam scans the various elemental areas of the target, a succession of electrical 
signals is produced, which are then amplified and transmitted to carry the picture 
information to the receiver. 

At the receiver end, the received signals are made to control the intensity of a 
second electron beam, synchronized with the first, as it in turn scans the fluorescent 
surface of another CRT behind the viewing screen of the receiver. 

This second CRT, with its fluorescent surface, is called the picture tube. Here 
is what it looks like — sideways on, and from in front. 




Timebase 



Electron 
Beam 



You will remember from what you learnt in Part 5 of Basic Electronics that when the 
fluorescent screen of a CRT is bombarded by a stream of fast-moving electrons, the 
area of the screen struck by this beam gives off light; and that the more intense the 
beam, the greater the degree of fluorescence which will ensue. 

You will also recall that means have been devised whereby the beam can be narrowed 
to a single spot at the moment it strikes the screen; and that if the beam is deflected 
very quickly across the screen, this spot will appear as a continuous line. Its apparent 
continuity is caused partly by the persistence of fluorescence (the "afterglow") in the 
CRT. and partly by persistence of vision in the human eye. 

This continuous line of light moving across the CRT in the receiver of a television 
set is known as the timebase, the trace, or the scan. 



1.30 [§3 

The Scanning Raster in the Receiver 

In the picture tube of the television receiver, the spot is made to move across the 
screen very rapidly indeed from left to right, and at the same time in a series of hori- 
zontal lines from top to bottom. When it reaches the bottom right-hand corner of 
the screen, it is returned very quickly to the top left-hand corner, and the scanning 
cycle is repeated. 

Tf this sequence is repeated fast enough (about 50 times per second) the whole 
screen will have been scanned, and the fluorescence of the top line will have been 
renewed by a second scan before the light of the first scan has had time to fade in the 
eye of the viewer. 

The image presented will be that of a series of parallel lines of light running very 
close together almost horizontally across the screen. This presentation — parallel 
lines of light having no picture content — is known as the scanning raster. 



Scan Starts Here 




tie 



sc/?nwm# 




Flyback Lines Scan ends Here 



In the picture of the raster the number of lines has been much reduced, for greater 
clarity. Note that during the unwanted (right-to-left, and bottom-to-top) move- 
ment of the spot — known as flyback periods — the electron beam is in practice sup- 
pressed altogether, and produces no trace. 

You will see that the motion of the spot when tracing out the raster is very like the 
movement of your eye as you read this printed page. Your eye starts at the left-hand 
top corner of the page, and scans the first line from left to right, fairly slowly. It 
then moves back very quickly to the beginning of the second line at the left of the 
page, "flying back" relatively very fast because it has no reading to do on the way. 
It then scans the second line as before. The sequence is repeated line by line down 
the page, until the last word in the bottom-right-hand corner is reached. 

One further point may look obvious, but you will see its importance later on. The 
depth of the type-area on this page is about a third greater than its width. But your 
eye will take much more than a third longer to scan the whole page than it will to scan 
a single line of it. In other words, its "page-scanning rate" is much longer than its 
"line-scanning rate". 



§3] 1.31 

Modulating the Raster 

You have just seen that the raster, in its normal state, produces only a series of 
parallel lines of light running horizontally across the screen, and presents no picture 
detail at all. But you also know that the intensity of the electron beam which produces 
the raster can be varied by the strength of the signals received from the transmitter. 

When this variation of signal strength occurs during the movement of the beam 
across the screen, the brilliance of the spot varies also — from very bright to almost 
dark, or to any degree of brightness in between — with the changes in brightness suc- 
ceeding one another with enormous rapidity. 

Tt is by building up an enormous number of rapidly-varying but controlled changes 
of bright-up, accurately synchronized with the scan at the transmitter, that the CRT 
in the television receiver can be made to trace out across the screen a picture of the 
image transmitted. 

In the illustration below, it is assumed that the intensity of the electron beam has 
been increased for nine regularly-spaced long-and-short periods during a single trace. 
The resultant scan will be punctuated by nine regularly-spaced long-and-short bright- 
ups, as they are called. (In an actual picture, of course, bright-ups will never be 
regularly-spaced longs-and-shorts, but will vary in duration and intensity with the 
tonal content of the scene.) 



A Difficulty of Bandwidth 

You might well think, after reading the above, that it was normal TV practice to 
make the electron beam scan the screen of the picture tube (and the corresponding 
beam scan the target in the camera tube) in a consecutive series of horizontal lines from 
top to bottom. 

It is indeed possible to produce a quite satisfactory image by this method; but in 
practice there arises a difficulty. 

You know that, if flicker is to be avoided, the rate at which a series of images must 
be presented to the eye is of the order of 50 times per second. So if flicker is to be 
avoided on the TV picture tube, the complete picture must be presented and re- 
presented to the eye at a rate no lower than that. 

But, as you will see later on, a picture-repetition frequency as high as 50 per second 
would call for a very wide frequency bandwith for the transmission of the video signal, 
and would therefore greatly restrict the number of channels which could be accom- 
modated within a given frequency band. 




1.32 [§3 

Interlaced Scanning 

An ingenious way has been found of getting round the difficulty described on the 
last page. It is called interlaced scanning. 

Instead of the target in the camera tube and the screen in the picture tube being 
scanned in consecutive lines, the beam is first made to scan all the odd-numbered lines 
in their proper order — that is to say, Lines 1, 3, 5, 7, 9, etc., down to the bottom line 
of all. It then scans all the even-numbered lines — 2, 4, 6, 8, 10 and so on — down to 
the end of the penultimate line of the raster. From there, it flies back to the beginning 
of Line 1, and the process is repeated. 

For reasons which you will see later on, the beam is not allowed to complete the 
bottom (odd-numbered) line before starting to scan the topmost even-numbered line. 
Instead, it is stopped half-way along the bottom line and made to fly more or less 
vertically upwards (with its trace of course suppressed) to half-way along the topmost 
even-numbered line, to resume its horizontal trace from that point. 



Start of Odd-line Scan © 



odd -line 



© Start of Even-line Scan 




Odd-line 
Flyback 



Even-line Flyback 



End of Even-line Scan 



End of Odd-line Scan 



This means, of course, that two vertical sweeps of the raster have to be made by 
the electron beam to produce one complete picture on the screen. The first half- 
picture is produced by the scanning of odd-numbered lines only; the second half- 
picture is produced by the scanning of even-numbered lines only. This second half- 
picture is then superimposed on the first, to make up the complete picture. 

You will see the advantages of this technique on the next page. 

Every half-picture presentation is nowadays called a field (// used to be called a 
"frame" in Britain; but this term has now been officially superseded). Every whole- 
picture presentation is called a picture. 

The number of half-pictures presented in every second is called the field frequency, 
and the number of complete pictures presented in every second is called the picture 
frequency. Since two fields equal one picture, the field frequency is always double 
the picture frequency. 



3] 



1.33 



Interlaced Scanning (continued) 

You know that a sequence of images must be presented to the eye at the rate of 
about 50 per second if nicker is to be avoided; but you have seen that a picture 
frequency as high as 50 per second presents problems of bandwidth. How does the 
technique of interlaced scanning help to get over this difficulty ? 

You know that the two fields making up one picture are superimposed on one 
another very quickly. You also know that the fluorescent coating on a picture tube 
screen has afterglow. And you will have realized, no doubt, that the vertical distance 
between two adjacent lines of a scan is optically tiny. 

The outcome of these three factors is that any line not being scanned at a given 
moment is closely sandwiched in between two lines, above and below it, which are 
being scanned at that moment. The even-numbered lines of Field No. 2 thus carry 
on the work of presenting a continuous picture which was begun by the odd-numbered 
lines of Field No. 1 ; and effective continuity of vision is achieved when only 25 scans 
of Field No. 1 plus 25 scans of Field No. 2 are completed in every second. 

The result is that a continuous picture without flicker can be achieved by means 
of interlaced scanning at a picture frequency of only 25 per second. 






Field No. I . Odd Lines 



Field No. 2. Even Lines 



3W t6e Pietune c4 Surftufc 



i 

2 
3 
■4 
5 
6 
7 
8 
9 





Field I + Field 2. Complete Picture 



1.34 



[§3 





OVD-UNE SCAN 




2S KtfdtUiow j4\ 
£\/eN-UN£ SCAN 







oohplsk piciufze wjhouj fuctcez 



§ 3] 1-35 

Scanning Frequencies 

It is possible to design TV systems making use of many different numbers of 
scanning lines per picture. As you probably know, both British television networks 
at present use a system based on 405 lines to the picture — a number chosen in the 
early days of TV back in 1936. 

In the course of the next few years, however, it is planned to change over gradually 
to a system based on 625 lines to the picture — as recommended by an internationally 
recognized body, the Comite Consultatif International des Radiocommunications 
(CCIR), as long ago as 1950. The change-over has already begun with the intro- 
duction of "BBC 2". 

The number of lines currently used in some national TV systems is : 



405 United Kingdom, Eire. 

525 U.S.A., Canada, Mexico, Cuba, Bermuda, Puerto 

Rico, Haiti, Trinidad, Costa Rica, Panama, Colom- 
bia, Peru, Brazil, Uruguay, Hawaii, Japan, Korea, 
Philippines, Cambodia, Thailand, Saudi Arabia, 
Kuwait and Iran. 

625 United Kingdom, Eire, France, Belgium, Holland, 

East & West Germany, Norway, Sweden, Finland, 
Denmark, Poland, Czechoslovakia, Switzerland, 
Austria, Hungary, U.S.S.R., Spain, Portugal, 
Italy, Yugoslavia, Turkey, Lebanon, Syria, Cyprus, 
Egypt, Morocco, Nigeria, Ghana, Kenya, Rhodesia, 
Iraq, China, Australia, New Zealand, Venezuela 
and Argentina. 

819 System becoming obsolescent, but has recently 

been in use in France, Belgium, Luxembourg, 
Monaco, Algeria, Tunisia and the Ivory Coast. 



Note that every system used contains an odd number of lines. An odd number 
of lines in the picture means that there must be a whole number of lines plus one 
half-line in every field. Thus the 405-line system gives 202| lines per field; the 625- 
line gives 3121; and so on. As you will see in detail later on, the purpose of this is 
to make interlacing automatic. 

Whatever the number of lines a system uses, the rate at which the lines are produced, 
or presented, is termed the line scanning frequency (sometimes the horizontal scanning 
frequency). If one complete picture consists of 405 lines and takes one twenty-fifth 
of a second to scan, it means that lines are being produced or presented at a rate of 
(405 x 25 =) 10,125 per second. 

The time-base generators which govern both line and field frequencies are (as you 
will discover) basically oscillators, and oscillators work at so-many-cycles-per- 
second. It is therefore customary to express line scanning frequencies in "Hertz", 
the unit of measurement now used for cycles-per-second. Thus the line scanning 
frequency of a 625-line system is always said to be (625X25=) 15,625 Hz. 



1.36 [§3 

Line Scanning Periods 

The time taken by the electron beam to scan any one complete line of a field is 
called the line scan period (or sometimes the horizontal scan period). It is calculated 
as follows. 

If 405 lines take one twenty-fifth of a second to be produced or presented, one line 
1 
must take - seconds. This works out at 0-0000988 seconds, or 98"8 micro- 

405 x 25 

seconds (one ,us being, as you will remember, one-millionth of a second). Since it is 
in practice impossible to avoid small variations in the field frequency, the line scan 
period for the 405-line system is generally taken to be a round 100 microseconds. 

The line scan period for the 625-line system works out at 64 microseconds. Since 
there are 312-5 lines in a single field in this system, and since it takes 64 /is to scan a 
single line, it follows that a whole field takes 20"$o5 /us to scan. In the 405-line 
system, there are 202-5 lines in a field, and each is scanned in about 100 ^.s. Once 
again, the entire field is scanned in approximately the same time of 20,000 ( «s. 

In neither system, therefore, is the horizontal or line scan rate less than some 
200 times greater than the vertical or field scan rate. In the 625-line system, indeed, 
it is over 300 times greater. Contrast this with the time it takes your eye to scan a 
line of this page, and then the page itself. 

Note that the 20,000 /.is which, in both systems, it takes to scan a field is equal to 
sVh of a second. This ties in with what you already know — namely, that 50 fields 
are presented in every second. 

Aspect Ratio 

The ratio of picture width to picture height on a TV screen is known as the aspect 
ratio, and is standardized in most systems throughout the world at 4 : 3. 

Many years ago, when suitable dimensions for a TV picture were being discussed, 
it was decided to make the aspect ratio the same as that of the then conventional 
cinema film — with the idea that the latter could be televised with the minimum of 
adjustment to either picture tube or camera. It has remained the same ever since. 

The true aspect ratio of a televised picture is determined in the camera. So although 
you can adjust the Height and Width controls on your TV receiver to give almost 
any aspect ratio you like, the picture will not appear natural if the ratio is other 
than 4 : 3. 

ASPECT 



• 



cess 

than norma/ 



GREATER 
than norma/ 



§4: THE PICTURE SIGNAL 



1.37 




The most important piece of equipment 
in the television studio is the TV camera; 
for it is there that the picture signal 
originates. This remarkable electronic 
eye, silent and invisible to the viewer, is 
continually watching every movement, 
every change of shape and every tonal 
content of the scene, and converting what 
it sees into a complex stream of electrical 
signals so that the scene may be repro- 
duced, almost instantaneously, many miles away in millions of homes. 

To perform this feat, the TV camera makes use of a special type of cathode-ray 
tube called a camera tube, which transforms the image to be televised into an equivalent 
picture composed of millions of tiny individual electric charges. This charge-image 
pattern is then scanned from top to bottom in a series of horizontal lines by a narrow 
beam of electrons which "reads" the electrical information contained in the pattern 
and converts it into a consecutive series of electrical signals, each proportional in 
amplitude to the brightness of a particular section of the original image. 

These picture signals, representing the tonal content of the image, occur one after 
another rather like the information presented on a ticker-tape machine. After 
amplification (and mixing, for reasons which you will shortly see, with synchronizing 
pulses), the picture signals, together with the separate sound signals, are carried by 
land line or by SHF microwave link to the transmitting station, where they are 
radiated as radio-frequency waves. 

Electron 
Gun 



Electron 

Scanning 

Beam 



Horizontal 

and Vertical 

Deflection Coil 

Assembly 



Viewfinder 



Light-sensitive 
Surface 




To Main Amplifiers 

and Camera Control 

Units 



How 
P/CTURB StCHALS 

are Produced 
in the TV Camera 

Television cameras, like photographic cameras, contain an optical lens system 
which collects the light from the scene and focuses it on to a light-sensitive surface. 

In the photographic camera, this surface is usually a section of a spool of film. 
In the TV camera, it is a specially coated surface inside the camera tube which is 
known as a photo-cathode, or (in some types of camera tube) as a mosaic. This 
photo-cathode itself forms part of a target assembly. 



1.38 



[§« 



Secondary Emission 

Before studying the behaviour of the light-sensitive target in the television camera, 
you must first understand something about a phenomenon known as secondary 
emission. This phenomenon is similar to the photo-electric effect described in an 
earlier Section in that it concerns the emission of electrons from a surface. It differs 
from photo-electricity in that it relies, not on light to cause the emission of electrons 
from the surface, but on the impact on that surface of a stream of electrons from an 
outside source. 

When the surface, usually of a conducting material, is bombarded by electrons 
exceeding a certain velocity, the impacts cause additional electrons to be released 
from the surface. These are known as secondary electrons. The number of secondary 
electrons released per impacting "primary" electron is dependent on the velocity of 
the primaries, and on the composition of the material being bombarded. 

The number of secondaries released for every primary electron impact is known as 
the secondary emission ratio of the material. It is typically between 2 and 10. One 
of the most used secondary-emissive materials is caesiated silver (i.e. silver oxide 
coated with caesium). This material, in addition to having very efficient photo- 
electric performance, also has the comparatively high secondary emission ratio of 7. 

The diagram below shows what happens when a single primary electron is attracted 
towards the surface of a secondary-emissive material. Provided that the positive 
attracting voltage is high enough, the electron will acquire sufficient velocity, and 
therefore sufficient kinetic energy, to knock secondary electrons from the surface of 
the material. 

In the diagram, seven secondaries are shown as being released by the impact of 
the single primary electron. The secondary emission ratio of the material is 
therefore 7. 



SECONDARY MISSION 

Secondary ^^^^^ 
Electrons ^\ V^^^^^ 




Primary 
Electron" 



+ Attracting Voltage 



§4} 1.39 

The Photo-Multiplier 

An important electronic device which relies on secondary emission for its operation 
is the electron multiplier. This is a current-amplifying device which is capable of 
enormous amplification — often greater than one million times. 

The principle on which it works is as follows. Primary electrons, originating from 
some thermionic or photo-electric source, are attracted towards a positively polarized 
secondary-emissive surface, from which secondary electrons are released when they 
strike it. These secondaries are then attracted towards a second positively-polarized 
secondary-emissive surface, which is held at a potential higher than that of the first. 
Thus if seven secondaries are released from the first surface, 7 x 7 = 49 secondaries 
will be released from the second surface — and so on for as many additional surfaces 
as you care to employ. 

The photo-emissive surfaces in an electron multiplier are called dynodes. In a 
5-stage electron multiplier in which each of the five dynodes has a secondary-emission 
ratio of 7, the overall gain of the device will be theoretically Ixlxlxlxl — 16,800. 
In practice, however, the overall gain will be appreciably less, for perfect collection 
and emission is never realized in practice. 




£och Dynode (Dl, D2, etc.) 

Primary/' \ \ 44 JJ ^\V^^ IlllllfJfff has a Secondary Emission 

Electron JtJ( II If V^V^VCvJfl////// Ratio of 2. So the number 

of Secondaries arriving at 
05 = 2x2x2x2= 16 



How the ELECTRON MULTIPLIER works 



Electron multiplier assemblies are often used in conjunction with photo-electric 
surfaces to provide large amplification of the tiny currents which are produced by 
photo-electrons released from the light-sensitive surface. 

In such an arrangement, a transparent light-sensitive surface such as a photo- 
cathode is placed at one end of an evacuated glass tube, with the electron multiplier 
immediately behind it. When the photo-cathode is exposed to light, it releases from 
its rear surface electrons which are then directed towards the first stage of the electron 
multiplier. 

Such an assembly (photo-cathode plus electron multiplier) is called a photo-multi- 
plier. A much-used TV camera tube called the Image Orthicon works, as you will 
see, on a principle very similar to that of the photo-multiplier. 



1.40 [§4 

The Television Camera 

Since the beginning of electronic scanning, many types of camera tube have been 
designed, some better suited than others for a particular application such as studio 
use, outside broadcasting, etc. ; but of them all the most widely used over the years 
have been the Iconoscope, the Image Orthicon and the Vidicon. 

The Iconoscope 

The Iconoscope camera tube, invented in 1925 by the Russian-American scientist 
Vladimir Zworykin, though now obsolete, has played an important part in the 
history of television. A British development of the Iconoscope known as the Standard 
Emitron was in regular use by the BBC over the period 1936-1939, and for a short 
time after the end of World War II. A camera tube of this type was used to televise 
the Coronation procession of King George VI in 1937— the very successful first 
"live" outside TV broadcast by the BBC. 

Although it is now largely superseded by camera tubes of more recent design, the 
principle of operation of the Iconoscope makes a good starting-point for your study 
of the television camera. 




/ue St<z*td&tot £mi£no*t (Zamefia 



§4] 



1.41 



The Iconoscope — Construction 

The envelope of the iconoscope shown in the diagram below consists of a spherical 
glass bulb about 200 mm in diameter. To it is attached a side tube similar in appear- 
ance to the neck of a conventional cathode-ray tube. This houses an electron gun 
similar in general construction to the type described on page 5.101 of Basic Electronics, 
but rather more complex so as to minimize the danger of secondary emission within 
the gun itself. 

Round the inside of the neck of this side tube there is deposited a thin film of 
conducting material, electrically connected to earth through a terminal on the outside 
of the glass bulb. This is called the final anode. 

The light from the scene is collected and focused by an optical lens system on to the 
light-sensitive surface of a rectangular target suspended in the centre of the glass bulb. 
The surface of the bulb through which the light passes (the optical window) is made 
flat so as to prevent geometrical distortion of the light image on the target. 




» TH E ICONOSCOPE - Standard Emitron 




WSSMBSBl, 



sm 




1.42 



[§« 



The Iconoscope — Construction (continued) 

The target itself consists of a very thin sheet of mica covered on one side with 
millions of tiny globules of oxidized silver coated with a thin film of caesium. Since, 
as you know, caesiated silver possesses the property of giving off electrons when 
illuminated, every globule on the target forms a tiny light-sensitive island. Every 
island is electrically insulated from its neighbour by the surface of the mica sheet. 

This side of the target is called the mosaic. 

On the other side of the thin mica sheet is a metallic coating called the signal plate. 
By reason of the insulating properties of the mica, there exists a small capacitance 
between every silver globule on the mosaic on one side of the target, and the signal 
plate on the other. The target thus becomes an assembly of millions of tiny capaci- 
tances set very close together, each sharing the same dielectric (the mica) and each 
having one of its electrodes (the signal plate) in common with all the others. 

As soon as the beam from the electron gun starts to scan the mosaic, the silver 
globules acquire a small overall negative polarity with respect to the final anode. 
What happens (to cut a fairly involved action short) is that the beam knocks a great 
many electrons out of the photo-sensitive material of which the globules are composed, 
by a process of secondary emission. Many of these liberated electrons are collected 
by the final anode and flow to earth. The remainder fall back more or less evenly on 
to the surface of the mosaic, thus leaving it slightly negative with respect to the final 
anode. 

In this way, every one of the millions of silver globules on the mosaic becomes the 
photo-cathode of a tiny photocell whose common anode is the final anode itself. 




Glass 
Bulb 



Capacitance 
between Silver 
Globules and 
Signal Plate 



Silver 
Globules^,^ 
( Photo-cathodes) 



Mica Sheet 



Mosaic 



FRONT VIEW 



The 

iconoscope 

TARGET 




1/ 



Signal 
Plate 



To Signal Load 
Resistor 

► 



MAGNIFIED SECTION 
OF END VIEW 



§4] 



1.43 



The Iconoscope — Operation 

Light reflected or originating from the scene to be televised is collected by means 
of the lens system on the front of the camera, and focused on to the mosaic surface of 
the target. The millions of tiny photocells making up the mosaic are affected by the 
light from the scene. More electrons are emitted from their photo-cathodes, and are 
collected by the final anode. 

The actual number of electrons emitted from any cell depends on the amount of 
light falling on it. In brightly illuminated regions of the mosaic, cell emission will 
be quite large; in darker regions it will be considerably less. If the illumination of 
the mosaic were uniform, as it would be if the scene consisted of a plain white surface, 
the emission from every photocell would be identical. 

You know from your study of Basic Electricity that whenever electrons are released 
from a body, that body becomes less negatively charged. So the light from the scene 
falling on to the mosaic causes every photo-cathode to acquire a more positive charge 
proportional to the amount of light falling on it. 

In other words, the brighter the light, the larger the positive charge on that part of 
the mosaic on which the light falls. 



Photo-electrons 



Low-intensity 
Light Beam 



Positive 
Charges 




How Light from the Scene 
. . .Strikes the Mosaic 



1.44 



[§« 



The Iconoscope — Operation (continued) 

The "picture" of the scene appearing on the mosaic thus consists of a pattern of 
electrical charges (some large, some small) stored in millions of tiny capacitances. 
The longer the light image remains on the mosaic, the longer will the capacitances 
have to charge up. Ideally, each should be allowed sufficient time to charge to a 
voltage corresponding to the saturation of its associated photocell on the mosaic. 

The charges stored in the capacitances are prevented from leaking to one another 
(and so destroying the charge-image) by the very high lateral resistivity of the mica 
sheet. It has been known for a charge-image to remain stored on a mosaic for several 
hours without appreciable leakage occurring. 




Positive-charge Image 



How the POSITIVE CHARGE IMAGE is Created 

The next problem is to convert the capacitive charge-image stored on the target 
into a useful electrical signal. 

Again the electron beam is made to sweep across the mosaic from top to bottom, 
in a series of horizontal lines closely spaced one below the other. As it does so, it 
"reads off" the picture information stored on the target by converting the optical 
image focused there by the camera lens into a train of electrical signals, each repre- 
senting a very tiny part of the scene. 

This train of signals is the picture signal. You must now see how it is created from 
the charge image on the mosaic. 



§4] 



1.45 



The Iconoscope — Operation (continued) 

What happens is that as the beam sweeps across the mosaic, it provides a discharge 
path for all the tiny capacitances in turn as it touches their associated photo-cathodes. 
This discharge path is the beam itself, the cathode supply, the signal load resistor, 
and the signal plate of the target — this latter being, you will recall, a common electrode 
for all the tiny capacitances formed with the mosaic. 

As every photo-cathode in turn is touched by the beam, the positive charge created 
on it by light from the scene is neutralized by electrons from the beam. The associ- 
ated capacitance discharges through the signal plate and the signal load resistor. 
When the more positive areas created on the mosaic by brighter areas of the scene are 
touched by the beam, more electrons flow to neutralize the charge on the relevant 
capacitance, and vice versa. 

Signal Plate 



Mica Sheet 



Silver Globules 



DISCHARGING 
WB MOSAIC 



Cathode of 

Electron 

Gun 



IT 




Electron 

Scanning 

Beam 



Discharge Current 
Path of Capacitance 



l-M-Hf— 



Capacitance between 

Silver Globules and 

Signal Plate 



Signal Load 
'Resistor 



As the discharge currents flow through the signal load resistor, a sequence of voltage 
pulses is developed across it proportional to the amount of positive charge on the 
particular area of the charge image which is being scanned at that moment. In this 
way, the image of the scene focused by the camera lens on to the mosaic is converted 
into a series of electrical signals following each other in very rapid succession, each 
representing in ordered sequence a tiny part of the scene. 

The little capacitances through the target, once discharged, are immediately re- 
charged by light from the scene. There exists, therefore, an obvious danger of their 
being discharged once more by the beam as it passes over them during its fly-backs 
to the beginning of a new sweep or to the start of a new field. The result would of 
course be a meaningless jumble of signals reaching the output. 

To prevent this, there is applied to one of the electrodes of the camera tube at the 
end of every line and field scanning period a voltage of appropriate polarity called a 
blanking pulse. The amplitude of this pulse (it comes from a control unit in the studio) is 
such that it completely suppresses the beam during its fly-back periods. No unwanted 
signals are therefore touched off as the beam returns to the start of a new line or field. 



1.46 



[§« 



The Iconoscope — Sensitivity 

You will recall that in the mechanical scanning systems described earlier (the 
Nipkow disk, for example), the photocell was exposed to the light originating from 
each elemental area of the scene for only so long as it took the disk to scan that area. 
This meant that the useful light received was only a small fraction of the light available 
during the whole scanning period. The system was therefore very insensitive. 

In the Iconoscope, the photocells of the mosaic target are continuously exposed to 
the light from every elemental area for the whole of the scanning period; and charges 
accumulate for that period in the mosaic/signal-plate capacitances. The total charge 
accumulated is thus many times greater, and it is this which accounts for the enormous 
difference in sensitivity between the Iconoscope camera tube and the Nipkow disk. 

In theory, this difference in sensitivity is as much as 100,000 : 1 ; but in practice the 
sensitivity figure for most types of tube working on the charge-storage principle is 
considerably less than the theoretical maximum. In the Iconoscope tube the overall 
efficiency is, for various reasons, as low as 6%— though even so its sensitivity is still 
very much greater than that of any mechanical scanner. 

The Iconoscope — Resolution of Detail 

In the primitive 25-cell image-reproducing system described earlier in this book, you 
saw that the smallest detail of the scene capable of being resolved by the transmitter 
was the area of the scene "viewed" by a single photocell. In the Iconoscope camera 
tube, as you have seen, the mosaic consists of a very large number of tiny photocells, 
each of which represents the smallest detail of the scene capable of being resolved 
(i.e., an elemental area). You might think, therefore, that, if the diameter of the 
scanning beam was made no larger than the area of a representative photocell on the 
mosaic, the Iconoscope would be capable of an enormously high degree of resolution. 

In practice, however, difficulties of manufacture cause unavoidable differences in 
size to exist between the individual particles of silver— with the result that there is 
uneven sensitivity between photocells. If the scanning spot were to be made no 
larger than an individual photocell, therefore, every single cell ("large" and "small" 
alike) would be sampled by the beam, and the variations in sensitivity would cause 
the image to take on a "grainy" appearance. 

To prevent this happening, the structure of the mosaic— that is to say, the number 
of silver particles on every square inch of its surface— is deliberately made much finer 
than it need be. The result is that at least ten photocells are covered at any one time by 
the scanning spot as it moves across the mosaic. In this way, differences in sensitivity 
of the photocells are averaged out; and a uniform degree of sensitivity is achieved. 

The actual size of the scanning spot is determined by the size of the target and by 
the number of scanning lines in every complete picture. It is usually arranged for 
the diameter of the spot to be made equal to the spacing between adjacent lines. When 
this is done, the discharge of the mosaic is uniform and complete during every scan. 
Another advantage is that a spot of such dimensions renders the line structure of 
the picture less evident to the eye of the viewer. 

In the 405-line system, the effective area of the scanning spot on the mosaic is about 
000008 sq. ins., corresponding to a spot diameter of 001 inches, or about 0-25 mm. 
A unit of that size represents the effective elemental area of the iconoscope mosaic. 



§4] 



1.47 



The Iconoscope — The D.C. Level 

The average value of the picture signals produced by a TV camera tube is called 
the d.c. level. It depends on the average brightness of the scene being televised. For 
example, the average level of picture signals derived from a scene bathed in brilliant 
sunshine will be considerably greater than the average level derived from the same 
scene when it is illuminated by moonlight. 

As the illumination of a scene is increased, therefore, so also should be the d.c. 
level of the picture signals produced by the camera. 

In the Iconoscope, however, it is not possible for a direct current to flow between 
the mosaic and signal plate of the target because the only coupling between these 
two surfaces is their mutual capacitance, and d.c. cannot flow across a capacitor. 
Consequently, the mean level of the output signals from the Iconoscope does not 
follow changes in average illumination. It is therefore necessary to introduce into 
the output signals before they are transmitted an artificial d.c. level as nearly as 
possible representative of the mean brightness of the scene. 

But you know from Basic Electronics that before you can restore a waveform to any 
given level you need an accurate reference voltage. In all TV cameras, this reference 
voltage is taken to be the voltage level representing black, i.e. the voltage level of the 
signal when a totally dark area of the scene is scanned by the beam. Unfortunately, 
as you will read on the next page, there are special difficulties in obtaining a good 
black-level reference voltage in the Iconoscope. 

Meanwhile, note in the illustration below how the d.c. level of the signal alters with 
the average level of illumination of the scene. In both waveforms, maximum signal 
amplitude occurs when the beam is scanning the whitest areas of the scene; minimum 
signal amplitude when it is scanning the blackest areas. Signal amplitudes lying 
between the white and black levels represent varying shades of grey. 

Picture Signal Waveforms Produced 
by Sean a Scene With 



High 

Illumination Level 



White 
Level 



White 
Level' 



Illumination Level 



Black 
Level 




Time 



Black . 
Level"*" 



Time- 



Though the black level of the scene is shown above as zero signal voltage, it is not 
necessary that black should be represented by that particular level of voltage. Either 
a positive or a negative level of voltage could be chosen instead. 



1.48 



[§« 



The Iconoscope — The Black-level Reference Signal 

It is a fundamental disadvantage of the Iconoscope that the output signal corre- 
sponding to black cannot be held steady. The reason for this is an effect known as 
shading, which you must now understand. 

When the beam scans the mosaic of the Iconoscope target, the high velocity of the 
electrons in it causes a large number of secondary electrons to be struck out of all the 
silver/caesium globules it touches. Many of these secondary electrons are collected 
by the final anode and flow to earth, but the majority fall back on to the mosaic- 
principally, but unfortunately not only, on to the photo-cathodes they have just left. 

In the top left-hand corner of the mosaic, the secondary electrons have nowhere to 
fall back save on to the globules they have left. Their return partially cancels the 
positive charge which their departure caused. As the beam scans further to the right, 
however, and more particularly as it scans succeeding lines, the secondary electrons 
it liberates are not only attracted back to the globules they have just left, but also 
to the still-positive neighbouring globules which have already been scanned. 

When the beam nears the right-hand edge of the mosaic and also as it scans the 
bottom lines, the secondary electrons it liberates behave as before, a number of them 
wandering off to partially-positive neighbours and so further cancelling their charge. 
But for the globules on the right-hand and bottom edges of the mosaic, there is no 
source from which they can attract their share of these wandering electrons— with 
the result that these areas have a permanent tendency to be relatively more positive 
than the rest of the mosaic. 

You already know that the scanning of more positive areas of the mosaic causes 
larger discharge currents to flow through the signal load resistor, and so for larger 
voltages to be developed across it. These larger signal amplitudes correspond to 
the scanning of brighter areas of the scene. 

The result is that signals generated by the scanning of the right-hand and bottom 
edges of the Iconoscope mosaic tend to represent these areas as being brighter than 
they really are. They show up on the picture screen of the receiver as a kind of 
whitish flare along its right-hand and bottom edges, and are known as shading signals. 




The effect of these shading signals is that the level of voltage representing black 
varies with the position of the scanning spot on the mosaic, being accurate only 
when the beam is in the top left-hand corner and least accurate when it is in the bottom 
right. In practice, the best that could be done was to introduce a genuinely black 
object into every scene, and to obtain an arbitrary black-level reference voltage from 
that object itself. But it was seldom very satisfactory. 

You will now see how the problem was solved in cameras of a later type. 



§4] 



1.49 



The Image Orthicon 

TV cameras of the Iconoscope type which you have just been studying were a 
great advance on any of the mechanical scanning systems which had preceded them. 
They were very stable electrically; and, when skilfully used in good lighting conditions, 
they were capable of producing pictures of reasonably good quality. 

But they had their disadvantages also. Although their sensitivity was far in 
advance of any mechanical scanning system, it was still not as great as could be 
desired, and studio scenes still needed pretty strong lighting before they could register 
well on the viewer's screen. Power consumption was therefore high, and the dis- 
comfort of the actors considerable. 

Iconoscope-type cameras had other inherent disadvantages as well. They tended 
to be large, heavy and of awkward shape. They needed fairly complex correction 
circuits to counteract the effects of shading signals. And they lacked, as you have 
seen, the major virtue of having a reliable black-reference level. 

Of the range of cameras which have been designed to overcome the defects of the 
Iconoscope, two in particular are of importance — the Image Orthicon and the Vidicon. 

The Image Orthicon has been by far the most widely used successor of the Icono- 
scope. Its tube has a sensitivity more than a thousand times greater than has the 
tube of its predecessor (some Image Orthicon tubes are even more sensitive than the 
human eye!). It is suitable for work under almost any lighting conditions, and it is 
extensively used in TV systems all over the world. It is therefore important that 
you should understand the principles on which it works, and the limitations to which 
it is subject, before you go on to learn about the Vidicon, which represents the latest 
development in present-day design. 

Here then, to begin with, is a sketch of what a typical TV camera of the Image 
Orthicon type looks like. 

A Typical IMAGE ORTHICON TV Camera 




1.50 



[§« 



The Image Orthicon — The Camera Tube 

The Image Orthicon camera tube is cylindrical in shape. It has at one end an 
extension of larger diameter which contains a light-sensitive surface. Its physical 
appearance is represented in the illustration below. 

Early types of Image Orthicon tube were 76 mm in diameter; but present-day 
tubes (as used by the BBC since 1954) are of 115 mm diameter. These 115 mm 
types are commonly about 480 mm in length. 

He IMAGB ORWICOH Camera Tube 





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The Image Orthicon camera tube works on the following principles. 

Light from the scene to be televised is focused on to a transparent photo-cathode at 
one end of the tube, which emits electrons from points on its inner surface in numbers 
corresponding to the tonal composition of the scene. These electrons are propelled 
towards a small and exceedingly thin glass target, on which they produce a positive 
charge image of the scene by the action of secondary emission. 

Because of the extreme thinness of the target glass, the charge-image so produced 
on its photo-cathode side "leaks through" to its reverse side and is there reproduced 
unaltered. Glass has a very high resistance in all normal circumstances (see later, 
page 1 .54); but the target of the Image Orthicon is so thin that its "through" resistance 
is very low and it becomes in this direction conductive. 

A low-velocity electron beam is then arranged to scan the reverse side of the target, 
where it neutralizes the charge-image by successively giving up enough of its own 
electrons to cancel the varying positive charges it encounters. The beam needs to give 
up more of its electrons to cancel the high positive charges which represent the brighter 
parts of the scene than it does to cancel the lower charges representing the darker tones. 

All electrons not given up by the beam as it scans each individual area of charge are 
reflected back towards the electron gun. More electrons will plainly be reflected 
back from capacitances corresponding to the darker tones in the scene. 

Before these "surplus" electrons return to the gun which originally projected them, 
however, they are deflected towards an electron-multiplier assembly. This assembly, 
as you already know, is capable of producing an output signal which is exactly 
proportional to every variation in the density of the returning electrons. 

In this way, the tonal composition of the scene is exactly reproduced in the form of a 
consecutive stream of rapidly-varying voltage signals. 



§4] 1.51 

The Image Orthicon — Construction 

The electrode assembly within the Image Orthicon camera tube may be divided 
into three main sections: 

(a) the Image section. 

(b) the Scanning section. 

(c) the Electron Gun/Electron Multiplier section. 

The Image section contains the photo-cathode, the target and all the other electrodes 
used in creating the charge image. 

The Scanning section contains an accelerating electrode for the electron beam in 
the form of a graphite coating deposited on the inside wall of the glass envelope, with 
the horizontal and vertical beam scanning coils mounted outside. 

The third section contains a conventional electron gun to produce the scanning 
beam, with a 5-stage electron multiplier assembled round the perimeter of the gun. 

Around the outside of the tube, covering both image and scanning sections, is 
placed & focusing coil whose function is to produce an axial magnetic field within the 
tube. 

All these components of the tube can be identified in the illustration below, to- 
gether with one or two others whose purpose you will shortly discover. 



• Image Accelerator Grid 



Optical / 
Window 



The Image Orthicon 
Camera Tube 



Dynodes 




Scanning Section 



| Electron \ 

Gun/multiplier 

I Section I 



1.52 



[§* 



The Image Orthicon — The Photo-cathode 

You should now consider the various component parts of the Image Orthicon 
camera tube in turn, beginning logically with the photo-cathode. 

Light from the scene is collected by the lens and focused on to the glass end-window 
of the tube. This has been coated on its inside surface with a special mixture of 
antimony, silver and caesium which forms a semi-transparent photo-electric surface. 
This coated end-window is the photo-cathode. It is normally connected to a large 
negative voltage (typically, minus 400 V). 

As you already know, the number of electrons released from the photo-cathode by 
the impact of light on its surface will vary from point to point according to the tonal 
composition of the scene — bright areas giving rise to large emission and darker areas 
to much less. 

When they leave the photo-cathode, the electrons are directed towards the target 
by the potential difference existing between the large negative voltage on the photo- 
cathode, on the one hand, and (as you will learn on the next page) the less negative 
voltage on the image accelerator grid and the positive voltage on the target mesh, 
on the other. 

A direct current is made to flow in the focus coil, producing an axial magnetic 
field which ensures that the electrons travelling towards the target do so in straight 
paths parallel to the axis of the tube — the photo-electrons being made to follow the 
lines of force produced by the current in the coil. 

Image Accelerator Grid 



Light 
from Scene 

/ 


Photo-cathode 
/ / 




Photo-electrons 

* // Target Mesh 


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The Image Orthicon CameraTube- IMAGE SECTION 



4] 



1.53 



The Image Orthicon — Creation of the Charge Image 

When they arrive at the target, the photo-electrons strike it with sufficient force to 
cause the emission of secondary electrons. Every point on the target from which 
secondary electrons have been "struck" in this way instantly acquires a positive charge 
equal in magnitude to the total negative charge removed by the loss of electrons 
emitted from that point. 

The result is that a positive charge-image is formed on the target, every small charge 
representing an elemental area of the scene. Bright areas of the scene are repre- 
sented by dense areas of large positive charges on the target, while darker regions are 
represented by lower concentrations of comparatively weaker positive charges. 

But what, meanwhile, has happened to the secondary electrons struck from the 
target b^ the varying light signals coming from the lens system and the scene? Left 
to themselves, they would quickly fall back on to the target from which they came, and 
in doing so obliterate the positive charge-image on it before it could be put to any use. 



To collect these electrons, there is placed in front of the target, but very close to it 
(about - 001 inch away), a closely woven mesh of very fine wires to which a small 
positive voltage is applied. The wires are so fine that they do little to impede the 
passage of the fast-moving photo-electrons on their way to the target; but the 
secondary electrons emitted from the target are travelling very much slower, and so 
are attracted to the mesh and dissipated in the supply circuit connected to it before 
they can do any harm by falling back on to the target. 



(+)l.5v 



Target 



I 



(End View) 



(Front View) 



Photo-electrons 



\ 



Target Mesh 








Secondary 
Electrons 



THE POSITIVE 
CHARGE IMAGE 



1.54 



[§4 



The Image Orthicon — The Target 

The target in the Image Orthicon consists of a rectangular shaped sheet of glass 
approximately 50 mm wide and 38 mm high. The glass is extremely thin, typically 
five thousands of a millimetre, or 0-005 mm — which is thinner than the finest of 
cigarette papers. The reason for this extreme thinness is, as you know, to keep the 
effective electrical resistance measured between the surfaces of the target (i.e., 
through the glass) so low that the positive charges "impressed" by light from the 
scene on to its photo-cathode side can leak quickly through on to its reverse (scan- 
ning) side. 

A piece of glass as thin as this needs to be kept small in area lest it vibrate enough 
to distort the collection and scanning of the charge image. Hence the small size of 
the Image Orthicon target. 

But why, you may ask, use glass at all? Why go to the trouble and expense of 
making it so exceedingly thin when some other less resistive material could be used 
which would leak the positive charges through from one of its surfaces to the other 
just as well, and with far less bother? 

The reason is that the target must have a lateral resistance which is very high indeed, 
in order to prevent individual charges from spreading into one another and so destroy- 
ing the whole pattern of the charge image. Glass is a material which possesses this 
high lateral resistance to the movement of electrons. 

Yet, even so, it is necessary to keep the spacing between adjacent picture elements 
on the target about 20 times the thickness of the glass target if smearing of the charge 
image is to be avoided. 

The diagrams below illustrate these points— but you must not forget, in studying 
them, that it has been necessary to show the thickness of the glass target as far greater, 
proportionally, than it really is, in order to show how the target works. 



Ohms 



Ohms 



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Charges on . . . 
THIS side leak through to . . . 
THIS side 



High Lateral Resistance 



Low 'Through' 
Resistance 



TAROBT *6StSTANC£ 




§4] 



1.55 



The Image Orthicon — The Scanning Beam 

The electron beam used to scan the target of the Image Orthicon is produced in an 
electron gun similar to that used in the Iconoscope. In the Iconoscope, however, 
the accelerating voltages of the tube were sufficiently great for the beam to strike the 
mosaic with considerable force during scanning. This resulted in secondary emission 
of electrons at the mosaic— which in turn gave rise to the undesirable shading signals. 

In the Image Orthicon, the electron beam is first accelerated by the positive potential 
of the wall anode in order to make sure that it is properly directed towards the target. 
But then, just before it reaches the target, it is deliberately slowed down by the 
repelling force of the decelerator grid. 

There is applied to this grid a carefully calculated voltage which is so much less 
positive than the potential of the wall anode that it appears to the approaching elec- 
trons to be negative. It slows them down so much that they can only just get through 
it; and they finally reach the target only because the relatively tiny attractive forces 
of the positive charge-image stored there pick them up and pull them "home". 
Repelling Force Attracting Force 



Electrons 



Electron 
Gun 




Target 



Decelerator 
Grid 



Wall Anode 



m i*A0i new Sccum &4&m 

When they do reach the target, the electrons act just as they did in the Iconoscope — 
neutralizing the total positive charge at the point where they hit, and giving up some 
of their number to do so. The higher the positive charge at any one point on the 
target, the more electrons have to be given up to cancel it out. There are therefore 
fewer electrons left over where the beam scans areas of high positive charge caused 
on the target by the brighter areas of the scene. 

Whatever their numbers at any point, the "survivor electrons" are at this instant 
of time (the cancellation of the charge image) virtually at rest, the attractive and repul- 
sive forces acting on them being momentarily in balance. Almost immediately, 
however, the still-positive potential of the decelerator grid begins to pull them back 
towards the electron gun. Very shortly thereafter they come under the still greater 
attractive force of the much more positive wall anode, which causes an enormous 
increase in their velocity. 

The returning electrons, of course, vary in density along the length of the beam 
according to the tonal content of the scene. 



1.56 



[§4 



The Image Orthicon — The Electron Multiplier 

When they get near the electron gun, the returning electrons are diverted on to the 
first dynode of the electron multiplier by an electrode aptly named the persuader grid. 
They strike the surface of the dynode, and secondary electrons are emitted — the 
actual number varying from moment to moment according to the density of the 
electron beam. 

These secondary electrons are attracted by the higher potential of the second 
dynode, from which additional secondaries are struck — the multiplication process 
continuing for a further three stages (making five in all), after which the now-much- 
more-numerous electrons are collected by the anode of the electron multiplier 
assembly. The signal appearing at this anode is a highly magnified version of the 
variation in current flow occurring in the electron beam returned from the target. 

This signal current is then made to flow through a load resistor connected in series 
with the anode; and the variations in voltage developed across this resistor by reason 
of the constantly varying current passing through it constitute the picture signal 
output of the camera tube. 

Despite all the complicated stages through which it has passed, this output signal 
carries a faithful representation of the tonal composition of the scene. 



Return Beam 



Persuader 
Grid 






Coupling Capacitor 




Scanning 
Beam 



You will see from the illustration that the output signal developed across the anode 
load resistor of the electron multiplier is connected to the signal amplifier by means 
of a capacitor. This means that no d.c. can pass from the tube to the amplifier, and 
that the output signal can at this stage be given no d.c. level (representing, as you 
know, the mean brightness of the picture). 

In the Image Orthicon, however, the target mesh prevents secondary electrons 
knocked out of the target face by the arriving photo-electrons from falling back on 
to it during periods of darkness. No charge-image is therefore produced on the 
target during these periods, and there is in consequence a very accurate black-level 
reference signal. That being so, the d.c. level can be restored to the output signal 
at any point in subsequent circuits, using the d.c. restoration technique you learnt 
about in Basic Electronic Circuits, Part 1 . 



§4] 



1.57 



The Image Orthicon — Blanking 

The accurate black-level reference signal mentioned on the last page is obtained by 
simulating the conditions of darkness at the target face for short periods at the end 
of every line and field scan. This is done by applying a large negative blanking poise 
to the target mesh at the appropriate moments, so completely repelling for the period 
of the pulse all photo-electrons travelling towards the target. 

The electron beam during these periods of blanking therefore returns with none of 
its electrons missing, and so can be used to form an accurate black-level reference. 



STOP/ 
During the 

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Miscellaneous Points 

Q You have seen that, when there is no charge image on the target, the density 
of the return beam is constant, and almost equal to that of the scanning beam. 
There is then no signal output. When a charge image appears on the target, 
however, the density of the return beam is reduced and an output signal results. 

It follows that, since the output signal is derived from a reduction in beam current, 
the output signal must be negative-going. 

Q The lateral resistance of the glass in the target is very sensitive to changes in 
temperature; and unless precautions are taken to minimize such changes, the reso- 
lution of the tube will be impaired and smearing of the reproduced image will occur. 

Most image orthicons have their image sections thermostatically controlled by a 
small heater coil wound round the section. Typical operating temperatures are 
between 35° and 45° C. 

Q An effect known as image burn-in, or image sticking, is sometimes experienced 
on the Image Orthicon target, especially as the tube ages. When the target has been 
exposed to a stationary scene for a long period (e.g., during the transmission of a test 
card), an image of the scene can be retained on the target long after the scene has been 
removed. The effect can generally be cured by exposing the photo-cathode to a 
uniformly bright scene for a fairly long time. 



1.58 



[§4 



The Image Orthicon — The Viewfinder 

The viewfinder used in the Image Orthicon (and in all modern TV cameras requiring 
manual operation) is fully electronic, and does not actually look directly at the scene 
at all. It forms, in fact, a miniature self-contained TV receiver within the camera 
itself. Part of the output signal from the camera tube is fed to the viewfinder, which 
displays a reproduced image of the scene on a rectangular CRT generally some 
180 mm across. 

The advantages of this type of viewfinder are that no separate lenses are needed, and 
that the brilliance of the image being displayed can be made as bright as desired. This 
latter facility is of especial value to the operator when a dimly-lit scene is being televised. 




The Lens Turret 

On the front of all modern TV cameras designed for general-purpose use there is 
an impressive array of lenses and lens hoods, all mounted on a rotatable turret. 
This turret can be controlled manually or, in some types, automatically from a remote 
position. 

Every lens is designed for a particular function — "close-up", "long-shot" or 
"zoom". A "zoom" lens (or, to give it its proper name, a zoomar) enables a gradual 
close-up of a distant point to be obtained without interrupting the scene, the camera 
being made to appear to move towards the scene. This facility is much used in the tele- 
vising of personal interviews, and in the stage appearances of well-known personalities. 

You will see in Section 6 how a zoomar works. 



§4] 



1.59 



The Vidicon 

The Vidicon represents one of the most recent developments in television camera 
design. 

Its camera tube is only about 150 mm long by 25 mm in diameter — which is very 
much smaller than the tubes used in either the Iconoscope or the Image Orthicon 
cameras. This small size, combined with its correspondingly light weight, makes the 
Vidicon very suitable for use in industry (monitoring, for instance, the operation of 
a process inaccessible to, or dangerous for, human beings) or in field work such as 
outside "telecasts" by the BBC or 1TV, in which ready portability is an advantage. 
The Vidicon tube is also useful in colour TV — for which cameras need to operate 
three tubes simultaneously to reproduce the three primary colours — red, blue and 
green — in the scene- ,.....;.:.:.»:.>:.... 




As against its advantages, however, the Vidicon lacks the extraordinary sensi- 
tivity of the Image Orthicon. Early models, too, responded slowly to moving 
objects in the scene, the reproduction of which on the screen was apt to be marred 
by "smearing". Great progress has been made towards overcoming this fault; but 
it remains true that the Vidicon is particularly suitable for the recording of fairly 
static scenes, in which it is frequently convenient to operate it by remote control. 

The Vidicon Camera Tube consists essentially of a cylindrical glass envelope 
containing an electron gun at one end and a light-sensitive surface at the other. 
There are few internal electrodes, and the complete tube generally weighs less than 
three ounces. 

Its essential difference from the Iconoscope and Image Orthicon tubes is that, 
whereas they employ the principles of photo-electricity and secondary emission 
respectively, the Vidicon produces a charge image on its target by making use of the 
photo-conductive effect. You will recall that in this effect the resistance — and thence 
the conductivity — of a photo-conductive material is affected when light falls on to it. 
Thus if the target of a camera be coated with a photo-conductive material, areas of 
high illumination in a scene focused on to this target will be made more conductive 
than areas on which less light falls. 

You should now see how this fact can be used to develop the chain of varying 
voltages which makes up the output signal of the camera tube. 



1 .60 [§ 4 

The Vidicon Camera Tube (continued) 

The envelope of the Vidicon camera tube is closed at one end by a flat, transparent 
optical window made of glass, which is coated on its inside surface with an equally 
transparent film of conductive material. This film (whose thickness has had to be 
greatly exaggerated in the illustrations which follow) forms the signal plate. It is 
electrically connected to a ring-type electrode encircling it, and projecting slightly 
round the outside of the tube. 

The inside surface of this signal plate — the surface facing the electron gun — is 
coated with a thin film of photoconductive material (usually antimony trisulphide) 
which forms the light-sensitive target on which the charge image is to be created. 
In darkness, both signal plate and target carry a small positive voltage of about 35 V. 



Wire 
Mesh 



Wall Anode 



Window 



Signal 
Plate 

Ring 
Electrode 




Focus 
Coil 



Scanning 
Coils 



Alignment 
Optical p^\\\\l\\\\\\K^V\\^\^3^ Coil 



Y//////////7/, 




Cathode ' 



V///////////A m 



k\\\\\\\\\\\\ \\\\S\\\\\\\\\\\\\M 



Photo-conductive 
Target 



The VIDICON Camera TUBE 



Situated some 2-5 mm inside the target, and stretched across one end of an open- 
ended tube known as the wall anode, is a fine-wire target mesh. Both mesh and 
anode carry a large positive voltage (200-300 V). One of the functions of the target 
mesh is to protect the target from bombardment by negative ions produced in the 
electron gun. These negative ions are electrically-balanced atoms which have 
captured an extra electron and have so acquired a negative charge. They are heavy, 
and would cause "burn spots" on the target if they were allowed to hit it. But, being 
slow moving, they are readily trapped by the high voltage on the mesh and dissipated 
in the supply. 

The target mesh has also another function which will be explained on the next page. 

The wall anode itself extends for most of the operational length of the tube. It 
forms the final electrode of the electron gun, and also provides a uniform accelerating 
field for the electron beam. 

The scanning coils, which bring about both horizontal and vertical deflection of the 
scanning beam, are arranged round the tube approximately midway along the wall 
anode. A further small coil, known from its function as the alignment coil, is also 
situated round the tube, close to the electron gun itself. 

Lastly, right round the whole of the outside of the tube, is a long solenoid known 
as the focus coil. The magnetic field produced by this coil "straightens up" the 
electron beam just before it reaches the wire mesh, and ensures that it scans the target 
squarely at right angles — in paths which are all parallel to the axis of the tube. 



4] 



1.61 



The Vidicon Camera Tube (continued) 

The second function of the target mesh is very important. Still with no light from 
the scene falling on to the target, the electron beam is set to scan it. Accelerated 
by the (-%) charges on the wall anode and the mesh, the electrons in the beam are 
moving fast when they reach the mesh. But when they pass through it, they suddenly 
find themselves facing a much less positive charge of only 35 V on the target. They 
are so much slowed down in consequence that they are able to do no more than 
just cancel the (— ) charge on the target and bring it down to zero. 

Since the mesh covers the whole of the target area, it is thus able to provide for 
the beam a uniform decelerating field between mesh and target, whatever part of the 
target the beam may be scanning. 

The situation in the tube is now this. The cathode of the electron gun, the electron 
beam and the target are at earth, with the beam forming a conducting path between 
them but with its velocity greatly reduced just before it reaches the target. The 
signal plate is at 35 V. With no light falling on it, the coating of photo-conductive 
material between signal-plate and target has a resistance so high that it can be regarded 
as an insulator. 

A strong electric field therefore exists between the 35 V on the signal plate and the 
earthed target, because the two are so close together. If anything were to happen to 
lower the insulation of the dielectric between them, current would at once flow from 
the target to the plate. 



the vidicon SIGNAL PLATE 
and TARGET 

(much magnified) 

Cathode 




Ring-type Anode 
Target 



Scanning Beam 



The target is now exposed to light from the scene — and the insulation of the 
dielectric is at once affected. You will see what happens on the next page. 



1.62 



[§« 



The Vidicon Camera Tube (continued) 

When light from the scene to be televised is focused on the optical window of the 
tube, it passes through the transparent signal plate and falls on to the side of the 
target closest to the scene. The efTect of this light is to lower the resistance of the 
photo-conductive coating on the signal-plate side of the target by amounts corre- 
sponding to the tonal composition of the scene — bright areas bringing about large 
changes in resistance, and darker regions having little effect. 

At every point on the target where the resistance has been lowered, a current 
flows from the earthed side of the target to the signal plate. These currents build up 
slowly, and are not in themselves of operational significance. They do, however, 
take electrons away from the earthed (scanning) side of the target, and in doing so 
cause the potentials on it to rise above earth — the amount of each such rise depending 
on the decrease of the resistance through the target at that point. 

In this way, a pattern of positive charges is formed on the scanning side of the target, 
the more positive areas of which correspond to the brighter areas of the scene being 
televised. 

These tiny charges — each of which forms a picture element — are prevented from 
spreading into one another by the very high lateral resistance of the antimony tri- 
sulphide (or other material forming the target film). 




Lens 




\ < + >, 35v T / get Low 20v Charge 



Resistance 



m 









Weaker 
Light-ray 



Signal 
Plate 



(+)20v 



(+)5v 



\ Medium 
Resistance 




5v Charge 




The charge image appearing on the target is then scanned by the low-velocity 
electron beam in the usual way; and electrons from the beam neutralize the positive 
charges they touch. Wherever this happens, the varying positive potentials of the 
charge image are brought sharply back to earth. 

Electrons not needed for neutralization are returned towards the cathode (electron 
gun), and play no part in the action; for in the Vidicon tube no use is made of the 
return beam. 



§4] 



1.63 



The Vidicon Camera Tube (continued) 

When electrons from the beam encounter the large, dense areas of charge on the 
scanning side of the target— those which represent the high-lights of the scene being 
televised— more of the electrons will be required for neutralization than will be the 
case elsewhere. Whatever their numbers, however, these neutralizing electrons 
flow in a swift current through the target, into the signal plate, through the ring-type 
electrode surrounding it and thence through the signal load resistor. 

The further end of the signal load resistor from the signal plate is connected to 

earth, through the low impedance of the 35 V supply. The complete current path 

is therefore as follows: cathode— scanning beam— areas of positive charge on the 

target— through the now-conductive parts of the target— signal plate— signal load 

resistor — 35 V supply — cathode. 

Signal _ Positive Charge 

„. Target 

Plate v . 6 , Area 




Cathode 



-•- - _ Sea; 



nnin : 



* Sea 



m 



E 



/ 



CURHEHT Path 



<+>!■(-> 



t 



35v 



Signal Load 
Resistor 

One of the drawbacks of the Vidicon tube is its slowness in responding to sudden 
changes in illumination. This is due, partly to the time it takes to neutralize the 
charge image on the target, more importantly to the time which the target needs to 
return to its fully-insulating state after the light which has reduced its resistivity has 
been removed. 

This phenomenon is known as photoconductive lag. Its value is typically "18% 
after one scanning period, for peak-white decay". This means that, Ath of a second 
after peak-white illumination has been removed from the target, picture signal output 
is still 18% of full amplitude. The effect, which will obviously be more pronounced 
at low levels of illumination, manifests itself in a tendency for rapidly-moving objects 
to become "smeared". 

The scanned area of the Vidicon target, and therefore the size of the charge image 
stored on it, is very small indeed — typically only some 13 mm by 9 mm. The 
diameter of the scanning spot is, as you know, about 0-25 mm. This means that it is 
difficult to avoid scanning overlap — with consequent impairment of resolution. Some 
Vidicon tubes can only take about 200 scanning lines without overlap; and a 405-line 
scan sometimes gives an overlap of as much as three lines. 



1.64 [§4 

The Picture Signal 

Whatever the type of camera tube used — Iconoscope, Image Orthicon or Vidicon — 
the overall shape of the output signal is identical. Since it does not matter which 
type of tube is taken as an example, therefore, let us look at the type of output signal 
which the Iconoscope produces. 

Look at the illustration of that rather pretty girl on the page opposite, and suppose 
that her face is the scene being televised. (Save when she is being photographed in 
the closest of "close-ups", of course, there will almost always be more in the scene 
than just bits of the girl's hair and face; but the principle will not be affected even if 
an over-simplified example be chosen to illustrate it.) 

Suppose, then, that this face is focused on to the Iconoscope target, and that a 
charge image of it is created on the mosaic. Suppose further (though this is, of 
course, impossible in practice) that this charge image is visible to the naked eye, and 
that it in fact appears on the mosaic exactly as does the girl's face in the illustration. 
Suppose, lastly, that the electron beam of the tube is scanning the even numbered 
lines across the mosaic, and that it has just reached the beginning of Line 200 at the 
moment shown. Consider the shape of output signal which this one scan of the 
beam, and the two or three scans immediately following it, will produce across the 
signal load resistor of the TV camera. 

As the beam moves from left to right across Line 200, it begins in a white area 
(represented by highly-charged regions of the charge image) and then moves into the 
very dark area of the girl's hair. The output signal consequently falls sharply from 
the comparatively high level which represents white to a much lower level only a little 
higher than that representing black. Then, leaving the girl's hair, the beam moves 
across her cheek, which appears in a black-and-white picture as varying shades of 
grey. The output signal therefore takes up a level lying between black and white, 
and remains there until the beam reaches the girl's nose. Note the two sharp dips 
in the output signal representing her nostrils. 

The signal continues to vary in this way until the beam reaches the end of Line 200. 
It is then suppressed by the blanking pulse and returned to the left-hand side of the 
mosaic. The next line in the scanning sequence is No. 202, which traverses the 
rather "uneventful" region of the girl's face between her nose and her mouth. Line 
204, however, has the job of scanning her lips— and it is perhaps not surprising that 
the corresponding output signal (you will find it in the signal train at the foot of the 
illustration) begins to jump about a good deal more excitedly than did its immediate 
predecessor! 

The most important point to note about the whole process is that, whereas the 
scanning lines across the mosaic lie one underneath the other in space, the output 
signals corresponding to them occur one after the other in time. Every signal in 
the resulting train occurs at a definite time after the end of its predecessor — and 
thereafter at a definite time after the beginning of the first line of the scanning field. 

This train of output signals is called the picture signal. It is taken, through an 
amplifying stage, to a control unit in the studio where (as you will see) special pulses 
are added to it to form the video signal. 

Remember that the polarity of the picture signal, which is positive-going in the Icono- 
scope scan pictured above and in the Vidicon, is negative-going in the Image Orthicon. 



§4] 



1.65 



Start of Scan 
(Line 200) 

\| 
White Level 



Black Level 



f?W tfie 



RE SIGNAL 



<& ftnaduced 




Blanking Period' 



White 



Black 



urn 



200 



IrvJ 



202 



|— 204— | 



206 



^-208- 



1.66 [§4 

REVIEW of the TV Camera 

All TV cameras operate by converting the tonal composition of a scene into equivalent 
charge images on a photo-sensitive target; and then reading these images off with an 
electron beam to form a train of electrical impulses whose amplitudes are proportional 
to variations in the brightness of those areas of the scene to which the impulses correspond. 
This train of electrical impulses is called the picture signal. 



The first effective TV camera to use the 
scanning process was the Iconoscope. It 
was in use for many years, and pictures of 
good quality could be obtained with it. 

But its inadequate sensitivity made it 
satisfactory only in strongly-lit studios or 
on sun-lit outdoor locations. Also, the 
presence of shading signals (caused by 
secondary electrons struck out of the mosaic 
by the scanning beam) not only called for 
special correction circuits and for expert 
operators to keep adjusting them during 
transmission, but also made it impossible to 
achieve a reliable black-level reference 
signal. 

For these reasons, the Iconoscope camera 
is now obsolescent, and the most commonly- 
used modern TV cameras are the Image 
Orthicon and the Vidicon. 




[e Accelerator Grid 



Tin Image Orthicon 
Camera Tube 



The Image Orthicon makes use of the photo-electric and secondary-emission effects 
to create the charge image on its target. Light from the scene is converted into a 
stream of photo-electrons whose 
density increases with the bright- 
ness of the scene. The stream im- 
pacts on the target and knocks 
electrons from its surface, to be 
caught by the target mesh. Posi- 
tive charges of varying size are 
created where these electrons 
have been knocked away, and leak 
through to the other side of the 
very thin target. 

There they are scanned by a low-velocity electron beam, which gives up enough 
electrons to neutralize every charge it encounters. The "survivor" electrons (now 
fewest where the light was strongest) are returned towards the electron gun and directed 
into an electron multiplier assembly, the amplified output from which constitutes the 
picture signal. 




§4] 



1.67 




REVIEW of the TV Camera (continued) 

The Vidicon is the smallest of all TV camera tubes, and is much used in closed-circuit 
systems and in colour television. Its charge image is created when the light from the 
scene is used to lower the electrical resistance through the target. Small currents, of 
no operational significance, flow from the earthed (scanning) side of the target to the 
positive signal plate, leaving positive charges of 
varying size behind them. When these charges 
are neutralized by the electrons in the scanning 
beam, the discharge currents flow through the 
target, signal plate and signal load resistor to 
form the picture-signal output of the tube. 

The overall shape of the picture signal is 
the same, whatever the type of camera used 
to produce it; but the polarity of the signal 
varies. Whereas the Image Orthicon 
camera produces an output which is nega- 
tive-going {i.e., the whiter areas of the scene 
are represented by more negative excursions 
of the output signal waveform), the Vidicon 
picture signal is positive-going — as is that 
of the Iconoscope pictured opposite. In the 
case illustrated, the lighter areas of the 
girl's face are represented by a (+) jump 
in the output signal waveform; the darker 
areas of her hair and nostrils by a signal not 
far above the black-level reference voltage of 
the picture. 

The important feature of all scanning systems is that, whereas the scanning lines 
across the target lie one underneath the other in space, the output signals derived from 
them occur one after the other in time. 

Another camera tube even more recently developed than the Vidicon is the Plumbkon. 
Although fully capable of producing a good black-and-white picture signal from a 
televised scene, this new tube seems likely to find its principal uses in colour TV. Since 
it is hardly, if at all, used in any operational British television system at the moment 
of writing, its characteristics have not been covered in this Series. 





1.68 



§5: THE VIDEO SIGNAL 



If the image seen by the viewer is to be a faithful reproduction of that sent out by 
the studio, it is essential that the scanning spot shall move across the picture tube in 
the receiver at the same speed and at the same time as the scanning spot moving across 
the target of the camera tube, and that it shall at all times occupy the same relative 
position in its scanning field. If any of these conditions are not realized, it will be 
impossible to keep the picture steady at the receiver; and it may either drift across the 
screen, dissolve into multiple images, or even break up altogether. 

To ensure accurate synchronization between transmitter and receiver, therefore, a 
series of synchronizing pulses are mixed with the picture signal in the studio, and are 
transmitted with it. At the receiver (as you will learn in detail in Part 2), these 
pulses are separated from the picture content of the received signal, and are used to 
synchronize the line and field timebase generator circuits feeding the picture tube. 

The sync pulses (as they are commonly called) take the form of rectangular-shaped 
pulses of voltage, of very accurately defined duration, introduced into the signal 
during the blanking periods which are needed at the end of every line scan and every 
field scan to allow the scanning beam to fly back to the beginning of the next line 
or field without being visibly traced out on the screen. 

The pulses are formed when signal amplitude is first reduced to blanking level, and 
then driven into the "blacker-than-black" region beyond the black-level reference 
voltage for the brief period of time needed for pulse formation. 

The composite signal containing the picture signal plus its associated sync pulses 
is known as the video signal. The illustration below shows a section of a typical 
video signal, consisting of two complete line scans and the three sync pulses associated 
with them. 




Time Occupied 
by one line 



Picture 
Signal Region 




H h- 



Sync Pulse 
Region 

(Blacker 

_ i 

than Black) 



THC VldCO SIGNAL -4 05 line system 



§5] 



1.69 



EFFECTIVE Period of the Line DURATION 



The Line Sync Pulse 

In all television systems, it takes many hundreds of lines to make up one complete 
field. A great many more line sync pulses than field sync pulses are therefore required. 
Since it is obviously essential that the comparatively rare field sync pulse shall be 
easily picked out and recognized by the circuit, its effective width is always made 
considerably greater than that of the line sync pulse— even though the amplitude of 
the two types of pulse is kept the same. 

Consider now the line sync pulses in more detail. 

As the scanning beams in the camera and picture tubes complete the scanning of 
every line, they are suddenly made to return to the left-hand sides of the target and 
screen respectively, so as to be back in the correct positions for beginning the scans of 
the next line. This sudden reversal of the scanning beams is (as you know) called the 
flyback or retrace; and the time taken for its completion is called the flyback (or 

retrace) time. 

To prevent the charge on the target from being destroyed during flyback of the 
camera tube beam, and to stop the flyback being visibly traced out on the picture tube 
screen, the two scanning beams are individually suppressed during the flyback periods 
by a negative blanking waveform applied to the two tubes. It is during the period of 
this blanking waveform that the synchronizing pulses are added to the picture signal. 

Flyback at the end of every line is 
initiated by the arrival of the line sync 
pulse. The time interval between any 
two successive line sync pulses therefore 
represents the full duration of a scanning 
line (including flyback). 

For reasons which you will see on 
the next page, the overall duration of 
the line blanking period is made about 
twice as long as the duration of the line 
sync pulse. This means that the effec- 
tive period of the line duration {i.e., the 
the period during which the picture 
signal is actually visible on the screen) 
is determined not by the time interval 
separating the line sync pulses them- 
selves, but by the time interval separat- 
ing the line blanking periods. 

The point is brought out in the 
illustration to the left, in which the 
spaces between the two pairs of dotted 
lines to the left and right of the picture 
tube are blacked out by the blanking 
period. In practice, of course, these 
two "blacked-out" spaces do not 
appear on the screen of the picture tube 
at all, for the length of the scan is always 
arranged to fill the whole width of the 
screen. The "blacked-out" areas exist, 
none the less, even if they are not seen. 




I- — These Portions of the — "l 

I Line Scan are Blacked Out 

by the Blanking Period 



1.70 



[§5 



The Line Sync Pulse {continued) 

The general shape of the video signal waveform during the presence of the line 
sync pulse is the same in all TV systems, the only difference lying in the relative values 
of the time intervals between different parts of the waveform. 

There are two reasons why the duration of the line blanking period is approxi- 
mately twice that of the line sync pulse. The first is that sufficient time must be 
allowed for the picture-signal content of the video waveform to fall to blanking 
level before the start of the line sync pulse. If this decay time were not allowed for, 
the shape of the leading edge of the sync pulse would be affected by the magnitude 
of the picture signal, and the precise time at which the timebase generator circuit was 
triggered would be ill defined. In other words, the time required for the sync pulse 
to reach full amplitude would depend on whether it happened to start from peak 
white, from black, or from some intermediate value. The result would be erratic 
triggering of the line timebase generator circuit, and consequent distortion of the 
reproduced image. 

In practice, the line sync pulse is introduced a few microseconds after the com- 
mencement of the line blanking period, this brief delay period being known as the 
front porch. You can see it in the illustration on the page opposite; but the reason 
why it is needed appears in the picture below, which shows what would happen if 
the front porch were not there. 



Why the FRONT PORCH is needed 



White Level 




Errors in Time 

.of Triggering v 

of Scanning Waveform 

The second reason for the long blanking period is to allow time for flyback of the 
camera and picture tube scanning beams at the end of every line to be properly 
completed. This is done by maintaining the picture signal at blanking level for a few 
microseconds after the end of every line sync pulse. This delay period is known as 
the back porch. The new scan starts from varying points along this back porch, as 
soon as flyback from the previous scan is complete. 

The general structure of the video signal during the period of a single line scan is 
shown, for both the 405-line and for the British 625-line systems, in the illustration 
occupying the whole of the page opposite. 



§5] M 

GENERAL STRUCTURE of the MEO SiGNAL 
DURING A SINGLE LINE SCAN 

WE 405-LINE SYSTEM 



Picture 
Signal 




— Peak White Level 



Black Level 
Blanking Level 



Sync Level 



Back Porch 



THE BRITISH 625-LINE SYSTEM 



Line Period 
'(64 ns) 



Picture 
Signal 



Sync Pulses 



Front Porch 
(1-55 -us) 




— Peak White Level 

— Blanking (and Black) Level 
Sync Level 



Back Porch 



1.72 



[§5 



The Field Sync Pulse 

When the scanning beams in the camera and picture tubes finish the scanning of a 
complete field (that is to say, when they have— in the 405-line system— scanned 
202|- lines), they are made to return very quickly indeed to the top of the target 
and screen so as to regain the correct positions for beginning the scan of the next field. 

This re-positioning of the beam at the end of every field is very similar to the 
operation of line flyback, except that the beam (as you will shortly see) does not 
travel in a straight line, and takes much longer to return to the top of the screen or 
target than it does to return to the start of a new line. The time taken to complete 
the re-positioning is called the field flyback (or field retrace) time. 

For the same reasons as in line flyback, the scanning beams of the camera and 
picture tubes are both suppressed during the period of field flyback by the application 
of a field blanking waveform applied to the two tubes. It is during the period of this 
field blanking waveform that the field sync pulse is applied. 

Exactly as with the line sync pulses, field flyback is initiated by the arrival of the 
field sync pulse. The time interval between any two successive field sync pulses 
therefore represents the full duration of a complete field. In both 405-line and the 
British 625-line systems, this interval is one-fiftieth of a second. 

You already know how important it is that the arrival of the field sync pulse shall 
be readily distinguished by the timebase generator circuits in the receiver from the 
many hundreds of line sync pulses which have gone before and which come after it. 
This recognition is ensured by making the duration of the field sync pulse more than 
forty times as long as the duration of the line sync pulse. Simple integration circuits 
preceding the timebase generator circuits can easily distinguish such differences in 
pulse duration, and can therefore recognize the field sync pulse as soon as it arrives. 
The separation of the line and field sync pulses from one another, and from the 
picture signal content of the video signal, is performed by what are called sync 
separation circuits in the TV receiver. These circuits then route the sync pulses to the 
appropriate timebase generator circuits. You will be learning about them in Part 2. 

The essential difference in the duration of line and field sync pulses is shown in the 
illustration below. 




'"9 |ls 
Line Sync Pulse 



approx. 400 ns- 
Field Sync Pulse 




9 us 
Line Sync Pulse 



| RBUTm DOMTtON of fine & field sync pulses \ 



§5] 



1.73 



The Field Sync Pulse {continued) 

You will naturally have supposed, from what you have read so far, that the field 
sync pulse is a single pulse of a few hundred microseconds' duration. In fact this is 
not so, and you must now see why. 

In neither of the British TV systems is the duration of a complete line scan more 
than 100 microseconds. During the presence of a field sync pulse of several hundred 
us duration, therefore, several of these line scans would be lost. The effective duration 
of the field sync pulse in the 405-line system, for instance, is about 400 microseconds, 
or the equivalent of four line-scan periods, each 100 ^s long. During this period the 
line timebase generator circuit would (unless something were done about it) be 
without synchronizing pulses at all, and would probably slip out of synchronism. 

This would not be serious in itself, for the scanning beam is blacked out during 
the period of the field sync pulse anyway. But when the field sync pulse is removed 
and the line sync pulses take over again, the line timebase would often take a few 
lines to recover its original synchronism ; and this would mean erratic triggering during 
the first few lines of every field, with consequent distortion of the picture on the 



screen. 



Steps must therefore be taken to ensure that line synchronization is not lost during 
the period of the field sync pulse. This is achieved by building up the field pulse itself 
out of a series of shorter pulses, recurring at twice line frequency, and each of some- 
what longer duration than the normal line sync pulse. The shorter pulses are called 
half-line pulses. Their function is as follows. 

You will remember that, in order to effect proper interlacing, the field flyback 
has to start half-way along the last line scan of every second field. One sync pulse 
must therefore be available at the end of a line for starting the odd-line fields, and 
another sync pulse half-way along a line for starting the even-line fields. 

In the field sync pulse as a whole, every alternate half-line pulse triggers the start 
of an odd-line field, and the remaining alternate half-line pulses trigger the start of 
the even-line fields. The moment of onset of every second half-line pulse coincides 
with the moment at which the normal line sync pulse would have fired had it not been 
suppressed during the field blanking period. 

This somewhat complicated arrangement is illustrated in the diagram below. 

How the Field Sync Pulse is Broken Up into HALF-LINE PULSES ^ 

Level 

Black 
i /Level 

Blanking 
' fr~ " Level 
Sync 
Level 




*\ Line y\ 
Sync Pulses I 



*\ Line A 
Sync Pulses 



Duration of Line Periods 



51 — *<-* 



"Duration of Field Sync Pulse- 



Effective Shape of 

Field Sync Pulse — 
AFTER INTEGRATION 



X 



r 



— Blanking Level 
Sync Level 



1.74 



[§5 



The Field Sync Pulse {continued) 

While the half-line pulses are doing their job of maintaining line synchronization 
during the period of the field sync pulse, an identical train of them is being applied to 
a simple integration circuit quite separate from the synchronizing circuit of the line 
timebase. The job of this integration circuit is to produce out of the train of half- 
line pulses a single wide pulse of equivalent length, which will serve for purposes of 
field synchronization. 

Field Flyback Path 

You may have been surprised to read, two pages back, that the distance which the 
scanning beam travelled during the period of field flyback was considerably 
greater than the distance travelled by the beam during the period of line flyback. 
The reason for this is that the scanning beam, on completion of a field, does not return 
to the top of the screen (or target) by a direct route (as was shown in the simplified 
diagrams in Section 3), but rather does so in a series of zig-zag lines. 

This is because the line timebase generator continues to work, and to be synchro- 
nized, during the full period of the field sync pulse, including its flyback time. The 
scanning beam is therefore deflected across the screen a number of times as it pro- 
gresses towards the top of the screen. 

The path traced out by the beam during its return to the top of the screen is 
normally prevented from appearing on the picture tube by the field blanking wave- 
form applied to the grid of the tube. 

It is, on some receivers, possible for you actually to see the path traced out by the 
scanning beam during field flyback simply by increasing the setting of the Brilliance 
control on your TV receiver. This has the effect of altering the point at which the 
beam of the picture tube is cut off— this point is normally adjusted to coincide with 
the black level of the video signal — until the flyback becomes visible. 

What you will actually see when you do this will vary from receiver to receiver, but 
it generally takes the form shown in the illustration below. 





ftilB 
FLYBACK 

wi^ooVXe 



§ 5] 1.75 

The Field Sync Pulse {continued) 

The short lines seen in the illustration on the last page are produced during the 
short blanking-level periods between the broad pulses of the field sync pulse; the 
longer lines during the longer periods between the narrow pulses of the post-sync 
field blanking period, which will be explained shortly. 

The lines will appear to be sloping at an angle much greater than that of the normal 
scanning lines, particularly towards the bottom of the screen. This is because the 
beam is being returned to the top of the screen at a rate much faster than that of its 
normal downward motion during the scanning period of the field. The variation 
in angle of slope of the flyback scans is caused by the non-linear shape of the deflection 
current produced in the timebase generator during flyback. 

Composition of the Field Sync Pulses 

The composition of the field sync pulses in the British 405- and 625-line systems are 
basically similar; but certain additions are made at the beginning and end of the 625- 
line pulse which you should know about. The field sync pulses of the two systems 
must therefore be described separately. 

The Field Sync Pulse in the 405-line System 

In this system the field sync pulse has an effective duration of four line periods 
(as you will see if you look at the illustration over the next page). It is made up 
of a cluster of eight half-line pulses, each of about 40 \±s duration, separated by seven 
return-to-blanking-level intervals each of 10 us duration. The full duration of the 
pulse is thus (8 x40)+(7 x 10) = 390 txs, which is virtually the equivalent of the four 
line periods mentioned above. 

The repetition rate of the 40 txs half-line pulses is (as you would expect from what 
you learnt two pages back) made twice that of the line sync pulses, so as to ensure 
synchronization of the line timebase circuit during alternate fields. 

The end of the field sync pulse is followed by a return to blanking level for a period 
approximately equal to the duration of ten lines, i.e., a total duration of 1,000 us. 
This period is needed so as to allow adequate time for completion of the field flyback. 
It is known as the post-sync field blanking period. 

It is, of course, just as important that synchronization of the line timebase circuit 
should be maintained during the post-sync field blanking period as it was during the 
period of the field sync pulse itself. Now, however, there are no complications of 
interlacing to worry about, for field flyback has already started by the time the post- 
sync field blanking period begins. The post-sync blanking period is therefore inter- 
rupted by line sync pulses at the normal repetition rate. 

You will have gathered from the above that the overall duration of the field blanking 
period is equal to about 1,400 |xs — a field sync pulse of about 400 fxs, plus a post-sync 
field blanking period of about 1,000 \xs. This is the equivalent of 14 line periods, 
and is a point of real practical importance. 

It means that, in every field of 202J lines, 14 lines are totally suppressed during the 
field blanking period. And since there are two fields in each complete picture, a total 
of 28 lines is lost in every picture. 

The effective number of lines in the 405-line system (that is to say, the number of 
lines which are available for presenting the actual picture information) is therefore 
not 405, but (405—28 = ) 377 instead. As you will see in the next Section, this 
fact has an important bearing on the size of the bandwidth required for transmission 
of the vision signal. 



1.76 [§5 

The Field Sync Pulse in the 405-line System (continued) 

The illustration opposite shows the detailed composition of the field blanking 
period in the 405-line system, for both the odd- and even-line field scans. For 
convenience and clarity of presentation, the lines are numbered in the order in which 
they are actually produced, i.e., 1 to 202J in the even-line field, and 202J to 405 in 
the odd-line field — although, as you well know, the even-line field really produces 
lines 2, 4, 6, 8, 10, etc., in the picture, and the odd-line field lines 1, 3, 5, 7, 9, etc. 

Note especially in the illustration that the field sync pulse occurs at the end of a line 
on completion of the even-line field ; but half-way along a line on completion of the 
odd-line field. 

S^K10SSSSSISliaiaHKIEIKIKIKII3lglEISEIElEliaSlEIISlEIKIEIE!KIElEliaEllEliaElKISlEllEliaKEIHISlElS!^KlKIKSKIK! 
The Picture-Sync Ratio 

Although it rather interrupts the sense, the rest of this page must be used for a 
brief discussion of sync pulse amplitude. Read it in conjunction with the illustration 
on page 1.107. 

The relationship of picture signal amplitude to sync pulse amplitude is termed the 
picture-sync ratio. The value of this ratio at the transmitter depends on the answer 
to the question: What is the highest picture-sync ratio which will give adequate 
synchronization under conditions of low signal strength at the receiver? Obviously, 
a very high picture-sync ratio in the transmitted signal will give a waveform with 
good picture-signal content, but it will be useless if synchronization cannot be main- 
tained; while a very low picture-sync ratio could give good synchronization but a 
picture signal so inadequate as to be no good to the viewer. 

In practice, the ratio chosen is generally the one which makes synchronization just 
possible when the picture signal is of just acceptable quality. 

The relative magnitudes of the picture signal and sync pulse contents of the video 
signal are usually expressed as percentages of the peak amplitude of the modulated r.f. 
carrier radiated from the transmitter. 

In the 405-line system, the sync pulses are always negative-going, and their maximum 
excursion is represented by almost zero carrier amplitude (0-3%). Blanking level 
is represented by 30% modulation of the carrier. Black level is determined by the 
contrast range of the camera tube, and usually lies between 37% and 39% modulation. 
(The gap between blanking level and black level is often called the pedestal.) 

Thus the final picture-sync ratio in the 405-line system is very nearly 2 : 1 — the 
amplitude of the sync pulse being from about 3% modulation to about 38%, and the 
amplitude of the picture signal from black level to peak white at 100% modulation. 

In the British 625-line system, the picture-sync ratio is almost the same as in the 
405-line system, but the modulation values of the carrier differ. The sync pulses 
are always positive-going; and an increase in picture signal amplitude brings about a 
decrease in carrier amplitude. 

The maximum excursion of the positive-going sync pulses is represented by 100% 
modulation of the carrier. Blanking level (which is also black level) is represented 
by 75% to 77% modulation. The sync pulses therefore occupy some 24% of the 
maximum carrier amplitude during modulation. 

The picture-signal content of the vision signal ranges from a black-level modulation 
of about 76% down to a peak-white level of 18-20% modulation. Carrier amplitude 
is never reduced to near zero, as it is in the 405-line system. 

3SEIISlSHElElEIEIKliaKIIEIEISEHEllSKlElEllSllSI3KIKlElKIKiHK10l3SEl 



§5] 



1.77 




UJ «> Z >L 



1.78 [§5 

The Field Sync Pulse in the British 625-Line System 

You learnt three pages back that certain additions are made at the beginning and end 
of the field sync pulse in the British 625-line system which are not present in the 405-line 
pulse. These additions are known as equalizing pulses. Their purpose is as follows. 

Perfect interlacing can only be achieved in any system when the scan of the last 
line of alternate fields is terminated exactly at its half-way point, and when the first 
line of the following field is started at the corresponding half-way point. The scan- 
ning lines of the odd- and even-line scans then fall evenly between one another, and 
the interlacing is correct. 

To achieve this, the field flyback for alternate fields must be initiated at precisely 
the half-way point of the last line, and the flyback periods for both fields must be 
identical. If flyback is started too soon or too late, or if its duration differs between 
the two fields, the spacing between lines of the fields will be uneven and the interlacing 
poor. This will make the line structure of the picture presented at the receiver more 
obvious, particularly on picture tubes of the larger sizes; and the effect will be more 
noticeable still if the moment at which flyback is initiated is liable to drift. 

In the even-line field of the 405-line system, flyback is started at the end of the last 
line, so that the time interval between the last line sync pulse and the start of the 
following field sync pulse is the duration of one line, or about 100 us. At the end of 
the odd-line field, however, the flyback is started half-way along the last line, so that 
the same time interval is only about 50 y.s. Similar differences exist at the end of the 
field sync pulse and the start of the line sync pulses in the post-sync field blanking 
period; and interlacing is thereby impaired. 

This defect of the 405-line system is obviated in all 625-line systems by the inclusion 
of a train of narrow "get-ready" pulses at the start and finish of every field sync 
pulse. These equalizing pulses have half the width of normal line sync pulses, and a 
repetition rate of twice line frequency, i.e., the same frequency as the half-line pulses 
which comprise the field sync pulse itself. You will learn how they do their job 
when you get on to the so-called separation circuits in the receiver, in Part 2. 

Apart from the equalizing pulses, the composition of the field blanking period in the 
British 625-line system differs little from that in the 405-line system. It is shown in 
detail in the diagram on the opposite page. 

The field sync pulse itself consists of a cluster of five half-line pulses, each of about 
27-5 us, separated by four return-to-blanking-level periods, each of 4-7 us. Effective 
duration of the sync pulse is therefore nearly 160 y.s, equivalent to 2 \ line periods. 
Repetition rate of the half-line pulses is twice that of the line sync pulses, as in the 
405-line system, and for the same reason. 

The sync pulse is preceded by a group of five equalizing pulses having a total dura- 
tion equal to 2\ line periods. The pulses occur at twice line frequency, and each has 
a duration of half that of a line sync pulse (2-3 fxs). A similar cluster of five equalizing 
pulses/bZ/oH'.s the field sync pulse ; and is succeeded in turn by a further blanking-level 
period of 800 us, interrupted by line sync pulses at line frequency. Overall duration 
of the field blanking period is thus about 1,300 (j.s, equal to the period of 20 lines. 

In the British 625-line system, therefore, 20 lines are lost in every field, making 40 in 
every picture. The effective number of scanning lines is thereby reduced from 625 to 585. 



§5] 



1.79 




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1.80 [§ 5 

REVIEW of the Video Signal 

The purpose of sync pulses is to ensure that the movement of the scanning beam in 
the picture tube is kept in perfect synchronism with the movement of the scanning beam 
in the camera tube, and that both beams are held at all times in the same relative position 
on their respective tubes. If this is achieved, the electrical impulses representing the 
transmitted scene will be delivered to the picture tube of the receiver in exactly the 
same order, and at exactly the same speed, as they were created in the first place. 

Two kinds of sync pulse exist — one for synchronizing the scanning lines, and the 
other for synchronizing the camera-tube/picture-tube fields. Both are rectangular 
pulses of voltage which extend into the blacker-than-black region of the video signal. 



Line Sync Pulses are single, narrow 
pulses added to the picture signal during 
the line blanking periods. They are used 
to initiate line flyback, and recur at the 
same frequency as the scanning lines. 
The duration of a line sync pulse is about 
10% of the line scanning period in the 
405-line system, and about 1\% of the 
corresponding period in the British 625- 
line system. 




Field Sync Pulses consist of broader (half-line) pulses which are added to the picture 



signal during the field blanking period, 
these pulses produce a single broad pulse 
which is used to initiate field flyback. 

The effective duration of the field sync 
pulse is deliberately made much greater 
than that of a line sync pulse so as to 
make it easily identifiable by the sync 
pulse separation circuits in the receiver. 



When passed through an integrating circuit, 



How tin FMd Sync PuIm Is Broken Up Into HALF-LINE PULSES 



Peak 
White. 
""Level 




V 

Duration of Line Periods 



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AFTER INTEGRATION 



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Equalizing Pulses are used in all 625-line systems to ensure that interlacing is not 
impaired by differences in the time at which the field sync pulse arrives at the starting- 
points of alternate fields. They take the form of groups of very narrow pulses, recurring 
at twice line-repetition frequency, inserted in the field blanking period immediately before 
and after the field sync pulse itself. 

All Sync Pulses are applied to the picture signal when it is passed through a piece of 
apparatus known as the Camera Control Unit situated in the television studios. You 
will see how this unit works in the next Section. 



The output of the control unit is the video signal complete. 



§6: THE TELEVISION STUDIO 



1.81 



It may come as a surprise to you to know that the televison studio and the television 
transmitter are two entirely separate entities, often situated many miles apart. For 
example, the BBC's studios at their big Television Centre near the White City in 
London are about 10 miles away from the nearest transmitter at the Crystal Palace, 
and over 400 miles from the Kirk-o'Shotts transmitter in Scotland. 



The TRANSMITTINGfitMiNi 




The STUDIO 



Yet both these transmitters, and all the other BBC transmitters in the United 
Kingdom as well, are for a large percentage of their broadcasting time supplied with 
signals originating in the London studios. 

There are also, of course, a great many smaller studios and outside broadcast units 
situated throughout the country, but even the programmes from these are routed 
through a central control point before being relayed to the national Transmitter 
Network. 

The object of this procedure is to enable programmes originating from different 
sources to be accurately mixed and phased with one another before being 
transmitted. A good example of the need for this accurate mixing is a sports-news 
programme, which will often include on-the-spot items derived from outside broad- 
cast units and from studios situated all over the country. You would be irritated 
indeed, if, every time the scene changed, your receiver suffered a momentary loss of 
synchronization ! 



1.82 



[§* 



Signal Routing 

It will help you to get an overall picture of the complex business of producing and 
distributing a television programme if you follow out the routing of the various signals 
which go to make up an imaginary Sports Round-Up televised by the BBC on a 
Saturday evening from Television Centre in London. 

The sports reporter who is compering the programme will be sitting in one of the 
many studios in the Centre, surrounded by a number of display boards and so on, 
enabling him to show (let us say) positions of the leading teams in the First Division 
of the Football League as a result of the day's matches. All the sound and vision 
signals emanating from this studio will be fed to a Central Control Room within the 
Centre, where they will be "married up" with the appropriate signals coming in from 
Regional studios, outside broadcast units, etc., carrying the sound and vision record 
of scenes from sports centres such as Aintree racecourse, the links at St. Andrews, or 
the football grounds of Sunderland, Aston Villa or Twickenham. 

From the Central Control Room, the signals representing the completed programme 
will be fed by cable to a Switching Centre situated in Broadcasting House some four 
miles away in Central London. From here the signals are relayed, also by cable, to 
a second Switching Centre in London which is owned and operated by the General 
Post Office. This GPO Switching Centre is responsible for routing the signals over 
GPO land-lines or by SHF microwave links to the several transmitting stations of the 
BBC National Network which are scheduled to broadcast the sports programme. 
(This GPO Switching Centre, by the way, provides an identical service for the trans- 
mitters of the ITA.) 

The diagram below shows how all these various points are inter-connected, the 
arrows indicating signal direction. You will see that the Switching Centre in Broad- 
casting House itself receives signals from outside sources; but these signals are all 
sent to Television Centre for technical assessment before being returned to the 
Switching Centre for subsequent transmission. 



^*^ ; 



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THE | 

TELEVISION CENTRE 



SIGNAL SOOTINC SySTM 



Outputs to 
Transmitters 
Holme Moss 



Lime 
Grove 
Studios 




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Other 
(local) 
Studios 






§6] 



1.83 



The TV Studio 

The layout and operating procedure of television studios varies in different countries 
throughout the world. Much depends on the scale of the programmes habitually 
produced, and on the type of equipment used. 

A good example of the more modern techniques in large-scale TV production is 
provided by the BBC Television Centre mentioned earlier; for this very large building, 
first opened for use in June 1960, presents the latest advances in TV engineering and 
architecture, its design being based on experience gained over more than 25 years of 
television broadcasting. The Centre is the nucleus of the entire TV organization of 
the BBC. From it are produced, or through it are routed, all the signals destined to be 
broadcast from its nation-wide transmitter network. 




The television studio itself varies widely in size, in function, and in the complexity 
of its technical equipment. It may be anything from a small room from which a 
news broadcast or a programme announcement is made or in which a personal 
interview is conducted, to an enormous stage holding scores of performers, masses of 
scenery and a bewildering array of complicated equipment. 

In the smaller studios the number of cameras and microphones used will rarely 
exceed two ; whereas in a large-scale production studio it is by no means uncommon 
for half-a-dozen cameras, and an even larger number of microphones, to be in use at 
any given time. 

The number of technical staff engaged on a TV programme depends, of course, 
on its complexity ; and also on whether it is being transmitted "live", as opposed to 
having been pre-recorded on video-tape or film for later transmission. A news 
broadcast, for instance, involving only a single announcer and perhaps a small 
number of filmed "shots", often makes use of pre-aligned cameras which are com- 
pletely unmanned and electrically controlled from a remote position. It therefore 
needs the services of very few controlling staff. 

On the other hand, the live presentation of a programme involving many different 
scenes, scores of performers, constant lighting changes and so on, will call for much 
planning, a great deal of rehearsal, and a large number of technical and artistic staff. 



1.84 



[§« 



Telecine and Video-Tape Machines 

Quite apart from signals coming into Television Centre from external sources — 
outside broadcasts, for example, and other studios both at .home and abroad — the 
producers of a TV programme are not confined to using only the scenes they them- 
selves create on the studio floor. There are, within the building itself, other sources 
of material on which they can draw when they want it. 

Two of these sources deserve mention. Essentially, they are both specialized 
pieces of recording apparatus, called respectively the telecine machine and the video- 
tape machine. They are permanently housed in a separate telecine suite in Television 
Centre, but can be switched on and off from the main Studio Unit (see next page) as 
and when required. Here is how they are used. 

The telecine machine converts ordinary cinematographic film into a sequence of the 
equivalent electrical signals, and delivers these signals when they are required for use. 
There are many types of scene (a large aircraft coming in to land is a good example) 
which obviously cannot be reproduced in the studio. The necessary shots must be 
filmed beforehand. When the picture of the landing aircraft is required in the 
production being televised, the telecine machine is switched on from the studio, the 
relevant length of film is processed in the machine, and the corresponding sequence 
of electrical impluses is fed at the appropriate point into the signal to be radiated. 

The video-tape machine works rather like the ordinary tape-recorder which you 
use to record a programme on the radio, or your own or your friends' voices, for 
later playing-back at your leisure. Instead of recording audio signals, however, the 
video-tape machine records on tape the video signals produced by a TV camera. 
This is usually one of the cameras being used on the studio floor, but there is no 
technical reason why it should not be an outside-broadcast TV camera instead. 

The sort of circumstance in which video-tape would often be used is in recording 
scenes in the studio which are to be artificially compressed in time in the televised 
programme. Imagine a comic sequence, for instance, of a man waking up late for 
work — leaping bleary-eyed out of bed, stumbling into the bathroom, gobbling some 
breakfast, dashing back because he has forgotten to kiss his wife good-bye, and then 
running like mad for his bus or his train ! The whole sequence might be effectively 
presented on the TV screen within the space of thirty seconds. No actor could 
possibly enact it "live" during so short a period ; so the various "bits and pieces" are 
pre-recorded on video-tape and fed into the final programme as required. 




Another use for video-tape is in the recording on tape of the electrical signals 
corresponding to a complete TV production, for re-broadcasting from another 
(possibly foreign) studio, or at a later date. 

Video-tape is quickly taking over much of the work hitherto recorded on photo- 
graphic film in the studio, or in "semi-fixed" installations outside. For work in the 
field, the greater portability of the photographic camera still gives it a big advantage — 
the film exposed being later converted into electrical signals by the telecine machine. 



§6] 



1.85 



How Programme Signals are Routed 

There are in BBC's Television Centre (in addition to a large number of studios of 
smaller size) seven major Production Studios, each designed to function as an inde- 
pendent unit. It will be useful to trace the path taken by signals originating in one 
of these production studios before they leave Television Centre on their way to the 
Switching Centres in Central London. 

Whatever their original source, all the sound and vision signals selected for trans- 
mission are passed to a Central Apparatus Room, which is the principal collecting 
point for all the signals intended to form part of the programme. In this Central 
Apparatus Room is housed much other technical equipment needed in the production, 
particularly the gear producing the synchronizing pulses for the cameras which you 
will read about very shortly. 

From the Central Apparatus Room, signals are fed through a routing unit to a 
Central Control Room, whose function is to maintain the continuity of the many 
programmes produced during the day's transmission, and to control the distribution 
of these programmes to the national transmitter network through the BBC and GPO 
Switching Centres. 

This Central Control Room itself forms part of what is called a Presentation Suite, 
the other unit of which is a small Presentation Studio from which will often be tele- 
vised programmes of the straightforward interview type, and also many of those 
scenes in which a personable announcer describes to the TV audience the attractions 
of the programme to follow. 



Production 
Studios 






'~E? 



Central 

Apparatus 

Room 



Telecine 



Routing 
Unit 



Presentation 
Studio 









Central 

Control 

Room 



Video 
Tape 



External 
Contri- 
butions 



PRESENTATION 
SUITE 



To B.B.C. ""' *»i 
and G.P.O. 
Switching Centres 

>- 



for distribution 
to National 
Transmitter 
Network 



BBC Television Centre PROGRAMME ROUTING SYSTEM 

The Studio Unit 

Every one of the seven major production studios in Television Centre contains its 



own studio floor, and all its own technical control facilities, 
unit contains four main areas: 



Essentially, each studio 



(1) The Studio Floor 

(2) The Production Control Room 

(3) The Lighting and Vision Control Room 

(4) The Sound Control Room 



1.86 § 6] 

The Studio Unit— The Studio Floor 

This is a very large, almost factory-like, room — typically as much as 100 ft. long, 
80 ft. wide, and 40 ft. or more in height. Slung beneath the ceiling of this enormous 
chamber, and running from side to side of it down its entire length, are a series of 
specially-shaped girders, supporting rails, brackets and the like, from which there 
hang and along which there slide a vast array of powerful lamps, pendant micro- 
phones, scenery and technical equipment of many kinds. 

Round the walls there will be set up a number of different stages, each complete 
with the scenery needed for a particular scene in the production being televised, and 
each awaiting only the actors to enter and speak their lines. The central areas of the 
floor will be largely occupied by a number of bits of machinery resembling hand- 
propelled tractors of different shapes and sizes, all mounted on rubber-tyred wheels 
and all having perched on their backs or sides shirt-sleeved technicians controlling 
cameras or swinging into position microphones hanging from long movable booms. 

On the floor itself there will be cables — hundreds of yards of cables of every size — 
cables connecting lights, cameras, microphones and technical equipment of great 
variety to their respective power supplies and control points. 

Carefully picking their way over this tangle of wires and between the array of 
"mobile machinery" will be people — actors, scene-shifters, camera and microphone 
crews, lighting and power technicians, secretaries, and the immediate staff of the 
Studio Manager himself. No rule can be laid down about the numbers of artistic, 
technical and other staff likely to be present on the studio floor at any one time. Like 
the quantity of scenery and of assorted lighting and electronic equipment used, all 
will depend on the nature and scale of the production being staged. For a really 
large production such as (say) a big court-room scene, a total of 50 people present at 
the same time on the studio floor would not be unusually large. 




§6] 



1.87 



The Studio Unit — The Production Control Room 

The Production, Lighting and Vision, and Sound Control Rooms are situated side 
by side at first-floor level at one end of the Studio Floor, with the Production Control 
Room in the middle. Each room has a large sloping observation window through 
which the controlling staff can see all that happens on the studio floor, and internal 
windows through which the Producer can signal to the other two rooms. 

The Producer himself is the man (or woman) who is in overall command of the 
whole Studio Unit while the particular production of which he is in charge is on the 
floor. From his seat at the Production Desk he controls in detail the way in which 
the production is to be finally presented to the viewer, choosing which of the pictures 
being produced by the cameras on the floor (there may be up to six of them on 
occasion) are best suited to the script at any moment, cutting in to them (maybe) 
pictures from sources outside the studio. This he may either do himself by the 
manipulation of a bank of electrical controls mounted on the desk before him; or 
(as is more usual) he may delegate the task to a person called the Vision Mixer, who 
then sits beside him. 

Assisting him in his task, and also sitting at the Production Desk alongside him, 
are his Secretary, who checks timing and calls out camera "shot" numbers; and the 
Technical Operations Manager, who is responsible for all the technical aspects of the 
production. 




Between the producer's team and the studio floor, a telephone "intercom" system 
makes it possible for instructions to be conveyed to the studio manager, to the 
microphone operatorsor to the camera-men at any time during the production. 

Standing in front of the production desk is a bank of nine 21-inch screen picture 
monitors. Every camera on the floor has its pictures displayed by one of these moni- 
tors ; while others display pictures derived from outside the studio. The ninth monitor 
displays the final transmission picture — that is, the actual picture which the producer 
has selected for transmission out of all the pictures displayed on the other monitors. 



1.88 



[§« 



The Studio Unit — The Lighting and Vision Control Room 

In this second of the three Control Rooms, the Lighting Supervisor is responsible 
for controlling all the lighting arrangements in the studio; and the Vision Supervisor 
for controlling the technical quality of the pictures produced by the studio cameras. 
These two operations are closely related technically, which is why the two supervisors 
share the same room. 

The two control desks are placed side by side in front of a bank of eight picture 
monitors displaying the pictures produced by the studio cameras. A ninth monitor 
stands on its own, and displays the final-transmission picture to the Lighting Super- 
visor alone. Of the eight monitors in the bank, up to six are linked to individual 
cameras on the floor; a seventh shows the final-transmission picture selected by the 
Producer in the next room; while the last is used for displaying pictures derived from 
outside the studio, which are about to be used in the programme. This last monitor 
can also be made to display pictures appearing on the camera monitors one after the 
other, in sequence. This greatly helps accurate matching of picture qualities, and 
ensures that false information derived from a defective monitor shall be detected 
before action is taken on it. 

Alongside every picture monitor, a narrow rectangular CRT displays the picture 
signal waveform produced by the camera associated with that monitor. This display 
helps the Vision Supervisor to maintain the black-level reference of the picture signal 
at its correct value. 

In front of the Vision Supervisor are six camera controls arranged in the same 
pattern as the six camera monitors, each control governing one camera down on the 
studio floor. By moving the appropriate control knob sideways along a quadrant, 
the Supervisor can adjust the lens aperture of any camera, thereby varying the light 
input to it. By rotation of the knob, he can alter the black-level setting of the picture 
signal waveform produced by the camera. By depressing the knob he can switch 
that camera's picture from its usual monitor on to the one used for comparison and 
matching. 




In the Sound Control Room, the Sound Supervisor has a desk from which he can 
control all the studio microphones; and also record-players, tape-recorders and 
special sound effects derived from outside the studio itself. This desk is positioned 
close to the observation window; and from it he can at all times see the position of 
every microphone on the studio floor. Should any microphone begin to be visible 
on the two picture monitors which he has in front of his desk, he can at once warn 
the operator over the intercom to move it out of the way. 



§6] 

The Camera Chain 



1.89 



One of the advantages of the modern TV camera is its manoeuvrability. This 
manoeuvrability can be effectively increased in two ways: by stripping the actual 
camera of every piece of accessory equipment which can possibly be put elsewhere ; 
and by freeing the operator from all possible duties which could interfere with his 
primary job of getting his camera into the best position all the time it is in use. 

To achieve these aims, there has been devised an arrangement called the Camera 
Chain. Every TV camera when it is in use will nowadays form part of a camera 
chain ; and no matter how many cameras are being used to record a production, each 
will require its own separate chain of ancillary equipment. All the parts of this 
chain, wherever situated, will be electronically interconnected as long as that particular 
camera is in use; and if one of the parts of the chain should fail, the whole of it will 
become unserviceable until the defective part can be repaired or replaced. 

There are five major links in a camera chain. They are: 

i) The master sync pulse generator situated in the Central Apparatus Room ; 

ii) The camera control unit (normally abbreviated to CCU) situated in the Vision 
Apparatus Room; 

iii) The camera itself and 

iv) The camera monitors, which you already know about; 

v) Camera signal control apparatus normally situated in the Production Control 
Room and in the Lighting and Vision Control Room. Its purpose is to enable the 
Producer and the Vision Supervisor to take effective action on the basis of the infor- 
mation presented to them by the camera monitors. 

The general layout of a camera chain is shown in block diagram form in the illustra- 
tion below. Remember that the purpose of this seemingly-complicated and expensive 
layout is to permit all possible technical assessment and control of camera perfor- 
mance to be exercised by specialists operating well away from the camera itself, so 
freeing the camera operator for his main job of keeping his camera accurately trained 
on the part or parts of a scene for which he has been made responsible. 



I Central Apparatus Room | 



)- 



To other Camera Chains -*- 



I 



"i 



Camera 



STUDIO FLOOR 



Master 
Sync Pulse 
Generator 

■31 _ J 
"1 



K- t^F^ 



Camera 
Control Unit 



I 



Vision Apparatus Room 



Other 



Monitors 



Camera 

Signal Control 

Apparatus 



Lighting.and Vision 

Control Room 

Monitors 



Production 

Control Room 

Monitors 



Sound 

Control Room 

Monitors 



I STUDIO CONTROL 



mm 



Transmission 
Signal 



SUITEjf} 



1.90 



[§« 



The Master Sync Pulse Generator 

It is the task of this very important piece of equipment to supply all the blanking, 
triggering and synchronizing pulses required by all the cameras and other pieces of 
equipment which are operating on a given line-system throughout the Television 
Centre. There are thus two master sync pulse generators operating in the White 
City TV Centre — one for the cameras, etc., working on the 405-line system, and another 
for all the equipment working on the 625-line system. 

The object of having all these essential, but "non-picture-signal", pulses supplied 
from a common source is to aid synchronization of all the equipment working on a 
particular scene. It frequently happens that the producer wants to use more than 
one camera, or even more than one signal source, as he builds up his presentation of the 
tale he is telling, and to feel free to jump from one signal source to another as he 
goes along. The chances of accurate synchronization are plainly improved if the 
various timing signals all come from a common source. 

The illustration below shows the block diagram of a master sync pulse generator 
controlling the equipment of a 405-line-system camera chain. The sync pulse genera- 
tor controlling a 625-line camera chain operates in much the same way, but is rather 
more complicated in view of the need to generate the equalizing pulses (see Section 5). 

The MASTER SYNC PULSt GENERATOR 





Line Blanking 

Pulse Generator 

(10,125 c/s) 












Line Sync 

Pulse Generator 

(10,125 c/s) 




























* 




' 


' 




Line Trigger 
Pulse Generator 




Frequency 

Divider Circuit 

(-2) 


























Blanking 
Mixer 






Sync Pulse 
Mixer 












* 






Field Trigger 
Pulse Generator 




Oscillator 
(20,250 c/s) 






















' 


' 




















4 






t 




Field Blanking 

Pulse Generator 

(50 c/s) 






Frequency 

Divider Circuits 

(*405) 






Field Sync 

Pulse Generator 

(20,250 c/s) 












< 




< 


r Line 

Tr 'gg 

Pulse 


er 
s 


' 


Field 

Trigger 

Pulses 




Mixed Sync Pulses 






> ecu: 






»» 






»► M 


r * 











The heart of the master sync generator is a very stable oscillator, whose sinewave 
output is made rectangular by being passed through a selection of the pulse-shaping 
circuits you learn about in Basic Electronic Circuits, Part 1 . This is done within the 
block marked "oscillator" in the illustration. 

The operating frequency of the oscillator is always twice the frequency at which the 
line sync pulses recur in the particular TV system in question. Operating frequency is 
therefore, 20,250 times per second in the 405-line system, and 31 ,250 times per second 
in the 625-line system (twice the number of lines per field multiplied by 50 fields per 
second). A basic frequency of this value is needed in order to provide the "raw 
material" for the half-line pulses of the field sync pulse, and for the equalizing pulses 
also needed by the 625-line system only. 



1 6] 1.91 

The Master Sync Pulse Generator (continued) 

The shaped pulses from the oscillator block are fed in two directions. In the 
first path, they go to the field sync pulse generator, which produces the half-line 
pulses making up the field sync pulse. 

These half-line pulses, in turn, take two paths. The first takes them through 
frequencer divider circuits of the type you met on page 1.79 of Basic Electronic 
Circuits, to trigger the field blanking pulse generator. The output of this generator is 
required to suppress the electron beam during field fly-back. It therefore needs to 
be a sequence of pulses recurring at a frequency of only 50 Hz. Thus the frequency of 
the half-line pulses needs to be reduced in the divider circuits by a factor of 405 in the 
405-line system, and in the 625-line system by a factor of 625. You learnt in Basic 
Electronic Circuits that the maximum practical limit of the synchronized multivibrator 
was about a ten-to-one count-down. To achieve a 405-times count-down therefore 
means that the half-line pulses have to be passed through three count-down circuits 
in succession, achieving frequency division of 9, 9 and 5 times respectively. 

The field blanking pulses, when formed, are fed to a blanking mixer stage where they 
are mixed with the line blanking pulses whose source you will be tracing in a moment. 

The 50 Hz output signals from the frequency-divider block are put to two other 
uses. In both systems, they are made to generate triggering pulses for the field 
scanning circuits in the respective CCU's (see next page) ; and in the 405-line system 
they are also fed to a frequency-control circuit (not shown on the block diagram 
opposite) whose job is to compare the frequency of the 50 Hz pulses with that of the 
mains supply and, if a difference develops, to generate a controlling voltage which 
brings the two frequencies back into synchronism by correcting the frequency of the 
master oscillator itself. (No such frequency-control arrangement is needed in the 
British 625-line system because of the improved circuit techniques which were available 
when the system was designed.) 

Back, now, to the field sync pulse. This pulse must recur, as you know, fifty 
times a second and consists of a group or cluster of shorter (half-line) pulses varying 
with the line-system concerned. The second job of the field sync pulse generator is 
therefore to marshal the half-line pulses it produces into the appropriate clusters, to 
add to them (in the 625-line system) the equalizing pulses, and to feed the field sync 
pulses so formed to a sync pulse mixer stage for mixing with the line sync pulses. 

Now for the generation and routing of the line blanking and line sync pulses. 

In the second signal path from the oscillator block, the signal is fed to a count- 
down circuit which divides its frequency by two, to produce a signal at the line 
frequency appropriate to the line-system (405 or 625) concerned. The first output 
from this circuit is used to generate the trigger pulses needed by the line scanning 
circuits in the respective CCU's; the second goes to the line sync pulse generator. 

The output of this generator is the stream of very short-duration pulses required to 
trigger the line scans in the camera (and, later, in the picture tube of the receiver). 
It again splits into two. The first output goes to the sync pulse mixer; the second 
to the line blanking pulse generator. In this block, the line pulses are shaped to the 
duration and amplitude required for their beam-blanking function during line flyback, 
and are then sent to join the field blanking pulses in the blanking mixer stage. 

The much-simplified block diagram of the Master Sync Pulse Generator on the 
page opposite shows, finally, how the mixed blanking pulses, the mixed sync pulses, 
and the line and field trigger pulses are all distributed to their respective CCU's, 
and to the other pieces of apparatus in the TV Centre which will need to use them. 



1-92 [§6 

The Camera Control Unit (CCU). 

Whereas the master sync pulse generator is common to all the camera chains 
working on a given line system throughout the TV Centre, there needs to be a separate 
CCU for every camera employed. 

The CCU is housed (as you know) in the Vision Apparatus Room which forms 
part of every studio unit. It provides the operating voltages needed by the camera 
and by several other pieces of equipment in the camera chain. Most of its other 
functions can be seen in the block diagram below. 



TV CAMERA 



Viewfinder 



Camera Tube 



Picture 

Signal 

Amplifier 



"I 



Scanning 

Waveforms 

Generated 



Operating 
voltages 



Blanking 
Pulses 
inserted 



The C.C.O. 



Line and Field Trigger Pulses 



Talk-back 
apparatus 



Clamping 

and 
Limiting 



VIDEO SIGNAL 

to Camera Viewfinder 

and all Monitors, 

and to Transmitter 

f t t t 



Sync Pulses 
Inserted 



Distribution 
Amplifier 



Amplifier 



T 



Amplifier 



Mixed Sync Pulses 



Mixed Blanking Pulses 

You will see, at the top of the diagram, where the line and field trigger pulses from 
the master sync pulse generator initiate the scanning waveforms for the camera. 
Next, moving downwards from the camera, you see the picture signal being amplified, 
and then having inserted into it in turn the mixed blanking pulses and the mixed 
sync pulses from the master sync pulse generator. In between these two insertions 
is a block in which the signal may be clamped to a suitable black-level reference, 
and will have restored to it the d.c. level lost in the camera coupling circuits when the 
signal passed through the picture signal amplifier. 

The complex video signal resulting from all these operations is further amplified 
in the distribution amplifier, and is fed to the viewfinder of the camera and all the 
other monitors attached to its camera chain. It is also, of course, sent on its way 
towards the distant transmitter, if it happens to be the one selected by the producer 
to tell that particular part of the story. 

Other functions of the CCU are to provide the one-way talk-back and cueing 
apparatus whereby the production control staff can pass instructions to the individual 
camera operators, and to perform a number of technical operations on the video 
signal before it is distributed — mainly with the aim of making it as free from distortion 
as possible, and of the correct amplitude relative to the amplitude of the sync pulses. 

The Camera Signal Control Apparatus 

The several pieces of equipment which go to make up this apparatus allow the 
producer to bring into play several of the various camera techniques described on the 
next few pages, and his technical assistants to keep check on the quality of the picture 
signals produced by a given camera and to adjust the quality of these signals when 
necessary. 



§6] 1.93 

Camera Techniques 

Many of the special techniques used in the studio to help the smooth presentation 
of a television scene go completely unnoticed by the average viewer. You should 
know, in outline, what some of these techniques are, what they are used for, and how 
they are achieved. 

Cutting 

Imagine that you are sitting facing two people in your own home, and listening 
to their conversation. As each one speaks in turn, your eyes normally turn automati- 
cally to the speaker as you listen to what he or she is saying; and then flick equally 
automatically to the face of the other as he starts to reply. It may sometimes happen 
that one wants to watch the face of the non-speaker to observe his reactions to another's 
speech — as when, for instance, a detective is watching a suspect being questioned 
at a police station — but the point is that the face which one wants to be watching is 
seldom the same one all the time. 

Now suppose two people are being televised holding a similar sort of conversation. 
Your eyes will now be replaced by a camera — and ultimately by a viewer watching 
from many miles away. If this viewer is to follow the conversation as intelligently 
as you did earlier, he too will want to turn his eyes to the face of principal interest- 
nearly always that of the person speaking — as the conversation ebbs and flows. Since 
he cannot do this for himself, the producer must do it for him. How ? 

The way it is done is to have two TV cameras, each focused on to one of the people 
being televised, and each operating all the time. In the Production Control Room, 
the producer watches the two pictures produced and selects the one he thinks most 
interesting at any given moment, constantly switching from one face to the other as 
the conversation proceeds. This switching from one camera to another must be 
instantaneous, with no sign of the blur which would result if a single camera were to 
swing back and forth from one person to another as the scene unwinds. 

This form of instantaneous switching is called cutting, and every separate switching 
operation is called a cut. The producer achieves a cut by depressing a button on a 
panel in front of him, which selects the wanted signal. 

Fade 

There are two kinds of fade used in the telling of a TV story— fade-out and fade-in. 
The first is commonly used to mark the slow, almost reluctant, ending of the action 
in one phase of the story; the second to usher in the beginning of a new episode. 
You could almost say that fade-out is one of the TV producer's ways of ending a 
chapter, and fade-in one of his ways of starting a new one. 

Fade-in and fade-out are achieved by slowly increasing or decreasing, respectively, 
the amplitude of the picture signal produced by the camera recording the scene. 
Control of this operation will be exercised directly by the producer or by one of his 
staff in the Production Control Room— not by the camera operator himself. The 
fader control is essentially a potentiometer controlling the amplitude of the picture 
signal content of the video signal sent to the Central Apparatus Room for transmission. 

The overall duration of a "fade-out/fade-in" between consecutive scenes will 
seldom exceed ten seconds, and will typically be about half that period of time. 



[§« 



1.94 

Camera Techniques (continued) 

A dissolve is the technical name given to an operation which consists of a fade-out 
from one camera and a fade-in to another one recording a different scene. By deft 
control of the relative picture signal amplitudes of the two cameras, the pictures they 
produce can be made to merge, or dissolve, into one another, until finally one picture 
supersedes the other altogether. 

A typical application of a dissolve would be the shot of an aeroplane taking off from 
a runway, merging into a scene showing the passengers inside the aircraft unfastening 
their safety belts once the plane has gained safe height. A smooth transition is thus 
effected to a subsequent episode of the story — generally to one taking place after the 
elapse of an uneventful interval of time. 

A normal dissolve would last for some five or six seconds from start to finish. 
Defocusing 

This operation achieves much the same purposes as a dissolve, and is done by 
slowly defocusing the camera recording one scene as the scene recorded by another 
camera is equally deliberately faded in. 

Defocusing is achieved by altering the setting of the camera lens in relation to the 
target of the picture tube. With some types of camera this can be done directly 
from the Production Control Room via the CCU; but it is usually controlled by the 
camera operator on the studio floor under instructions transmitted to him by the 
producer over the "talk-back" apparatus already described. 

Panning 

This is the name given to the slow revolution of a camera on its swivel base as it 
records in turn (let us say) the expressions on the faces of a jury as they take in the 
significance of a dramatic piece of new evidence. 

The Camera SWiMS along toe «Iiiry Bench... 




and a PANNING Shot 

Results on the Viewer's Screen 

A camera so revolving is said to be panning. The picture recorded by it as it 
revolves is called a pan shot. 



§6] 



1.95 



Camera Techniques (continued) 

The normal way of producing in the studio a gradual close-up of part of a scene 
first viewed from a distance is to move a camera slowly towards the scene until it is 
close enough to produce the desired effect. This is generally done by pushing the 
camera dolly (the rubber-tyred trolley on which the camera is mounted) silently by 
hand into the required position. 

In outside broadcasts, however, or in situations in which permissible camera 
movement is restricted (e.g., the recording "live" of an opera at Covent Garden), 
the close-up desired by the producer is achieved by the gradual expansion of a powerful 
lens system mounted in front of the lens of the camera itself. Such a system is known 
as a zoomar lens. It is only fair to warn you that you will not fully understand the 
brief account of it which follows unless you already know the meaning of a few basic 
photographic terms. 

There are three separate lenses in a typical zoomar system, mounted in line with 
only the centre lens movable. The fixed front and back lenses are of the positive 
convergent type, the movable centre lens of the negative divergent type. When the 
centre lens is moved towards the front lens, the lens system acquires a longer effective 
focal length, and the effect of close-up (or zoom-in) is achieved. When the centre 
lens is moved backwards towards the rear lens, the effective focal length is reduced 
and zoom-out results. 

An important advantage of the zoomar lens is that although the apparent size of 
the object viewed can be made to vary with the degree of zoom-out or zoom-in, the 
ratio of focal length to lens diameter (the / number of the lens system) remains un- 
changed. This means that the lens aperture does not have to be altered as zoom-in 
proceeds, and the brightness level of the object being viewed does not vary with the 
degree of close-up. 





200*1 



0tfT 



00JM 



ftf 






Tilt 

The eye of a camera leaves street level and slowly travels up, up, UP the vertical 
flank of a skyscraper in New York City, until it comes to rest at last on the 
immensely foreshortened uppermost floors silhouetted against the sky. . . . 

This operation is known as a tilt. It is effected by the camera operator himself. 
The need to take tilt shots when required is one of the reasons why a modern camera 
needs to be very carefully balanced on its swivelling base. 



1.96 



[§« 



Camera Techniques — Special Effects 

Three types of special effects developed in the cinematograph industry are being 
put to increasing use in TV. They are back projection, inlay and overlay. Collec- 
tively, they are sometimes called montage effects. The object in each case is to 
persuade the viewer that something is happening on his screen which it would be 
either difficult, impossible or very expensive to photograph direct. 

In the back projection technique (it is also known as process projection), either an 
ordinary cinema projector or a "magic lantern" is set up behind a translucent screen 
and is made to project on to it either a moving or a still image of a desired piece of 
scenery. The actors perform in front of the screen, and the TV camera field takes in 
not only them and any foreground "props" which the action requires, but the pro- 
jected background as well. 

In this way it is possible to fake pictures of a man talking to a girl in the front seat 
of a moving car with the road unwinding behind them, or of the same man posing 
with the same girl in front of the Taj Mahal, without (in the first case) mounting a 
backward-pointing camera on the bonnet of a moving car or (in the second case) 
sending man and girl, plus technicians and much heavy equipment, all the way to 
India. 





■v.. rift 


ft m 


n ifi 



Difficulties of the back projection technique include the following: 

a) It takes up a lot of space on the studio floor. 

b) The translucent screen must be very powerfully lit so as to acquire a level of 
brightness which the TV camera can use; and this brightness must be uniformly 
spread over the whole screen surface. 

c) Noise from the projector must be so low that the microphones recording the 
main action will not pick it up. Projector noise includes the whirring of a moving- 
picture camera, and the hiss of the escaping air blast commonly used to cool the slide 
of a "still" projector. 

Lighting difficulties are the most serious; for if light from sources other than the 
projector (spill light, as it is called) is not kept to a minimum, the average brightness 
of the projected image will be too low to "look right" to the TV camera. The 
problem is particularly acute in the first few feet immediately in front of the screen. 

You will see that back projection is essentially an optical means of combining the 
output of two cameras, or of one camera and one "still" projector. Inlay and overlay 
are electronic methods of doing the same thing. 



§«] 



1.97 



Camera Techniques — Special Effects (continued) 

Inlay is a technique used for combining the outputs of two signal sources in such a 
way that, in the composite picture shown to the viewer, the foreground scene replaces the 
background scene over certain areas of the latter which remain fixed throughout the process. 
Here is how it might be used in the shooting of a TV adventure story. 

At a time of national crisis, the President of Ruritania steps out on to the balcony 
of his palace to make a pronouncement to the anxious crowds below. The distinctive 
facade of the Presidential Palace is known throughout Europe; and it is undoubtedly 
that facade which you are watching as the President speaks. How, short of hiring 
the palace for the day from its present incumbent, has the scene been shot? 

What happens is that one camera (or a telecine film scanner) will be kept throughout 
the speech trained on to a library photograph or slide of the palace itself. Elsewhere 
in the studios a second camera will be focused on to a facsimile balcony standing on 
an empty stage, from which the actor impersonating the President will deliver his 
oration. The outputs of the two cameras will then be matched in the Production 
Control Suite in such a way that the balcony, and the President within it, will appear 
anchored in its correct location on the palace wall. 




The principal drawback of the technique would arise if the "President" were to be 
so carried away by his own eloquence as to move outside the strict limits of his stage 
balcony. Not only would he then appear to the viewer to begin moving crabwise 
and without visible means of support across the vertical facade of his palace. He 
would also turn into a semi-transparent ghost as he did so. 

The reason is, of course, that the background signal would no longer be suppressed 
as the foreground signal moved over it. Instead, both signals would appear simul- 
taneously on the viewer's screen. 

The inlay technique is thus a means of cutting what amount to "electronic slots" in 
certain fixed areas of the background scene, and of fitting into these slots appropriate 
bits of separately-photographed foreground scene. In briefest outline, it is done by 
placing an opaque mask between a short-afterglow CRT of the type used for film 
scanning, and a photo-multiplier type of photocell. The mask has a cut-out of the 
correct size and shape, and in the correct area, to hold exactly a foreground shot of 
the President gesticulating on his balcony. The scanning beam of the CRT is 
synchronized with the scanning beams of the two cameras. When the beam is 
scanning areas obscured by the mask, the photocell receives none of the glow of the 
CRT, and the signal produced by the camera recording the background scene goes 
forward for transmission. 

When the beam of the CRT moves into the area of the slot, however, the photocell 
picks up its glow and at once produces what is called a silhouette pulse. This pulse 
operates an electronic switch, whose function is to select for transmission only the 
picture signal produced by the camera recording the foreground scene — and that only 
for as long as the silhouette pulse is present at the switch. In this way, only one 
signal at a time is accepted from the two picture-signal sources. 



1.98 [§6 

Camera Techniques — Special Effects {continued) 

An interesting example of the inlay technique allows a whole range of effects to be 
produced by the use of false perspective. In this way, the miniature figure of a girl 
can be made to dance on top of a piano played by a pianist of normal size, or a giant 
can be made to inhabit an ordinary house by having his blown-up head "slotted in" 
so that it looks out of one of the windows. 

Another adaptation of the technique enables any desired type of transition to be 
made from one scene to another, with the second scene superseding the first as a line 
of demarcation passes gradually across the screen. This is known as a wipe, and is 
achieved by passing a mask of the desired shape across a small optical image of the 
scanning raster. 

Overlay is essentially similar in the results it produces to back projection. The 
difference is that it uses electronic means of placing a desired background behind an 
"acted" foreground, whereas back projection uses optical means of doing so. Over- 
lay, however, can provide a greater range of effects than can back projection, and it 
has the advantage of not requiring a translucent screen or the space taken up by a back 
projector. All it needs is a plain screen either totally unlit or else more conveniently 
lit from in front. 

There are several ways of achieving overlay, but in all of them a signal representing 
a moving outline of the foreground figure (in effect a more complex type of silhouette 
pulse) is made to operate an electronic switch. The switch selects the background 
signal from Camera 1 until a silhouette pulse arrives, when it selects instead the fore- 
ground picture from Camera 2 as long as the pulse is present. 

The actor performs in front of a plain screen, which may be either white or black. 
A particularly wide range of effects can be produced when a black background is 
used. For instance, if the actor's face, neck and hands are swathed in black cloth, 
and if the lighting is so arranged that they will not be visible against the black back- 
ground, an empty suit of clothes can be made to walk about a room and a very 
creditable Invisible Man produced on the screen. 

What has happened electronically is that the silhouette pulse which would normally 
have been generated as soon as the scanning beam encountered (say) the actor's face 
has been literally "bluffed out of existence", with the result that the background signal 
is transmitted in its place — and the picture over the mantlepiece appears unobscured 
between the Invisible Man's collar and the brim of his hat! 

Camera Techniques — Video Effects 

Other special effects can be produced in TV by other electronic means. An example 
is the deliberate exaggeration of the contrast range of the picture produced by a 
camera, by manipulation of its appropriate amplitude control. 

A troupe of dancers, let us say, is performing on a dimly lit stage. They are all 
wearing black make-up or black face masks, and all are dressed in black from head to 
foot save for a pair of white gloves each. By slowly reducing the ambient lighting 
and simultaneously increasing the amplitude of the video signal, the swinging white 
gloves can be made to stand out more and more against a steadily darkening back- 
ground, until the very movement of the dancers themselves become invisible and only 
the white gloves swing and swing 



§ 7: THE TRANSMITTER AND AERIAL 



1.99 



Once it has left the TV studio, the video signal passes through the various Switching 
Centres you learnt about in the last Section, to the transmitting station situated from 
a few to several hundred miles away. It is accompanied by the audio signal. 

By no means all TV transmitting stations are equipped to radiate television signals 
only. Many serve as combined TV and sound-radio broadcasting stations, with 
the sound side radiating a frequency-modulated carrier of very high frequency (i.e., 
a "VHF/fm station"). Such a combined station will often be radiating as many as 
three different radio programmes as well as a single TV programme. Both the TV 
and the radio transmitters will be installed in the same building; and will be super- 
vised by the same team of engineers, assisted by automatic alarm devices common to 
both transmitters. 

The TV side of such a composite transmitting station will itself comprise at least 
two separate transmitters — one to handle the video signal and the other to handle 
its associated audio signal. The two transmitters will be situated close together, often 
side by side. Their outputs will be the vision signal and the sound signal respectively. 

These two signals are then usually combined in the last stage of their processing 
in the transmitter building, and are taken over a common transmission line to the 
aerial, from which they are radiated as a combined signal to the distant viewers. 

You will remember from what you read in Part 4 of Basic Electronics that, before 
you can radiate over any distance waveforms of voltage or current such as those 
which are produced by a microphone or television camera, you must first convert 
the waveforms into high-powered radio-frequency signals. These r.f. signals must 
then be propagated as electromagnetic waves. 

The first of these tasks is performed by the transmitter, the second by the aerial. 
Together, these two elements make up the transmitting station. 

Both sound and vision transmitters work on much the same principles. The 
transmitter consists of a very stable oscillator which produces a high-frequency 
carrier wave of fixed frequency and amplitude; of a modulator stage in which the 
signal is combined with the carrier in such a way as to modulate either its frequency 
or its amplitude ; and of a power amplifier in which the modulated carrier is raised to 
a power high enough for the aerial to be able to propagate it to the desired range. 



The Transmitting Station 



Carrier 



Modulated 
Carrier 



Oscillator 



Modulator 



Audio or Video Signal 



Power 
Amplifier 




Aerial 



1.100 [§7 

The Vision Transmitter 

The illustration below sets out in block diagram form the essential features of a 
typical vision transmitter. 

The first stage in the transmitter will normally be a highly stable crystal-controlled 
oscillator circuit. It is important that the carrier wave be maintained at a very stable 
frequency, both for reasons of general transmitter efficiency and so that the frequency 
of the radiated signal shall be maintained at its assigned value throughout the period 
of transmission. 

It is generally easier to control an oscillator operating in the VHF range when the 
frequency is kept as low as possible. For this reason, the frequency of the crystal- 
controlled oscillator is arranged to be lower than the frequency at which it is intended 
to radiate the signal. 

The frequency of the "basic carrier" is then stepped up to the intended operating 
frequency of the transmitter by means of one or more frequency-multiplier circuits 
such as those you studied in Basic Electronics, Part 4. These circuits, as you know, 
are capable of producing an output whose frequency is an exact multiple of the 
frequency of the input. Since this input frequency was highly stable in the first 
place, the stability of the stepped-up frequency will be maintained without impairment. 



"S%. 




Signal from 

Sound Transmitter 




mm mm 



Crystal 
Oscillator 



Frequency 
Multipliers 



R.F. 
Amplifiers 



VV 



Video 
Amplifiers 



COMPOSITE VIDEO SIGNALS 
FROM STUDIO 



VISION SIGNAL 



§ 7] 1.101 

The Vision Transmitter (continued) 

The carrier emerging from the frequency-multiplier circuits is then amplified to a 
level suitable for modulation by being passed through one or more r.f. amplifier 
circuits ; and is then fed to the modulating stage. 

Also fed to this modulating stage, after considerable amplification in a number of 
video amplifying circuits, are the composite video signals reaching the transmitter by 
high-quality coaxial cables (or, in some cases, by SHF microwave relay) from the 
switching centres and the TV studio. 

The effect of applying to the same stage both the varying voltages of the video 
signals and the r.f. carrier with its highly stable frequency and amplitude is to make 
the amplitude of the carrier vary in exact accordance with the variations occurring in the 
amplitude of the video signals. The effect is shown in the illustration below. 



This ..*J$***^ This e $ This 

CARRIER ^g^*<" VIDEO SIGNAL «rO* W VISION SIGNAL 





The vision signal resulting from the above process is, of course, none other than an 
amplitude-modulated wave of the type with which your work in Basic Electronics 
has made you familiar. Note carefully that, unlike the sound carrier which may be 
either amplitude-modulated or frequency-modulated, the vision carrier is always 
amplitude-modulated. 

After modulation, the carrier is again amplified many times by a number of powerful 
amplifiers, so that the radiated signal shall have sufficient strength to meet the range 
requirements of the transmitter. The signal is then passed through an impedance 
matching stage to a short length of feeder cable. This feeder takes the signal into a 
circuit called a sideband filter, whose object is to limit the frequency bandwidth 
required by the radiated signal. 

The vision signal (less most of one of its sidebands) is then combined with the 
sound signal (complete with both its sidebands) in a combining unit; and the two 
signals are fed to the aerial through special low-loss transmission lines. From the 
aerial the two signals are radiated in the form of electromagnetic waves. 

Up to the time when it emerges from the Power Amplifier stage, the vision signal 
is handled in much the same way (in principle) as is the amplitude-modulated sound 
signal. It will be convenient to break off for a moment, therefore, and have a look 
at how the TV sound transmitter works. 



[§7 



1.102 

The TV Sound Transmitter 

The television sound transmitter is in principle like any other transmitter used for 
the transmission of sound signals. Indeed, radio and TV sound transmitters of the 
BBC have often been used in conjunction with one another for the transmission of 
experimental stereophonic programmes. 

TV sound transmitters, like their radio counterparts, are of two main types — 
those in which the amplitude of the carrier is modulated, and those in which its 
frequency is modulated. The present British 405-line TV system employs amplitude 
modulation for the sound carrier, as do the French 819 and 625-line systems. 
Countries using the CCIR-recommended 625-line system, however, use frequency 
modulation of the sound carrier; and it is this type of modulation which is used in 
the British 625-line system (BBC 2). 

Frequency modulation, as you know, differs from amplitude modulation in that 
the FM carrier is maintained at constant amplitude while its frequency is altered by 
amounts which vary with the amplitude of the modulating signal. In amplitude 
modulation it is the other way about — the frequency remaining steady while the 
amplitude varies with the signal. The waveforms below show what an r.f. carrier 
looks like when it is modulated by either method. 



AMPLITUDE 
MODULATION 



FREQUENCY 
MODULATION 





AF MODULATING 
SIGNAL 



RF CARRIER 






CUKKIIK MODOl^ 



0* 



The first stage in the generation of the sound carrier, as it is of the vision carrier, is 
an oscillator circuit (usually crystal-controlled) operating at a fundamental frequency 
considerably lower than that of the carrier which is eventually to be radiated. The 
carrier produced by this oscillator is fed to a modulator stage, where it is mixed with 
the sound signal. 




Generation *f the 
TV SOUND CA**t£* 



(1st. STAGE) 



Audio 
Amplifier 



Modulator 




§7] 1-103 

The TV Sound Transmitter {Continued) 

You learnt in Part 4 of Basic Electronics how amplitude modulation of the carrier 
wave is effected. Here is a diagram to remind you of the process. 









F 


requency 

Multiplier 

Stages 














Crystal 

Controlled 

Oscillator 




1 


2 


3 


n 




R.F. 
Amplifiers 


Modulated 
Amplifier 


















F« 




From Studio 


A.F. 

Amplifiers 
























Aerial 



AMPLITUDE-MODULATED Transmitter 



Oscillators tend to be more stable at low frequencies in the VHF range, so a crystal 
oscillator with a frequency of 5 MHz might be used to provide the fundamental 
frequency for a 50 MHz carrier. It would be brought up to the required frequency 
by being passed through a number of frequency-multiplier stages. 

The carrier is then amplified (in the R.F. Amplifiers block in the diagram above) 
until it is strong enough to drive the power amplifier in the next stage. 

The audio signal will seldom come in from the studio with an amplitude of more 
than about one volt. It must therefore be put through a number of a.f. amplifier 
stages before it can be applied to the modulator. 

As you learnt in Basic Electronics (pages 4.79 onwards), the modulating signal can 
be applied either to the anode or to the grid of the final power amplifier of the trans- 
mitter. In either case, PA output current (and so the amplitude of the carrier) is 
varied in precise ratio to variations in the amplitude of the audio signal. 

The final amplifying stage, when modulated in this way, is known as the modulated 
amplifier. Besides combining the audio signal and the carrier, the stage produces 
most of the power required to radiate the signal from the aerial. 

In some systems, modulation of the carrier is performed at an earlier stage in the 
transmitter, and the signal is amplified to its final strength only after modulation. 
Such a process is called low-level modulation, as opposed to the high-level modulation 
technique described above. 



1.104 



[§7 



The TV Sound Transmitter — Frequency Modulation, Direct Type 

Frequency modulation of the carrier wave was explained in Part 6 of Basic Elec- 
tronics. There are two main methods, both of which are currently used in trans- 
mitters of the BBC. The first, the direct type of frequency modulation, employs an 
LC oscillator whose frequency is directly modulated by a reactance modulator. The 
advantage of this method is that frequency-modulation of the carrier is easily achieved. 
Its main disadvantage is that the stability of the mean frequency of the oscillator 
is often poor, so that a special frequency-stabilizing stage has to be added to the 
circuit (see next page). 

The block schematic of the reactance-modulated, direct-type FM transmitter 
which follows should be studied in conjunction with the more detailed explanation 
of its working given in Basic Electronics, Part 6. 




Aerial 



Briefly, what happens is this. The audio signal received from the studio is ampli- 
fied, and then fed to a reactance modulator stage whose function is to modulate the fre- 
quency of the master oscillator in direct proportion to the amplitude of the audio signal. 

The reactance modulator is a circuit which is made to behave in a manner very 
like a variable capacitor. It is connected to the fundamental oscillator in such a 
way that its effective capacitance is placed in parallel with (and therefore forms part 
of) the tuned circuit of the oscillator. In the absence of a signal input, the effective 
capacitance of the modulator is steady ; but when a signal is applied, the capacitance 
varies by amounts which depend on the amplitude of the signal input — large amplitude 
signals causing much greater changes in effective capacitance than signals of lesser 
amplitude. 

You know that the frequency of an LC oscillator is governed by a formula in which 
the variables are the total inductance and the total capacitance in its tuned circuit 

( f = 2n\/Tc ) ' ^° ^ e f re< l uenc y °f tne master oscillator must vary every time an 
audio signal applied to the reactance modulator changes the effective capacitance 
in the oscillator tuned circuit — and it must vary by an amount which is directly 
proportional to the changing amplitude of the audio signal. 
The amount of this variation is called the frequency deviation. 



§7] 1-105 

The TV Sound Transmitter— Frequency Modulation, Direct Type (continued) 

From the fundamental oscillator, the now-modulated carrier passes to a number of 
frequency multiplier circuits like those used in the vision transmitter. These circuits 
not only multiply the fundamental frequency itself, but also (and in exactly the same 
proportion) the deviations in its frequency caused by the audio modulations. 

Take, for instance, an oscillator operating at 5 MHz and having a maximum fre- 
quency deviation (corresponding to the maximum amplitude of the modulating signal) 
of 375 kc/s. If the final operating frequency of the transmitter is to be 100 MHz, 
the frequency-multiplier circuits must clearly step up the carrier frequency by a 
factor of 20. This exact multiplier will also be applied to all the deviation frequencies, 
and will step up the maximum deviation to 20 times 375, or 75 kHz. 

Later stages of the transmitter increase the power of the modulated carrier to a 
level high enough to meet its range and signal-strength requirements. Typical 
power outputs actually radiated from the aerial vary from about 3 kilowatts, up to 
120 kW for the longer intended ranges. 

Control of the rather unstable mean frequency of the master oscillator in the 
frequency-modulation (direct type) circuit can be achieved in more than one way. 
One such method was described in Part 6 of Basic Electronics. Here is another. 

Part of the output signal from the master oscillator is fed to a. frequency-comparator 
circuit (see block diagram on preceding page). This circuit compares the frequency 
of the master oscillator signal to the frequency of a very stable crystal-controlled 
reference oscillator. If the frequency of the master oscillator differs from that of the 
reference oscillator, a difference-frequency (or beat-frequency) signal is produced. 

The comparator uses this signal to produce a controlling voltage for a small servo 
motor. The greater the difference in the frequencies of the two oscillators, the greater 
will be the amplitude of the controlling signal. 

The shaft of the servo motor is mechanically linked to the shaft of a variable 
capacitor which forms part of the tuned circuit of the master oscillator. If the 
frequency of the master oscillator is higher than it should be, the polarity of the 
controlling signal will be such that the motor is turned in a direction which will 
increase the value of the variable capacitor. From the frequency formula, you know 
that this will cause a reduction in the frequency of the master oscillator. 

Conversely, if the frequency of the master oscillator is lower than it should be, the 
polarity of the beat-frequency control signal will be such as to turn the servo motor 
in the opposite direction. The value of the variable capacitor in the tuned circuit of 
the master oscillator will be reduced; and the effect of introducing a carefully measured 
lower value for C into the equation will be that the frequency of the master oscillator 
is increased by the amount needed to bring the two oscillator frequencies back into 
balance. 



1.106 [§7 

The TV Sound Transmitter — Frequency Modulation, Indirect Type 

The second currently-used method of frequency modulation is the indirect type. 
It employs a crystal-controlled master oscillator having excellent frequency stability, 
the output of which is (with more difficulty than in the direct type) frequency-modu- 
lated at a later stage. Here is a block diagram of the circuit arrangement. 



Crystal- 
Controlled 
Master 
Oscillator 




Phase 




R.F. 
Amplifiers 




Power 
Amplifiers 




Modulator 






















Audio 
Correction 
Network 


Feeder 




. 


t 






A.F. 
Amplifiers 




V0i; Audio Sigr 
V#£v& from St 


al 
idio 


. 


. 


... Al*' . 




Aerial 



The 



<&**- 



No controlling device is needed by a crystal oscillator save for a thermostatic 
arrangement to keep the crystal itself at a constant temperature. 

Modulation of the carrier is achieved in the next stage, by means which you studied 
on page 6.18 of Basic Electronics. You there learnt that when a constant-frequency 
r.f. signal is applied to a network consisting of either a capacitive or an inductive 
reactance in series with a potentiometer, the phase of the output signal can be shifted 
by varying the resistance. Substitute a triode or transistor for the potentiometer and 
apply the audio signal to its grid (or to its base or emitter as the case may be), and 
the constant frequency of the oscillator signal will be shifted in phase in proportion 
to changes in the amplitude of the audio signal, and at a rate dependent on its 
frequency. 

After modulation in this way, the carrier is passed through a normal series of r.f. 
and power amplifiers on its way to the aerial. 

The need for the audio correction network shown in the diagram between the a.f. 
amplifiers and the phase modulator was explained on Basic Electronics, page 6.19. 
The tendency in the phase-modulated FM transmitter is for the amount of frequency 
deviation in the modulated carrier to increase in proportion, not only to the amplitude 
of the audio signal but also to its frequency — with appreciable distortion of the 
modulation of the carrier arising in consequence. 

The basic method of eliminating this danger is to decrease the amplitude of the 
audio signal in proportion to any rise in its frequency, by passing the signal through 
a simple RC voltage-divider network of the type described and illustrated on the 
page quoted. 



§7] 



1.107 



The TV Vision Transmitter — Impedance Matching 

You last left the vision signal, you remember, some five pages back when it was 
emerging from the Power Amplifier stage. The illustration shows the shape of the 
signal at this point, in both the 405-line and 625-line systems. 

Also shown on the illustration are the relative amplitudes of the various signal 
levels which you studied when you were reading about the picture-sync ratio on 
page 1.76. 



The Structure of the VISION SIGNALS 




Peak White 

Level 100% 

Black Level 

.^37-39% 

Blanking 

Level 30% 

Sync Level 

0% "% 




-—Sync Level 100% 
-—Black Level 76% 
(also Blanking Level) 



Peak White, 
Level l»% 



«... . . «. * (PosmvE 
405-Line System modulation) 



- ,», ^ «oe ■• •• <. (NEGATIVE 

British 625-Line System modulation) 



The vision signal is now a very powerful one, and the physical difficulties of handling 
it are considerable. 

A typical PA stage, for example, may consist of a pair of triodes connected either 
in "push-pull" or in parallel. The voltage supplied to the anodes of these valves 
may well be of the order of 6000 volts, and very large currents are thereby caused to 
flow. The heat generated in the triodes by the resulting high power dissipation 
(P = VxI) is great enough to call for measures of forced cooling. Some types of 
PA valve are partially encased in water-jackets and are cooled by re-circulated water, 
as in the engine of your car. Others are cooled by an air blast, the resulting warm 
air being sometimes used to heat the transmitter building. 

When it leaves the PA, the vision signal passes through an impedance-matching 
stage, the purpose of which you will understand if you look back at §5 of Basic 
Electronics, Part 4. You there learnt that, if maximum transfer of power is to be 
achieved from a transmitter through a feeder to an aerial, and then radiated by that 
aerial, it is important that the output impedance of the transmitter, the characteristic 
impedance (Z ) of the feeder, and the input impedance of the aerial should all be 
closely matched. 

Now the output impedance of the vision transmitter ("looking back into it", as it 
were) is typically not less than 4000 ohms. The input impedance of a dipole aerial, 
as you will learn later in this Section, is affected by a number of factors, but is always 
very much lower — typically of the order of 50-80 O. Feeders can be manufactured 
with widely varying characteristic impedances ; but those with a Z of around 50 to 80 
ohms have considerable practical advantages. 

It is therefore common practice to pass the vision signal through an impedance- 
matching stage capable of lowering the Z to a value of around 50-80 £2. The signal 
is then taken to the aerial along a feeder with a Z of that approximate value. 

At the lower VHF frequencies, impedance-matching is carried out easily enough 
with the aid of a transformer whose secondary is tapped in such a way that the number 
of turns in it can be varied at will. 



1.108 [§7 

Impedance Matching (continued) 

The impedance of the primary of a transformer (Z p ) is related to the impedance of 
its secondary (Z s ) by the expression:- Z p = N 2 Z S , where N is the turns ratio of the 
transformer. 

Thus in order to match a transmitter impedence of 4000 O into a feeder impedance 
of 75 Q, the turns ratio of the matching transformer would need to be calculated as 
follows: ,- z - ™- 

N = ^/— =/y/^ r =^53 = about 7-3. 

Thus the matching transformer should have a turns ratio of about 7-3:1 in order to 
achieve the desired impedance match. Note, though, that in order to handle the 
power and high voltages involved in a big TV transmitter, the matching transformer 
will need to be a very sizeable affair. 

At UHF, the ordinary type of transformer cannot be used— because (for one 
reason) capacitive losses between the windings become too high. It is therefore 
usual for the output of the transmitter to be coupled to the feeder by means of specially- 
made lengths of transmission line called lecher lines. 

Lecher lines are hollow metal tubes, typically between 10 and 20 millimetres in 
diameter, whose surfaces are silver-plated in order to minimize resistance to current 
flow at very high frequencies. (You will learn more about this in Part 2, when you 
meet a phenomenon called skin effect in the UHF receiver.) 

A lecher (pronounced lekka) line of appropriate length is similar to a quarter-wave 
stub, except that it is of better quality so as to keep power losses at very high frequen- 
cies to a minimum. When one such line is placed close to another one, which itself 
forms part of a tuned circuit, a voltage is induced in the second line by normal 
transformer action. If this second, or coupling, lecher is made exactly a quarter of a 
wavelength long and is shorted at one end, its impedance will vary from zero at the 
shorted end to a very high value at the open end. 

Somewhere along the length of the coupling lecher there will therefore exist a value 
of impedance suitable for matching into the feeder. Adjustable "fingers" connected 
to the two conductors of the feeder are moved along the length of the coupling lecher 
until the desired Z is found. 

IMPtDANCB MATCHING by LBCHBR UNi 



Coupling Lecher 



H.T(+) 



To Aerial . 



Aerial Feeder Line 



c 



Adjustable Fingers : 



4TD 



P.A. Output 
Valves 



*€LD 



Tuned Circuit Lecher 



Note that lecher lines are used at the higher VHF frequencies in Band 3, as well 
as at UHF. It is simply a matter of which works best — lecher line or transformer. 



§7] 1.109 

The Sideband Filter 

You will remember from what you learnt in Part 4 of Basic Electronics that when- 
ever a low-frequency signal is used to modulate the amplitude of a high-frequency 
carrier, two additional frequencies are produced which are called sidebands. These 
two frequencies lie respectively above and below the frequency of the carrier; and are 
distant from that frequency by equal amounts. 

The amount of this divergence from the carrier frequency depends on the frequency 
of the modulating signal. When, for example, a 200 kHz carrier is modulated by 
a 5 kHz signal, one sideband frequency (known as the upper sideband) will be 5 kHz 
above the carrier, and the other (the lower sideband) will be 5 kHz below it. 

There are, therefore, three separate frequencies present in the modulated carrier — 
the carrier frequency itself of 200 kHz; the upper sideband frequency of 205 kHz; 
and the lower sideband frequency of 195 kHz. Note that the two sidebands are 
separated from one another by double the frequency of the modulating signal. 

You know that the highest and lowest frequencies present in the modulated carrier 
are those corresponding to the highest and lowest frequencies present in the modulating 
signal. If, then, the intelligence contained in the transmitted carrier is to be faithfully 
reproduced at the receiver, the bandwidth of the receiver must be sufficiently wide to 
accept the full range of frequencies produced during modulation. 

There is no great problem with the comparatively low-frequency signal which is 
used to modulate the sound carrier; for doubling the frequency of this signal produces 
no very wide total bandwidth. But with the much-higher-frequency signal used to 
modulate the vision carrier, the two sideband frequencies produced are much farther 
apart. In the 405-line system, both frequencies extend to at least 2-5 MHz — which 
means that if the full range of modulation frequencies are to be transmitted, the total 
bandwidth required for satisfactory reception will be at least twice 2-5 MHz. 

Add to this total vision-carrier bandwidth of not less than 5 MHz the full band- 
width required by the modulated sound carrier, and you get an overall transmission 
bandwidth which is very high. The original London TV transmitter at Alexandra 
Palace, indeed, used to transmit a vision carrier of 45 MHz, plus both its sidebands. 
Overall width of the vision carrier was therefore the wide one of 45 ± 5 MHz. 

Double-sideband transmission on the vision carrier clearly restricts the number of 
channels which can be fitted into a given frequency band. A technique has been 
worked out, however, which allows the 
greater part of either one of the side- 
bands to be suppressed without impair- 
ing the quality of the picture reproduced 
at the receiver — provided that the 
receiver has been appropriately de- 
signed. The overall bandwidth required 
by the modulated vision carrier is in this 
way much reduced. 

This type of transmission is now used 
by TV systems all over the world. It is 
known as vestigial sideband trans- 
mission, by reason of the fact that only 
a "vestige" of one of the two side-bands 
is transmitted. 




1.110 



[§7 



The Side-band Filter (continued) 

The suppression of one of the two vision sidebands is carried out at the transmitter. 
It is done in one of two ways — either by deliberate mistuning of certain tuned circuits, 
or by the insertion of a sideband filter circuit in which one sideband is filtered off and 
caused to dissipate its power in an absorber resistance. Filters are usually lengths 
of concentric cable connected across the signal feeder and terminated in the resistance. 

In the 405-line system it is the upper sideband which is restricted, the lower sideband 
being transmitted in full. The diagram below shows (on the left) how the sound 
and vision carriers are situated in the frequency spectrum, and where the sideband is 
is reduced. Note that the upper sideband of the vision carrier is transmitted in full 
up to 0-75 MHz, but rapidly attenuated thereafter until it is completely suppressed at 
1*25 MHz. The lower sideband, on the other hand, is transmitted at full amplititude 
for approximately 3 MHz, and then reduced to zero so as not to interfere with the 
sound carrier being radiated at 3-5 MHz below the vision carrier. The two sidebands 
associated with this sound carrier are so small (less than 20 kHz each) that they cannot 
be shown on the diagram without exaggerating them right out of scale. 

You will see that the upper and lower extremities of the channel are plus 1-25 MHz 
and minus 3-75 MHz relative to the frequency of the carrier, thus making an overall 
channel width of 5 MHz. 



405-LINE SYSTEM 

Sound Vision 

Carrier Carrier 







British 625-LINE SYSTEM 

i' Vision 
Carrier 



Vestigial 
Sidebands 



4 35 3 2 I { I I 

j Frequency (MHz) .' 75 ■ 

h CHANNEL — H 

WIDTH 




H 



375 



5 MHz 



1-25 



1-75 



I 2 3 

(Frequency MHz) 

— CHANNEL 

WIDTH 
8 MHz 



5 | 6 
5-5 i 



625 



The technique of vestigial sideband transmission is also used in the British 625-line 
system; but there are nevertheless important technical differences between the two 
systems. The principal ways in which the 625-line system differs from the 405-line 
system can be summarized (in so far as their respective frequency spectra are con- 
cerned) as follows: 

U In the 625-line system, it is the lower sideband which is suppressed, not the upper. 

Q The overall channel width is 8 MHz, instead of 5 MHz. 

Q Within this overall channel width, the width of the unsuppressed vision side- 
band is 5-5 MHz, instead of 3 MHz. 

O The vestigial sideband is restricted to 1-25 MHz, not to 0-75 MHz. 
Q The sound carrier is radiated at a frequency 6 MHz above the vision carrier, 
instead of 3-5 MHz below it. 



§7] 



1.111 



Combining the Sound and Vision Signals 

The job of the combining unit is to feed the separate sound and vision signals 
into a common output feeder line in such a way that there is no risk of the vision 
signal getting back down the line into the sound transmitter or of the sound signal 
getting back down the line into the vision transmitter. 

There are several ways of combining the outputs of the two transmitters without 
this danger developing, one of the simplest being that shown in the diagram below. 




Shorting Element 



The emimm 



The outputs from the two transmitters are joined at Point A. At intervals equal 
to one-quarter of the wavelengths of the respective signals back along each line from 
this point, there are connected a number of pairs of quarter-wave (X/4) short-circuited 
stubs. You learnt about stubs of this kind on page 4.55 in Basic Electronics. In the 
diagram above, they are the vertical elements connected to the twin wires of the two 
feeders. Each stub is "shorted" by a horizontal element joining it to the end remote 
from the feeder of its stub pair. 

These stubs resonate at one frequency only — that at which the physical length of 
the stub corresponds to one-quarter of the wavelength of the signal. If, then, the 
lengths of the stubs on the sound and vision feeders are made exactly one-quarter 
of the wavelengths of the sound and vision signals respectively, these signals will 
encounter a very high impedance "looking into" every stub in turn and will therefore 
pass on their way to the common feeder unimpaired. 

But when a signal whose frequency differs significantly from the frequency to which 
a stub is resonant is applied across the stub, the impedance of that stub will drop; 
and the more the frequency of the applied signal differs from the stub's resonant 
frequency, the lower will this impedance fall. If the frequency difference is great 
enough, the impedance will fall to a very low value, and the stub will act virtually 
as a short-circuit. 



1.112 [§7 

Combining the Sound and Vision Signals (continued) 

You know that the frequencies of the sound and vision signals in a TV trans- 
mission always lie some MHz apart. Thus when a signal tries to intrude down the 
wrong feeder— say, for example, the vision signal down the sound line — it inevitably 
has a frequency different from that to which the stubs on the forbidden feeder are 
designed to resonate. The first stub encountered will therefore offer to the intruding 
signal a very low impedance. The signal will flow down the stub; and much of its 
energy will be dissipated in the virtual "short" which it offers to the signal. 

Moreover, you also know that a low impedance at one point across a line is auto- 
matically reflected as a very high impedance across the same line one-quarter of a 
wavelength back along it. Since the stubs are carefully spaced at quarter-wave 
intervals from Point A (in the illustration on the last page) the intruding signal, on 
meeting the first low-impedance stub, automatically sets up a very high impedance 
to any signal of its own frequency across the feeder back at Point A. 

It is rather like a band of escapers from a prison castle groping their way along 
a secret tunnel, but each unconsciously setting off a trip-wire which progressively 
closes the door into the tunnel every time it is tripped. Thus every extra man entering 
the tunnel makes it harder for a companion to follow him through the steadily 
narrowing entrance. 

The reduced current getting past the first stub flows on up the forbidden feeder 
until it encounters the second stub — whereupon an identical process of "tripwire" and 
"trapdoor" is repeated, with the very high impedance at Point A now reinforced by a 
second high impedance set up across the line at the first stub by the action of the 
"tripwire" at Stub 2. 

Both the sound and (especially) the vision signals have been raised to very con- 
siderable power by their passage through the Power Amplifier stage. So a succession 
of stubs (seldom less than three) may be needed before the last of the energy of an 
intruding signal is dissipated in the virtual "short" offered by the last stub to which 
it manages to struggle through the successive barriers set up in the line against it. 

Meanwhile, back at Point A the common feeder to the aerial, which has no "signal- 
trapping" stubs attached to it, offers a path of very low impedance to sound and 
vision signal alike. Both therefore prefer to take this easy path rather than try to 
force their way along the feeder respectively forbidden to each. 




To Aerial 



High Z caused by 
irst Stub across the 
Vision Feeder 



To Aeriai 



HighZ 
caused by Fi 
Stub across the 
Sound Feeder 




Path of the 
Vision Signal 



The feeders in the diagram of the combining unit on the preceding page were shown 
as being of the open-wire type — largely for convenience of illustration. But the 
same principle can be applied with feeders of the concentric type. The stubs then 
consist of X/4 lengths of cable connected across the feeder, while across the other 
end of the stub the two conducting elements of the concentric cable are connected 
together to form the short-circuit required. 



§7] 1.113 

Aerial Feeders 

You learnt a good deal about transmission lines, or feeders, in Part 4 of Basic 
Electronics. The type of feeder most commonly used in TV transmitting stations is 
a semi-flexible concentric cable similar to that described and illustrated on page 
4.57 of that Series. 



CONCENTRIC 
LINE 




Outer conductor 



Inner conductor 



The diameter of the cable will range from about an inch up to six inches or so, 
normally increasing in proportion to the amount of power which the feeder is required 
to carry. The thicker types of cable would be used to feed the higher-powered 
transmitters radiating from a hundred kilowatts up to as much as 1000 kW of power. 

A particular advantage of the concentric type of cable is that its cylindrical outer 
conductor can be earthed in order to screen the inner conductor against electrical 
losses caused by radiation from the line itself. This is an important consideration, 
for the distance between transmitter and aerial is seldom less than 1000 feet (or well 
over 300 yards). Electrical loss from a cable of that length could dissipate an impor- 
tant fraction of the power being transmitted to the aerial, if screening was less than 
adequate. 

Another possible source of electrical loss is the accumulation of moisture caused 
by condensation in the dielectric spacing between the two conductors. At worst, 
this could cause a full short-circuit between the two. Some feeders have warm air 
ducted under pressure through them (i.e., between the two conductors), in order to 
counteract this danger. 

Transmission lines of this type can be constructed with varying characteristic 
impedances, generally somewhere between 50 and 150 ohms, depending on the input 
impedance of the aerial. This impedance, as you will see, depends on a number of 
factors; but, whatever its value, it is normally the impedance of the aerial which 
determines the Z specified for the transmission line, and therefore the impedance to 
which the matching unit must lower the output impedance of the transmitter itself. 

Feeders with a characteristic impedance much higher than the 50-150 Q range 
mentioned above can be built, and are still in use in some of the earlier transmitting 
stations ; but they need to be of the open two-wire type pictured on Basic Electronics, 
page 4.57. 



TWO WIRE 
OPEN LINE 




Insulated spacer ' Parallel wires 



Feeders of this type, however, possess the disadvantage of being unscreened. They 
therefore tend to lose power by radiation. It is also difficult to maintain the exact 
spacing of the wires along the line in rough weather, with the result that their impe- 
dance fluctuates. Mismatching results, and power loss occurs. 



1.114 [§7 

The Aerial 

The function of any transmitting aerial — whether it is being used for radio, for 
television, for radar, or even for trying to bounce back a signal through space from 
Mars — is to convert the variations in current or voltage of the modulated signal into 
an electromagnetic wave, and to radiate this wave in the desired direction (or in all 
directions at once). 

You learnt in Part 4 of Basic Electronics that a type of aerial much used in trans- 
mitters is the Hertz aerial, or half-wave dipole. The frequencies used in TV are 
especially suitable for the half-wave dipole, and its employment in TV transmitting 
stations is general. 

Refer back at this point to pages 4.59 to 4.70 of Basic Electronics, where the basic 
principles of how all aerials work are explained. You learnt there, in particular, 
that a half-wave dipole behaves like a series-resonant tuned circuit. It is as if its two 
halves were the last bits of an open-ended transmission line bent back at right angles 
at a point exactly one-quarter of a wavelength back from the open end. You also 
learnt that, when a.c. flows in a half-wave dipole, the voltages at all corresponding 
points on the two halves of the dipoles are at all times equal in amplitude but opposite 
in polarity. 

There therefore exists between the two halves of the dipole when current is flowing 
in it an electric field, having a direction towards the more positive charge, which is 
similar in nature to the electric field existing between the plates of a capacitor. When 
a large current flows in the dipole, this field will be strong. 

When the current changes direction, the voltages in the two halves of the dipole 
change polarity, and the direction of the field changes with them. The field therefore 
collapses very rapidly, and equally rapidly builds up again in the opposite direction. 
The general pattern of the field is shown in the illustration below; but remember that 
the plus and minus signs are constantly changing position so that the direction of the 
arrows is constantly reversing as the field builds up, collapses, and builds up again 
with opposite polarity. 



eiECTRI Ft Bid 
Stt r round/ n $ 
an Aerial 




Remember that the electric field of a dipole is capacitive in nature ; and that the 
plane of the field is the same as that of the two halves of the dipole. 



[§7 1.115 

The Aerial (continued) 

Besides the electric field described on the last page, the a.c. flowing in the dipole 
gives rise also to the type of magnetic field which always builds up round a conductor 
in which current is flowing. When the current flow in the dipole reverses, this 
inductive field collapses and builds up again rapidly with opposite polarity. Again 
you have the alternating effect — but this time in a plane at right angles to the direction 
of current flow, and therefore at right angles to the dipole. The best way to picture 
it is to imagine that you are poised vertically above the tip of the aerial mast and 
looking down on it. The magnetic field radiates from the aerial exactly as would 
ripples if you dropped a stone into a pool of water from this position. 



Tfie 
MAGNETIC N$U> 
Surrounding 
4* Aerfal 



The electric and magnetic fields around the dipole are therefore at right angles to 
one another. They alternate about the dipole aerial — building up to a peak, collap- 
sing and building up to another peak in the opposite direction — at the same frequency 
as that of the current flowing in the aerial, and varying slightly (in amplitude, in 
frequency or in both) according to the variations imposed on the carrier by the audio 
and video modulating signals. 

How the Electromagnetic Waves Are Radiated 

You have seen that the capacitive-type electric field is caused by voltage differences 
in the dipole, and the inductive-type magnetic field by variations in the flow of 
current. You know that current and voltage in a tuned circuit are out of phase with 
one another. When circuit resistance is low (as it is in a well-designed dipole), they 
are almost 90° out of phase. Thus the electric and magnetic fields round a dipole 
expand and collapse out of phase also. When one is at its peak, the other has almost 
collapsed. It is as if at one moment all the energy fed into the dipole were stored in 
the capacitances and at the next moment in the inductances. The reinforced alterna- 
ting effect goes on for as long as a.c. flows in the dipole. 

The power delivered to a modern transmitting aerial is very considerable — several 
hundreds of kilowatts would not be unusual. So the fields generated are strong. 
As. these strong fields repeatedly and rapidly build up and collapse, portions of both 
escape from the aerial altogether and are radiated into space as electromagnetic 
waves carrying the transmitted intelligence to distant receivers. 



1.116 [§7 

The Half-Wave Dipole 

You know from Basic Electronics, Part 4, that an effective half- wave dipole is formed 
when the wires of an open-ended transmission line are bent back at right angles at a 
point one-quarter of a wavelength back from the open end. With the conductors 
opened out in this way, the fields surrounding them can be radiated. When they were 
close-spaced, the fields were mutually cancelled out. 

In practice, however, the aerial is seldom formed by bending back the wires of a 
line. It almost always consists of either a hollow or a flattened rod of the correct 
electrical length. Such a rod has important advantages over a pair of wires. It 
possesses rigidity; it stands up better to bad weather; it can handle much more 
power; and (as you will shortly see) by increasing the diameter of the rod relative 
to its length, the bandwidth transmitted by the aerial can be usefully broadened. 

Nevertheless, the simile of the bent-back transmission line is useful, for it points 
up some essential facts about any half-wave dipole aerial. 

First, the length of each arm of the dipole must be almost exactly one-quarter of 
the wavelength of the signal which the aerial is designed to radiate (for the "almost", 
see later). This means that a half-wave dipole of given length can radiate with full 
efficiency only a signal whose wavelength is four times the length of one of the dipole 
arms (or twice the dipole's own overall length). If you want to radiate a signal 
having a different frequency (and therefore a different wavelength), you have to use 
a dipole of different length. 

The illustration below will remind you of the reason for this critical quarter- 
wavelength dimension of each arm of the half-wave dipole. Observe the standing 
waves of current and voltage which exist at a point one-quarter of a wavelength (X/4) 
back from the open end of the transmission line — voltage minimum, current maximum. 



Standing 
Waves 

on 
OPEN 
CIRCUITED 
LINE 




OPEN 



mm 






Note that the same conditions of maximum current and minimum voltage also 
occur three-quarters of a wavelength back from the open end of the transmission line. 
If you bent back the line at this point, you would get a dipole with the same properties 
of an LC tuned circuit resonating to a signal of the same wavelength as before. But its 
length would be unnecessarily great, and its propagation characteristics different. 

You may say, therefore, that the half-wave dipole represents the shortest length of 
aerial in which the conductor arms behave as a tuned circuit resonant at the desired 
frequency. 



§7] 1.117 

The Half- Wave Dipole (continued) 

The distribution of current and voltage along a half-wave dipole at resonance is 
now easy to follow. Current was maximum and voltage minimum at the point 
where the transmission line was notionally bent back to form the aerial. Current 
will therefore be maximum and voltage minimum at the centre of the dipole, and 
vice versa at its two ends. 

If you now apply Ohm's Law, in the form Z = V over I, to the voltage and current 
at various points along the dipole, you will find that the impedance "looking into" 
each end of the dipole is very high. Theoretically, it is infinite; for current, having 
nowhere to go to, cannot flow there. In practice, a little current does flow, and the 
impedance at the ends of a dipole is typically of the order of 3000 ohms. 

In the centre of the dipole, on the other hand, voltage is theoretically zero, so 
impedance should be zero also. In practice, most TV transmitter dipoles have a 
centre impedance of the order of 50 to 80 ohms (though, as you will see later in this 
Section, there are a number of factors which affect the Z of a dipole in different ways). 

You now see the importance of those matching transformers back in the sound and 
vision transmitters. The output impedance of the transmitters has a value which 
cannot be altered, and which is typically of the order of 4000 ohms. The job of 
matching transformers is to see that maximum power is transferred from a generator 
with an output Z of that order of magnitude into a load having a Z IN of the order of 
50-80 ohms. 

But, you may say, I know that the ends of a dipole have an input impedance of 
some 3000 ohms. Surely no grave mismatch would occur if the transmitter was 
connected across the ends of the dipole, using a feeder having a Z of between 3000 
and 4000 ohms ? Dipoles of this kind are indeed in use (though they are usually 
fed at one end only and a special form of matching is employed to take the place of 
the other terminal). The drawback to them is that a feeder with a Z of 3000-4000 
ohms needs to be of the unshielded, twin-wire type, with wide spacing between the 
wires. There are considerable difficulties in maintaining this spacing uniform through- 
out the length of the feeder, especially in bad weather or in high winds ; and there is 
also a risk that the open wires will themselves start acting as aerials and will so dissi- 
pate some of the power they are supposed to be carrying to the aerial. 

For this reason, transmitters typically use fully-screened and weatherproof con- 
centric feeders of about 50 to 80 ohms impedance. At the dipole, the central con- 
ductor of the feeder is connected to one half of the dipole and the outer conductor 
to the other half. 

A dipole so fed is described as current-fed, because the current flow in the dipole 
is maximum at the point of feed. An end-fed dipole is said to be voltage-fed. 

The Half-wive 

Centre £/>*/* 

Feed Point r 



Transmission Line 
to Transmitter 




1.118 [§ 7 

The Aerial — Polarization 

It is customary to describe an electromagnetic wave by reference to the inclination 
of its electric field. In the diagram below, this inclination is vertical. The radiated 
wave is therefore said to be vertically polarized. 



\ \ > 



N 
\ 

\ \ 
\ \ 

-. \ \ 

\ S V 

I \ I 

> I ! 

1 1 ' 

/ I I 

i I I 

' I ' 
/ I 



The B ^magnetic Wayes 
Radiated by this Dipole are 
V£l *lty POLARfStD 



If the aerial pictured above were to be turned through 90° from the vertical to the 
horizontal, both fields would likewise move through 90°. The electric field would 
become horizontal, and the resulting electromagnetic wave would be said to be 
horizontally polarized. 

It is usual, in both radio and TV, for the transmitting aerial to be mounted either 
vertically or horizontally — seldom at any angle between. Radio and TV signals 
thus normally travel with a polarization either at right angles to the surface of the 
Earth, or parallel to it. 

It is the practice in Great Britain to transmit most (but not all) programmes in the 
VHF Bands 1-3 with vertical polarization. All UHF programmes (save re-trans- 
missions from low-power "fill-in" stations) are to be transmitted with horizontal 
polarization. 

The efficiency of a receiving aerial is greatly affected by its positioning with regard 
to the polarization of the signals to which it is required to respond. Its efficiency 
will only be maximum if its orientation corresponds to the polarization of the signal 
being received. In other words, an aerial must always be mounted vertically for 
best reception of a vertically-polarized signal, and horizontally for best reception of a 
horizontally-polarized signal. 

This polarization-sensing property of a receiving aerial can greatly help in prevent- 
ing mutual interference between two stations transmitting on the same frequency. In 
Channel 1, for instance, the Crystal Palace (London) transmitter radiates a very 
powerful signal on 45 MHz for vision and 41-5 MHz for sound. West Cornwall is 
served by a separate transmitter operating on the same frequencies. Despite the 
intervening distance, mutual interference would be possible, for reasons which you will 
see later on; but it is prevented by transmitting the London signal with vertical polar- 
ization and the West Cornwall signal with horizontal polarization. 

It is for this reason that the familiar "H" and "X"-shaped aerials on which Bands 
1 and 2 are commonly received are mounted vertically in and around London; whereas 
in Cornwall they are mounted horizontally, on a plane parallel to that of the earth. 



§7] 



1.119 



The Aerial — Dimensions 

You know that the same television aerial is commonly used to radiate both sound 
and vision signals, and that the frequencies of these two signals differ by as much as 
3-5 MHz in the 405-line system and 6 MHz in the 625-line system. You also know 
that an aerial cannot radiate a signal of any frequency with full efficiency unless it is 
resonant to that frequency. How can a single aerial be simultaneously resonant to 
two frequencies several megacycles apart? 



RESPONSE CURVES 

of the THIN and THICK DIPOLES 




The answer is, of course, that it can't 
be; and that the frequency to which the 
transmitting aerial is made to resonate 
has to be a compromise. The solution 
is to make the aerial resonant to a mean 
frequency lying between the sound and 
vision signal frequencies, and to make 
the response curve flatter and wide 
enough to embrace both signals. 

This can be done by increasing the dia- 
meter of the dipole relative to its length. 
In this way, selectivity is deliberately sac- 
rificed in favour of a response which is 
adequate at both sound and vision frequ- 
encies, but less than optimum for either. 

Another drawback of thickening the 
dipole is that its centre-point impedance 
may fall low enough to cause added 
problems of matching. 

The compromise frequency normally chosen is the geometric mean between the 
vision signal frequency (call it f,) and the sound frequency (fj). The frequency to 
which the aerial is tuned (f r ) is thus calculated by the equation: f r = -y/f , x f 2 . 

Take as an example the aerial used for radiating BBC 1 signals from the trans- 
mitter at the Crystal Palace. 

The frequency of the vision signal radiated by this aerial is 45 MHz, and that of 
the sound signal 41-5 MHz. The aerial must therefore have a frequency response 
which is nearly flat over the 3-5 MHz which separate the two signals, so that its 
efficiency may be approximately the same for both. What "wavelength" is to be 
taken, in these conditions, for calculating the proper length of the dipole? 

The geometric mean of the frequencies 41-5 MHz and 45 MHz is v/41 .5 X 45, which 

works out at about 43-21 MHz. The wavelength corresponding to this frequency is 

3 x 10 8 
— — — — (see Basic Electronics, page 4-45), or about 7 metres. Taking half this 

wavelength, the theoretical length of the dipole should therefore be about 3-5 metres. 
But because signals of all frequencies travel somewhat slower through the metal 
of which a dipole is composed than they do through space, the physical length of a 
practical dipole is always made slightly shorter than that which would be indicated 
by theoretical calculation (so that a "half-wave dipole" is in practice more like a 
"0-48X dipole"). 



1.120 



[§7 



The Aerial — Directivity 

When an aerial radiates electromagnetic waves, the radiation will always be stronger 
in some directions than in others. Aerials are said to be directional along their lines 
of strongest radiation. In the case of the half-wave dipole, maximum radiation 
occurs in the plane at right angles to the dipole passing through the point in it at 
which maximum current flows — i.e., through its centre. 

Radiation patterns for half-wave dipoles mounted horizontally and vertically are 
illustrated below. Both aerials are shown, theoretically, as being mounted in free 
space. In practice they will be mounted close enough to the ground for their radia- 
tion patterns to be appreciably distorted from the ideal patterns shown. 



ylVmi'illli'il 



iRadiationl 



TlWIff^IHiE 



RADIATION PATTERNS 
of Half-Wave Dipoles 



s 



*«#* 

\fSfi 




Minimum 
Radiation 

♦ 



You will see from this diagram that the directivity of the vertical dipole aerial is 
predominantly horizontal— and the directivity of the horizontal dipole predominantly 
vertical. This is because of the shape of their respective polar diagrams. (The 
polar diagram of a transmitting aerial may be thought of as a contour line of equal 
field strength (seepage 1.124), all round the aerial, showing the relative effectiveness of 
transmission in all directions from it.) 

The reason why the polar diagrams are of this shape lies in the distribution of 
current flow along the dipole. Very little current flows at either end of the dipole, so 
very little radiation occurs from these ends and minimum signal is transmitted out 
along the line of the axis of the dipole. Maximum current, on the other hand, flows 
near the centre of the dipole, and maximum signal will be observed radiating in all 
directions from this point by an observer "looking into" the centre of the dipole from 
any position at right angles to its axis. 

Note that the horizontal directivity of an aerial system (in other words, its ability 
to maximize signal strength in a given horizontal direction) can be increased by 
stacking two or more radiating dipoles one above the other. The effect is to flatten 
the horizontal radiation pattern and to concentrate more of the radiated energy at the 
bottom of the beam. 



§7] 1.121 

The Aerial — Directivity (continued) 

When it is desired to concentrate the radiated signal in a particular direction 
(e.g., predominantly North, as with the BBC and ITA transmitters in the Isle of 
Wight), special elements called reflectors and directors are placed behind and in front 
of every dipole. These greatly reduce radiation behind the dipoles and substantially 
increase radiation in front of them. You will learn more about reflectors and direc- 
tors in the Section on receiving aerials in Part 2. 

Note, however, that the presence of reflectors and directors reduces the input Z 
of the dipoles. It is therefore another of the factors affecting the compromise which is 
always involved when the output impedances of the sound and vision transmitters 
have to be matched to the Z of the aerial. 

The Aerial — Service Area 

The service area of a TV transmitting station is the geographical area over which 
reception of its radiated signal is possible. Given a "non-directional" transmitting 
aerial (that is, one which radiates at equal strength in all directions), the service area 
viewed from above is theoretically circular; and if it were not for hills, tall buildings 
and other obstructions would extend to the same limits in all directions. 

The radius of this theoretical circle is some 60 km (over 35 miles) in all directions 
from the aerial, with reception beyond this range generally unpredictable. The effec- 
tive range of a TV broadcast is thus much less than that of the normal radio transmis- 
sion. The reason lies in the much higher frequency of the TV sound and vision signals. 

Recall what you learnt about the theory of wave propagation on pages 4.67 onwards 
of Basic Electronics, and look in particular at the explanation of the frequency spec- 
trum on page 4.70. You will see that at TV frequencies (the lowest used in Britain 
is the 41-5 MHz available for the sound signal in Channel 1 of Band 1) neither ground 
waves nor sky waves are effectively usable, and that reliance must be placed on 
space- wave transmission. 

At VHF, and still more at UHF, radio waves travel in a virtually straight line; so 
that the maximum distance over which a signal can be radiated is theoretically limited 
by the curvature of the Earth to the furthermost points, in all directions, which could 
be seen from the top of the aerial by the keenest of eyes in perfect weather. The 
position of these points relative to the aerial form its optical horizon; and the radiation 
of signals from the aerial to this optical horizon is called line-of-sight transmission. 

TV aerials are almost invariably set up on high ground in order to increase their 
optical horizon. This gives rise to another kind of wave which reaches the receiving 
aerial by a slightly different route. Thus at TV frequencies the electromagnetic wave 
at the receiving aerial is made up of two components. The direct wave (or space 
wave proper) travels from the transmitting aerial straight to the receiving aerial. The 
reflected wave strikes the ground between the two aerials and is bounced off it on its way. 

The DIRECT and REFLECTED waves in Line-of-Sight Transmission 



Reflected Wave 




Receiving 
Aerial 



1.122 [§7 

The Aerial — Service Area (continued) 

You will gather from the illustration on the last page that the main factors governing 
the usefulness of the direct wave are the curvature of the Earth's surface, and the 
height of the transmitting aerial (and, to a lesser extent, of the receiving aerial) above 
ground. The relative strength of the reflected wave depends primarily on the angle 
of incidence (O in the illustration) which the wave makes with the surface of the 
Earth, on the nature of that surface at the point of reflection, and on the polarization 
of the wave itself. 

Remember that you are still assuming that the terrain of the service area is flat. 
If that assumption does not hold good (and it seldom does), steep hills, large buildings 
and extensive sheets of water tend to deflect and distort the reflected wave. You will 
learn more about the effects of these distorting factors in Part 2; but for the moment 
go on assuming that the service area terrain is flat and normally reflective, and con- 
sider the theoretical factors which govern the strength of the transmitted signal at 
differing points within the service area. 

Field Strength 

The strength of the transmitted signal at any given point within the service area is 
known as the field strength at that point. It is expressed in either millivolts-per- 
metre (mV/m) or microvolts-per-metre ((xV/m), this being the measure of the voltage 
induced by the signal in an aerial per metre of its effective electrical length. 

The first factor affecting field strength at a given point in the service area is the 
amount of power transmitted by the aerial. The higher the transmitted power, the 
greater (other things being equal) will obviously be the field strength at a given point in 
the service area. The rule is: 

Field Strength Varies as the Square toot 
of the Power of the Radiated Signal 

Thus the power of the transmitter would need to be increased by a factor of 16 to 
achieve a fourfold gain in field strength. 

The second factor affecting field strength is the wavelength of the transmitted 
signal. As the frequency of this signal rises, so the flux density of the electromagnetic 
waves (that is, the number of lines of electric and magnetic force per unit of area) 
rises also — with the result that a field-strength measuring device will pick up a stronger 
and stronger signal as the wavelength of the signal progressively shortens. 

Unfortunately, the exact opposite of this increase in signal strength will be recorded 
by a receiving aerial situated quite close to the measuring device. The reason is 
that, as signal wavelength shortens, so the physical length of the aerial must be corres- 
pondingly reduced so that it can continue to resonate to the higher-frequency signal. 

The result will be a steady reduction in the aperture of the dipole, as it is called. 
In other words, there will be less of it available to pick the signal out of the air. 
You can say, in fact, that 

effective Field Strength Varies Inversely 
as the Wavelength of the Radiated Signal 



§7] 1.123 

The Aerial — The Service Area (continued) 

The third factor affecting field strength at a given point in the service area is the 
distance of that point from the receiving aerial. The rule is that: 

field Strength Varies inversely as the Square 
of the Distance from the Receiving Aerial 

Thus field strength at a point some distance away from the aerial is theoretically 
only a quarter of that at another point lying half-way along a straight line joining the 
first point to the aerial. Distance from the aerial has only been doubled, but field 
strength is reduced by a factor of four. 

Normally, therefore, TV signals are strongest in areas close to the transmitter. Up 
to 15 to 25 miles away from it, the signal is generally strong enough to give a good and 
consistent picture with only a modest receiving aerial, and VHF frequencies in Bands 
1 and 3 will be adequately picked up by indoor aerials mounted on the receiver itself. 
An outdoor aerial, however, will usually be required for reception of the UHF signals 
in Bands 4 and 5. 

At distances beyond about 25 miles out to the optical horizon, signal strength is 
still adequate to give a good picture provided that a more efficient aerial is used. 

The outermost limits of the service area depend partly on the height of the aerial 
above the surrounding countryside (the higher it is, the more distant its optical 
horizon), and partly on the frequency of the radiated signal. At the comparatively 
low frequencies employed in Band 1, the signal is able to bend slightly over the optical 
horizon and to follow the curvature of the Earth for a short distance beyond, so. that 
even aerials situated below the horizon can sometimes receive the radiated signal. As 
the frequency of the signal increases, however, the rate at which its strength decays 
below the optical horizon rises fast. This is particularly true of the UHF signal, which 
tends to behave more like a beam of light and so will not readily bend round corners. 

This fringe area just beyond the optical horizon lies normally some 35 to 40 miles 
distant from the transmitting aerial. Reception within it is unpredictable. Elabor- 
ate outdoor aerial systems are always required, with double-banked arrays having 
very high gain being needed to produce an adequate picture. 

And yet, despite all the normally-accepted service area limits given above, the 
TV signals of BBC 1 have been known to be received with fair definition in Australia! 
What happens is that, at certain times of the day at certain seasons of the year and in 
certain conditions of weather, strongly ionized layers are formed in the atmosphere 
and move in a certain way around the globe. These ionized layers are capable of 
raising very considerably the critical frequency above which (see pages 4.68 and 4.69 
of Basic Electronics) sky waves radiated from an aerial fail to be refracted back to 
Earth. With a convenient disposition of such layers around the Earth, the VHF 
signals reaching Australia would have got there by being bounced several hundred 
times between ionized layers in the atmosphere and the surface of the Earth, risking 
with every bounce failure to strike a waiting layer with suitable electrical properties 
or a normally reflective area of the Earth's surface. 

In freak reception of this kind, there are always large areas between transmitter 
and receiver in which no signal at all is received. As you learnt in Basic Electronics 
these no-signal areas form the skip distance of the transmitter. 



1.124 



[§7 



The Aerial — Service Area (continued) 

The zone limits detailed on the last page are not necessarily true of every transmitter 
anywhere in the world. Service area limits are affected by many other factors, 
particularly by the nature of the ground within the area, by its contours, and by the 
presence or absence of large structures such as tall office blocks or full gasholders. 

The sketch map below, issued by the Engineering Information Department of the 
BBC, shows the service area of the BBC station at Manningtree (Essex), which 
transmits at VHF in Channel 4 on a vision frequency of 58-25 MHz and on a sound 
frequency of 6 1 -75 MHz. Polarization is horizontal. 

The thin unbroken contour lines show the average values of field strength in milli- 
volts per metre of aerial length, measured with the receiving aerial fixed 30 ft above 
ground level. The heavy contour line broken by dots marks the limit of a service 
area free from interference from other channels for 90 per cent of reception time. 



Ba/ton o 



Aldeburgh 




Hartest 

o 



Sudbury o 



Wakes 
O Colne 



Walton-on-the-Naze 



Clacton-on-Sea 



Maldon 



Kilometres 



The service area of a transmitter is determined by making measurements of signal 
strength at a large number of points in the surrounding area, and by plotting these 
measurements on a map. Points of equal strength are joined up, and a contour line 
similar to the equal-height contours marked on an Ordnance Survey map is the result. 
The receiving aerial is mounted on a vehicle having an adjustable mast. Signals are 
measured with the aerial raised to a uniform height— usually 9 m above ground level, 
which is an internationally accepted approximation to the height of a receiving aerial 
mounted on the roof of a two-storey house. 

The equal-signal-strength contour lines are known as field strength contours. 



§7] 1.125 

Operating Frequencies of TV Transmitters 

The frequencies used for the broadcasting of radio and television programmes 
in all countries are allocated by international agreement, so as to minimize the 
chances of mutual interference. All the frequencies allotted lie within certain 
ranges (or bands, as they are called) of the electromagnetic spectrum, every band 
being identified by a number. 

It must be remembered that the "traffic" which has somehow to be fitted into the 
air without avoidable mutual interference includes much besides the broadcast of 
news, instruction and entertainment to the general public. Channels are also 
provided for the Police; for the Armed Forces; for the Foreign Office and other 
Ministries of State; for Space Research; for the many users of Radar in its various 
forms; for the numerous and enthusiastic amateurs who glory in the name of "ham" ; 
and for several other classes of special user as well. 

As far as British TV is concerned, the frequency bands allocated to it are as follows: 

Band l,whichrangesfrom 41 to 68MHz(VHF) 
Band 3, „ „ „ 174 to 216 MHz (VHF) 
Band 4, „ „ „ 470 to 582 MHz (UHF) 
Band 5, „ „ „ 614 to 854 MHz (UHF) 

Band 1 frequencies are at present used exclusively by the BBC, and those in Band 
3 by the Independent Television Authority (ITA). Bands 4 and 5 are reserved for 
use by the BBC and (one day) the ITA for the broadcasting of programmes on 625 
lines and in colour. These allocations have been made — like all allocations of 
operating frequencies within a country — by purely British internal arrangement. 

Each of the four Bands is itself divided into a number of smaller bands of consecu- 
tive frequencies, which are called channels. Every channel is in turn identified by a 
number. These channels are used by individual transmitting stations operating within 
a given frequency band. The London (Crystal Palace) transmitter of the BBC, for 
instance, operates on Channel 1 in Band 1 ; while the London (Croydon) transmitter 
of the ITA operates on Channel 9 in Band 3. 

The maximum number of programmes which it is possible to transmit within a 
band of frequencies is determined by the number of channels into which that particular 
band can be divided. For technical reasons, the minimum spacing between channels 
in the 405 -line system is 5 MHz. This means that Band 1, with its bandwidth of 
(68—41=) 27 MHz, could theoretically accommodate six programmes. In fact, 
however, the extra-wide spacing between Channels 1 and 2 allows only five — but this 
is a special case. Band 3, with its bandwidth of (216— 174=) 42 MHz, can hold 
eight programmes at a time. 

The minimum spacing between channels in the British 625-line system is 8 MHz, 
which means that 14 programmes could be accommodated in Band 4, and 30 in 
Band 5. But the provision of nation-wide coverage of a particular programme without 
mutual interference between transmitters requires the use of more than one channel 
for a given transmission. 

The result is that the number of different programmes which can be simultaneously 
transmitted on a national scale within a particular band cannot always be calculated 
by simple reference to the theoretical channel accommodation. 



1.126 



t§7 



The Aerial — Physical Structure 

The illustration on this page gives an impression of the 750-foot high mast support- 
ing the aerial system at the BBC transmitting station at Holme Moss, situated high 
up in the Pennines on the borders of 
Yorkshire and Lancashire. Note the 
comparative size of the substantial one- 
storey buildings at the foot of the mast. 

A powerful inland transmitter such as 
this, situated high up and in the midst of 
large centres of population on almost all 
sides, will be of the non-directional type 
having an almost circular service area. 

Pictured overleaf in closer detail are the 
topmost sections of the mast, on which 
the radiating elements themselves are 
situated. 




The MRM MAST 
at Holme Moss 





§7] 1-127 

The Aerial — Physical Structure (continued) 

The uppermost part of the big mast at Holme Moss consists of two separate 
sections, both indicated in the illustration to the right. The structure at the very 
top is the aerial used for radiating TV programmes in Channel 2 of Band 1. The 
aerial consists of a two-tier stack of half-wave dipoles, mounted in the vertical plane 
one above the other. (xx»<mmmmmsm 



TheTV Aerial 



Note the two slender-looking wires 
attached to the mast just under the platform 
at the base of the TV aerial. There are 
actually four of these wires "anchoring" 
the mast at this level. They are in reality 
thick steel cables, each over 800 feet long. 
Their lower ends are fixed to huge concrete 
plugs let into the ground in a nearly-400 ft. 
radius from the foot of the mast. 

Other groups of steel cables help to secure 
the mast at three other levels down its 
length. Their purpose is to help the mast 
withstand the tremendous pressures exerted 
by wind on a structure of such a shape. 

The section of the mast immediately 
under the TV aerial forms the aerials used 
to radiate three sound-radio VHF/fm pro- 
grammes in Band 2, which form part of the 
national network of the BBC. The section 
looks rather like a number of the small 
corner-turrets found in mediaeval castles, 
set one on top of another without their 
pepperpot roofs! The openings which 
resemble the "arrow-slits" in these struc- 
tures are vertical slots, each of which acts 
as a half-wave dipole radiating in the 
horizontal plane. 

A still taller TV mast is that at Emley Moor, in Yorkshire, which carries (among 
other things) an aerial system for transmitting future VHF/fm radio broadcasts in 
Band 2, an aerial system for radiating TV signals in Band 3, and (at the very top) 
another aerial system for radiating TV signals at UHF in Bands 4 and 5. This 
mast is 1265 feet — nearly a quarter of a mile — high. 

For reasons which you will see on the next page, dipoles are often stacked in tiers, 
one set above the other. On the Emley Moor mast, the section housing the stacked 
dipoles is enclosed in a fibre-glass tube 12 ft. in diameter. The object is to weather- 
proof the dipoles, and to make the servicing of them easier and safer. The effect is 
to make the mast look completely cylindrical from top to bottom. 




1.128 



[§7 



The Aerial — Physical Structure (continued) 

It is frequently desirable to distort deliberately the polar diagram of an aerial 
system in order to beam the radiated signal with greater power in one direction than 
in another. This might be made necessary by the nature of the terrain or by the 
concentration of the population in the service area. It is often essential in the 
immediate vicinity of the mast itself, for beaming can sometimes be too efficient and 
can send the radiated energy right over the heads of viewers living close to the aerial. 

Distortion and downward tilting of the polar diagram can be achieved by stacking 
the dipoles in a special way, and by feeding them through different lengths of trans- 
mission line. This results in some of the dipoles receiving the signal fractionally 
later in phase than others, so altering the radiation pattern of the resulting electro- 
magnetic wave. 




TILTING the POLAR DIAGRAM 

The radiating dipoles of the Sutton Coldfield (Birmingham) transmitter of the 
BBC are mounted in two tiers of four dipoles apiece, facing North — South — East — 
West respectively round the 750-foot mast, and with the two tiers placed one above 
the other exactly one wavelength apart. The dipoles are made of galvanized steel 
strip, and have 7-5 kW heaters to prevent ice from forming on them in winter. (At 
Emley Moor, the dipoles are of the rectangular-hollow-tube wave-guide type, to 
minimize electrical loss at ultra-high frequencies.) 

The Aerial — Precautions Against Breakdown 

It is common practice nowadays to guard against the total breakdown of a trans- 
mitting station, affecting millions of viewers, by duplicating the entire transmitter 
set-up and by feeding the output of each pair of transmitters to a different half-section 
of the aerial system. Then if one transmitter or one part of the aerial system should 
fail, at least a fairly good signal would still be available from the other part. 

Say, for simplicity of explanation, that one tier of dipoles in a two-tier aerial 
system is made completely separate, electronically, from the other. Each tier is 
served by a separate vision transmitter and by a separate sound transmitter, the out- 
puts of which are combined in a separate combining unit and fed to the appropriate 
tier along separate transmission lines. In practice, a somewhat more complicated 
arrangement is required in order to balance the phase of the signals fed to the aerial 
system as a whole, and parts of one tier are "teamed up" with parts of the other. 
But the principle of duplicating the basic operations of the entire transmitting station 
is not affected. 

The dipoles themselves are so designed that, should one-half of the system fail, 
the other can have the full power of the four transmitters (two vision and two sound) 
switched to it, and can radiate the resulting electromagnetic waves without any help 
from its faulty partner. 



§7] 

REVIEW of the Transmitter and Aerial 



1.129 



The TRANSMITTING: 




The STUDIO 



The Transmitting Station consists of not 
less than two transmitters (at least one for 
vision and one for sound) connected by 
special feeders to an aerial. The function 
of the station is to convert the video and 
audio signals received by landline or 
SHF radio link from the studio into 
electromagnetic waves, and to radiate 
these waves to receiving aerials situated 
within the service area of the station. 

The Vision Transmitter produces a 
carrier wave of very stable frequency and 
amplitude. The amplitude of this carrier 
is then modulated by the video signal from 
the studio, and the resulting vision signal 
is raised to a high power in a number of 
power amplifiers. The signal then passes in turn through an impedance-matching stage, 
a sideband filter circuit and a combining unit on its way through several hundred feet of 
feeder to the aerial. 

The purpose of the Impedance-Matching 
stage is to reduce the output impedance of mpe9MCe MAT C«m by UCMK U*£ 
the transmitter to approximately the Z„ 
of the feeder. The value of this Z„ is 
normally chosen so that it matches the 
impedance of the aerial at the point where 
the signal is fed into it. Matching is 
normally done at VHF by means of a 
transformer, at UHF by means of lecher 
lines. 

Vestigial Sideband Transmission is a technique used to reduce the bandwidth required 
by the radiated vision signal, with the object of fitting a greater number of channels into 
a given band of frequencies. It is achieved in the sideband filter circuit by suppressing 
part of one of the two sidebands which are produced during modulation of the vision 
carrier. Both of the sidebands produced during modulation of the sound carrier are 
transmitted in full. 

The Sound Transmitter uses the signals produced by the microphones in the TV studio 
to modulate either the amplitude of its carrier (in the 405-line system) or the frequency 
of its carrier (in the British 625-line system). There are two main types of frequency 
modulation — the direct type in which the frequency of an ordinary LC-type oscillator is 
directly modulated and then stabilized in a later stage, and the indirect type in which a 
more stable type of crystal oscillator is (though with more difficulty than before) modu- 
lated in a later stage. 

In both types, the frequency of the sound carrier is made to vary in accordance with 
changes in the amplitude of the audio signal. 




fcarljt feeder Una 



Tuned Circuit Ucbar 



[§7 



1.130 

REVIEW of the Transmitter and Aerial (continued) 

The task of the Combining Unit is to 
feed the separate sound and vision signals 
into a common feeder line, so that they 
can be radiated together from a common 
aerial. It is necessary to ensure that the 
vision signal cannot get back down the line 
into the sound transmitter, nor the sound 
signal down the line into the vision trans- 
mitter. The job is done by means of 
quarter-wave short-circuited stubs situated 
at intervals of one-quarter of the wave- 
lengths of the respective signals back 
along each line from their point of function. 

Electromagnetic Waves are generated as a result of the rapid build-up and collapse 
of the electric and magnetic fields set up round an aerial when a powerful alternating 
current is fed into the aerial. The two fields are at right angles to one another, and the 
wave is said to be polarized by reference to the inclination of its electric field. 




The Half-Wave Dipole is the shortest 
length of aerial in which the conductor 
arms behave as a tuned circuit resonant 
at the frequency which it is desired to 
radiate. The signal is generally fed to 
its centre point. 




Tat H*lf-w<¥* 



Transmission Line 
to Transmitter 



Th» ONttCT Mtf RCFUCTED » 



m in LAw«f-$J|fct rrwMMJMtM 

DlmWM 



_^_ ""-J — ™ kritcnd Win 



When an aerial radiates electromagnetic waves, the radiation is always stronger in 
some directions than in others. Aerials are said to be directional along then- lines of 
strongest radiation. The directivity of an aerial system in a desired direction can be 
improved by such devices as stacking the dipoles one above the other, or by placing 
reflectors and directors respectively behind and in front of every dipole in the system. 

The Service Area of a TV transmitter 
theoretically corresponds with the limits of 
the optical horizon in all directions from 
the top of the aerial mast, but is in practice 
much affected by the nature of the terrain. 

Signals travel by line of sight transmission from the transmitting to the receiving aerials 
in two types of wave — the direct and the reflected waves. 

The Operating Frequencies used in British TV are as follows: BBC 1 and TTV broad- 
cast on 405 lines at VHF in Bands 1 (41—68 MHz) and 3 (174—216 MHz) respectively; 
BBC 2 broadcasts (and all future 625-line programmes) at UHF in Bands 4 and 5 (470— 
854 MHz). 

In the 405-line system, the frequency of the sound carrier is 3-5 MHz below that of 
the corresponding vision carrier. In the British 625-line system, it is 6 MHz above it. 



§8: SIGNAL BANDWIDTH 



1.131 



You will recall that the picture elements of a CRT scanning device are individually 
extremely small, and that the total number of them contained in a single picture image 
can run into many hundreds of thousands. The scanning beam of the camera tube 
has to convert all these hundreds of thousands of picture elements into equivalent 
electrical signals, and it has to do so within the repetition period of a single picture. 

Since this repetition period is as short as one-twenty-fifth of a second, it is clear 
that the rate at which picture elements need to be converted into electrical signals is 
enormously rapid. To take a practical example: If the image displayed on the target 
of a camera tube is composed of 400,000 elemental areas, and if the complete image is 
scanned once every twenty-fifth of a second, the rate at which conversion must take 
place is 400,000 x 25, or 10 million picture elements per second. You must now find 
out why this formidable conversion rate demands a much greater frequency bandwidth 
for the radiated vision signal than is needed for radiation of the sound signal. 

Recall what you learnt about amplitude modulation in Part 4 of Basic Electronics. 
You learnt, in particular, that when an r.f. carrier is amplitude-modulated, the effect 
is to add new frequencies to the transmitted signal in addition to the frequency of the 
carrier itself. These extra frequencies, you read, are called sidebands ; and it is they 
— not the carrier frequency itself — which contain the intelligence of the transmission. 
You also found that the frequency bandwidth of the modulated carrier (that is to say, 
the range of frequencies extending between the limits of the upper and lower sidebands) 
increases as the frequency of the modulating signal increases. 

The same effect occurs when the vision signal from a TV station is transmitted from 
an aerial — only in this case the sidebands developed are very much wider than those 
produced by a radio transmitter because of the much higher frequency. 

The actual width of the sidebands produced by a TV transmitter depends on the 
rapidity with which the elemental areas of the scene being televised are being converted 
into electrical signals. This, in turn, depends largely on the tonal composition of the 
scene. When this is, say, a quiet seascape on a dull day, a great many of the picture 
elements composing the scene will merge into one another in a kind of overall grey. 
From the point of view of the scanning beam this reduces their number, and so the 
rapidity with which they have to be converted into electrical signals. 

To put the point in more homely terms : You need a greater frequency bandwidth 
to televise the final of the men's 100 metres on a sunny day at the Olympic Games 
than you do to televise an angling competition off the pier at Southend in dull, 
overcast weather. 



- Carrier Frequency 




+ 

Bandwidth 



Frequency 



-Carrier Frequency 



WWk 



o + 

Bandwidth 



Frequency 



1.132 



[§• 



The Frequency Content of the Video Signal 

You know that the video signal sent to the transmitter for radiation is composed of 
two elements: (a) a sequence of regularly-spaced synchronizing pulses for line and 
(much less frequently) field scans, and (b) a picture signal of continuously varying 
amplitude. It is the task of the designers of a TV set to determine the minimum 
bandwidth which such a video signal will require, in the most exacting conditions of 
tonal contrast which the camera is likely to be asked to handle. 

There is no difficulty about the two sync pulses, which form distinct and regularly- 
recurring components in the frequency spectrum corresponding to the repetition rates 
of the line and field sync pulses. As you know, the repetition rate of the line sync pulses 
is determined by the number of pictures presented per second, divided by the number 
of lines composing a picture. The field sync pulse repetition rate is twice the number 
of times a complete picture is presented per second. In all European TV systems, 
this picture presentation rate is 25 times a second, so the field sync pulse repetition rate 
is 50 times a second. 

Neither type of sync pulse, therefore, causes any large change in the frequency 
spectrum; so neither causes the designer any difficulty in this respect. The only 
proviso is that the bandwidth selected must be wide enough to accept some of the 
harmonic frequencies contained in the rectangular sync pulses, in order to preserve 
their "squareness". For if the shape of the sync pulses is unduly distorted, accuracy 
of synchronization will be impaired. 

It is the frequency components of the picture signal which in practice determine the 
signal bandwidth which a TV system requires for efficient operation. 

You know that it is necessary, if you are to get good definition of the reproduced 
picture, that the bandwidth of the transmission shall be wide enough to accommodate 
the highest frequencies produced by the camera. It is therefore usual to determine 
the bandwidth needed by a TV system by reference to the most severe tonal condi- 
tions which the system could theoretically encounter in practice. 

Such conditions are represented by a scene composed entirely of rows of alternate 
black and white squares arranged in chessboard fashion, as shown in the illustration 
below. Each of these squares represents an elemental area, and is assumed to be of a 
size equal to the area of the scanning spot (which is very small indeed). 



As the scanning beam moves across the chessboard scene, a pulse of current will 
flow every time a white square is scanned, and a minimum no-signal current (or even 
no current at all) will flow when a black square is scanned. The shape of the current 
waveform produced when a line of the chessboard is scanned will therefore be: 

UNE OF CHESSBOARD 
Scanned 




WAVEFORM OF CURRENT I I 1 I 1 I II II j I 1 White 

Produced U--U.--U..-LLU- -LLL.SZ 

The illustration above assumes that the camera used produces a positive-going 
picture signal. For a camera producing a negative-going picture signal, the current 
waveform is simply shifted along one square to the right. 



§8] 



1.133 



The Frequency Content of the Video Signal (continued) 

You know from Basic Electronic Circuits, Part 1, that a square wave such as that 
at the foot of the preceding page is in reality a composite wave, made up of a funda- 
mental sine wave having the same period as the square wave, plus a large number of 
harmonics of the sine wave. The frequencies of these harmonics, you will recall, 
are always multiples of the frequency of the fundamental; and the more of them there 
are present in a waveform, the nearer will that waveform attain to "squareness". 

It follows that the square waveform theoretically produced by a scan of one line of 
the chessboard contains a large number of harmonics of the fundamental sine wave, 
and that to transmit all these harmonics would involve an extremely high modulating 
frequency. The bandwidth of a carrier modulated by such a high-frequency signal 
would need to be very large — so large as to cause considerable problems of channel 
allocation, and of design in both the transmitter and receiver. 

A perfect picture of the chessboard scene cannot, therefore, be economically trans- 
mitted by a normal TV system. Fortunately this does not matter so much as you 
might expect, for in ordinary everyday scenes contrasting tones as severe as those of 
the chessboard practically never occur. Something a good deal lower than theoretic- 
ally-perfect definition can therefore be accepted. 

It has, in fact, been found that a picture of acceptable quality can be transmitted 
if all (or nearly all) the harmonics of the chessboard-produced square wave are omitted, 
and if a bandwidth be chosen for the system which is wide enough to accommodate the 
fundamental frequency of the square wave only. 

This fundamental frequency is shown in the illustration below. 



■ LINE OF CHESSBOARD 
Scanned 




SQUARE WAVEFORM OF I f— 1 I - 1 r~ 1 |— 1 [— 1 rnGrey 

CURRENT Produced |_J |_J L_J I I I I T I | Black 

FUNDAMENTAL FREQUENCY 
OF SQUARE WAVEFORM 




7U 



of the SQUARE WAVE 



Given a camera producing a positive-going picture signal, every positive peak of 
this sine wave represents a white dot in the tonal composition of the scene, and every 
negative peak a black dot. If the picture signal is negative-going, these polarities are 
of course reversed. 

The problem of determining the maximum frequency which the picture signal is 
ever likely to produce (and therefore the minimum bandwidth which the video signal 
will require) thus resolves itself into a matter of calculating the fundamental fre- 
quency of the signal produced when the chessboard is scanned. 

You should now see how this can be done. 



1.134 



[§« 



Calculating Maximum Picture Signal Frequency 

The first step in determining the maximum picture signal frequency produced when 
the chessboard is scanned is to calculate the total number of picture elements contained 
in the scene. Obviously, one of the factors affecting this total is the number of 
scanning lines composing the picture, for this governs the number of elements which 
can be fitted into the height of the picture. Call this number N. 

Now recall that the ratio of the width of the observed picture to its height is known 
as its aspect ratio (A). Whatever the actual dimensions of the picture, its width will 
always be A times its height. If the height of the picture contains N contiguous 
elements, each square in shape, the number of elements in any one line across the 
width of the picture must be A xN. 

With the picture itself composed of N lines, the total number of elements in the 
complete picture must be N times the number of elements contained in any one line, 
orNx(AxN) = AN 2 . 

You will follow the basic reasoning of this quite clearly if you look at the simplified 
sketch of a TV picture below. 



With the Aspect Ratio 4:3, 
Total Number of Elements 
in the Picture is 

4^ 

3 




AN 2 = ~ x 36 = 48 



AxN 



Now the signal waveform produced by a scanning spot moving across an individual 
line of the chessboard scene is composed of two distinct regions, one representative 
of the white elemental areas and the other representative of the black areas. The 
instantaneous signals produced from the two regions will be of opposite polarity, 
the signal representing black lying below the mean level of grey, and the one represent- 
ing white lying above it. 

Since the black and white elements are adjacent to one another, the signal waveform 

produced from every line of the chessboard will consist of a repetitive sequence of 

alternate positive and negative signals. One cycle of the signal is therefore completed 

for every pair of adjacent elements — one black, one white; and the number of cycles 

of the signal waveform produced from the complete picture will be exactly half the 

AN 2 
number of elements contained in the picture. In other words, it will be . 

The number of cycles of the signal waveform produced during the scanning period 
of the complete chessboard scene represents the maximum picture signal frequency, 
expressed in terms of cycles per picture. You have only to multiply that number 
by the picture repetition rate (P, expressed as the number of pictures scanned per 
second), and you get what you are looking for — namely, the maximum picture signal 
frequency expressed in cycles per second. 

The complete formula is thus 

MAXIMUM PICTURE SIGNAL PAN 2 

FREQUENCY = "IT CydeS per SeC ° nd 



§8] 1.135 

Calculating Maximum Picture Signal Frequency (continued) 

The formula derived at the foot of the preceding page can be used to calculate the 
maximum picture signal frequency (f max ) of any TV system by substituting the appro- 
priate values for picture repetition rate, aspect ratio and number of scanning lines. 

In the 405-line system, the values are: P = 25, A = 4/3; N = 405. Therefore 

_ PAN 2 _ 4x25x405x405 

m " ~ 2 3x2 

= 2,733,750 cycles per second, or Hertz 

which is approximately equal to 2*7 MHz. 

In practice, there is room to increase the vision signal bandwith of the 405-line 
system to 3 MHz, which permits transmission of the maximum picture signal fre- 
quency, plus a small number of its harmonics as well. 

In the British 625-line system, on the other hand, in which the values are P = 25, 
A = 4/3; and N = 625 

_ PAN 2 _ 4x25x625x625 
W — 2 ~ 3x2 

= 6,510,417 Hertz 
which is approximately equal to 6*5 MHz. 

This is a bandwidth too great to be fitted conveniently in the channel available, 
so part of the f max is sacrificed and the width of the upper sideband (which carries 
the intelligence) is limited to 5*5 MHz. Some slight loss of definition results on the 
screen of the picture tube — rather as if you were to use a film of coarser grain in your 
holiday camera. 

In both the above calculations, you will note, two factors have been ignored. The 
first is that the number of lines which are in practice lost in the field blanking periods 
reduces N, which increases the value of A. The second is that the proportion of each 
line which is inactive during the line blanking periods reduces the value of A. The 
degree of error so introduced is on balance small and may (in these two cases) be ignored. 

Note also the importance which the technique of interlacing assumes in evaluating 

PAN 2 
the equation f max = — - — , and so establishing the bandwidth needed by a particular 

TV system. 

You know that, in the British and all other European TV systems, the presentation 
rate of the picture (which needs to be a minimum of 50 pictures a second if objection- 
able flicker is to be avoided) is in practice reduced to 25 pictures a second without 
harm to the received picture. Thus P (the picture repetition rate) is effectively halved 
by interlacing, with consequent reduction of the bandwidth required for the system. 

To take a simple example. If it were not for interlacing, every one of the lines 
making up the picture in the British 625-line system would need to be presented not 
less than 50 times a second if flicker were to be avoided. The equation would then 

bC: «■ 4x50x625x625 

f max = — Hz, or nearly 13 MHz 

instead of the 6-5 MHz which you know to be all that is actually needed. 

It is easy to see the important part which the interlacing technique has played in 
easing the problems of channel congestion. 



1.136 [§8 

REVIEW of Signal Bandwidth 

The bandwidth required for transmission of the vision signal is much greater than is 
that needed for transmission of the sound signal; and the higher the definition of the 
transmitted picture, the higher will be the bandwidth required. 

The reason is that the many hundreds of thousands of elemental areas contained in the 
picture have to be converted into equivalent electrical signals within the very short 
space of time which the beam takes to scan the scene. In other words, the conversion 
rate of the elemental areas is very high. A high conversion rate calls for high picture- 
signal frequency; and this, in turn, requires that the vision signal at the transmitter shall 
have a wide sideband. 




OWE Of CHESSBOARD 
Scanned 



SQUARE WAVEFORM OF I [—] p... f^l fTl ._ 1^1 ...r^W 



CURRENT Produced 



FUNDAMENTAL FREQUENCE 
OF SQUARE WAVEFORM 



LJ~lJ : T_FhFl_J-LFlf 



The maximum frequency content of the 
video signal is generated by a TV camera 
when it is set to scan a scene composed of 
the most severe tonal contrasts it is possible 
to devise. Such a scene is accepted to be 
a chessboard pattern of alternate black and 
white elemental areas, each square in 
shape and each the exact size of the cross- 
section of the scanning beam. 

The signal frequency so produced is here designated f„, ax . 

The formula for calculating the maximum frequency content of a TV system when it is 
scanning the chessboard scene is 



7U 



%**u«e^ ^W*"* 



of the SQUARE WAVE 




where P is the picture repetition rate (in number of pictures presented per second), A 
the aspect ratio of the observed picture, and N the number of lines composing the picture. 
Note that, because f max is proportional to the square of the number of lines used in a 
given TV system, any 625-line system will require a much greater bandwidth for satis- 
factory transmission than will a 405-line system. 

The absolute size of the f max of a TV system — and therefore of the bandwidth it will 
require for satisfactory reception — is much reduced by the technique of interlacing. If 
50 pictures per second need to be presented to the observer's eye to avoid the sensation of 
flicker, this condition can be satisfied by presenting instead 50 fields interlaced so as to 
form only 25 pictures. 



The value of P in the 



PAN 2 



equation is thereby halved, and with it the maximum 



bandwidth needed by the system. 



INDEX TO PART I 



Aerial, Transmitting 
Afterglow 
Aspect Ratio 

Back Porch 

— Projection 
Band, Frequency 
Bandwidth, Signal 
Black-Level Reference Signal 
Blanking Pulse 

— Waveform 
Bright-Up 
Burn-In, Image 



1.114 et seq., 1.126 

1.29 

1.36, 1.134 

1.70 

1.96 

1.125 

1.131 et seq. 

1.48 

1.45, 1.57 

1.69, 1.72 

1.31 

1.57 



Caesium 1.12, 1.38, 

Camera Chain 

— Control Unit 

— Tube 
Carrier Modulation 

— Wave 
Central Apparatus Room 

— ■ Control Room 
Channel, Frequency 
Close-up 

Colour (Nature of) 
Cutting 

D.C. Level 

Decelerator Grid 

Definition 

Defocusing 

Directivity 

Director 

Dissolve 

Dynode 

Electromagnetic Waves 
Electron Gun 

— Multiplier 
Equalizing Pulses 
Eye, Human (How it Works) 

Fade-in, fade-out 
False Perspective 
Feeders, Aerial 
Field 

— Flyback Path 

— Frequency 

— Strength 
Contour 

— Sync Pulse 

Final Anode (of Iconoscope) 
Final Transmission Picture 
Flicker 
Flyback Period 

— Time 
Focus Coil 
Frame 
Frequency Content (of Video Signal) 

— Deviation 



1.42, 1.52 

1.89 

1.92 

1.29, 1.37 

1.101 

1.100 

1.85 

1.82, 1.85 

1.125 

1.95 

1.8 

1.93 

1.48 
1.55 
1.16 
1.94 
1.120 
1.121 
1.94 
1.39 

1.115 

1.41 

1.39, 1.56 

1.78 

1.17 

1.93 
1.97 

1.113 

1.32 

1.72 

1.32 

\A22et seq. 

1.124 
1 .72 et seq. 
1.41 
1.87 
1.26 
1.30 
1.72 
1.52 
1.32 
1.132 et 
seq. 

1.104 



Frequency Multiplier Circuit 1.100 

Fringe Area 1.123 

Front Porch 1.70 

Fundamental Frequency 1.133 

Half-Line Pulses l .73 

Half-tone Photo-engraving 1.15 
Half-wave Dipole 1.114 e/ seq. 

Heat (Nature of) ] .8 

Iconoscope Camera 1 .40 et seq. 

Image Accelerator Grid 1.52 

— Orthicon Camera 1 .49 e t seq. 

— Sticking 1.57 
Inlay 1.97 
Interlaced Scanning 1.32 et seq. 

Lecher Line 1.108 

Lens Turret 1 .58 

Light (How Converted into Signal) 1.12 

— (How Produced) 1 .9 et seq. 

— (Nature of) 1.7 
Lighting Supervisor 1.88 
Line-of-Sight Transmission 1.121 
Line Scan Period 1.36 

— Scanning Frequency 1.35 

— Sync Pulse 1 .69 et seq. 

Master Sync Pulse Generator 1.90 

Maximum Picture Signal Frequency 1.133 

Microphone 1 .6 

Monitors 1.87 

Montage Effects 1 .96 et seq. 

Mosaic 1.37, 1.42 

NipkowDisk 1.27 

Operating Frequencies (of British 

T. V. Transmitters) 1 . 1 25 

Optical Horizon 1.121 

— Window 1.41,1.60 
Overlay 1 .97 



1.69 



Panning 

Pedestal 

Persistence of Vision 

Persuader Grid 

Photo-cathode 

Photocell 

Photo-conductivity 

Photo-conductive Lag 

Photo-electricity 

Photo-multiplier 

Photosensitivity 

Picture 

— Elements 

— Frequency 

— Signal 

— Sync Ratio 

— Tube 
Polar Diagram 



1.94 

1.76 

1.18 

1.56 

1.13,1.37,1.52 

1.13 

1.14, 1.61 et seq. 

1.63 

1.12 

1.39 

1.12 

1.32 

1.16 

1.32 

1.44, 1.64 

1.76 

1.29 

1.120 



Polarization (of Transmitted Wave) 1.118 


Spectral Colours 




1.8 


Post-Sync Field Blanking 


1.75 


Spectrum of Visible Light 




1.8 


Presentation Suite 


1.85 


Spill Light 




1.96 


Producer 


1.87 


Studio, Television 




1.81 et seq. 


Production Control Room 


1.87 


— Floor 




1.86 


— Studio 


1.85 


— Manager 
Switching Centre 




1.86 
1.82 


Raster, Scanning 


1.30 


Sync Pulses 




1.68 


Reactance Modulator 


1.104 








Reflector 


1.121 


Talk-back Apparatus 




1.92 


Resolution 


1.16, 1.46, 1.63 


Target 


1.29, 


1.37, 1.54, 1.60 


Resolving Power 


1.16 


— Mesh 




1.52, 1.60 


Retrace Time 


1.69 


Telecine Machine 
Tilt 




1.84 
1.95 


Scan 


1.29 


— Angle 




1.128 


Scanning 


1 .29 et seq. 


Timebase 




1.29 


— - Frequencies 


1.35 


Tonal Content (of Scene) 




1.15 


— Raster 


1.30 


Trace 




1.29 


Secondary Electrons 


1.38 


Transmitter, Vision 




1. 100 et sea. 


— Emission 


1.38 


Transmitting Aerial 


\A\4ct seq., 1.126 


Ratio 


1.38 








Selenium 


1.14 


Vestigial Sideband Transmission 


1.109 


Sensitivity (of Iconoscope) 


1.46 


Video Signal 




1.68 


Sequential Scanning 


1.24 


— Effects 




1.98 


Service Area 


1.121 et seq. 


Video-Tape Machine 




1.84 


Shading Signals 


1.48 


Vidicon Camera 




1 .60 et seq. 


Sidebands 


1.131 


Viewfinder 




1.58, 1.60 


Sideband Filter 


1.109 


Vision Mixer 




1.87 


Signal Bandwidth 


1.131 


— Signal 




1.5 


— Load Resistor 


1.45 


— Supervisor 




1.88 


— Plate 


1.42, 1.60 


— Transmitter 




1.100 


— Routing 


1.82 








Silhouette Pulse 


1.97 


Wall Anode 




1.55, 1.60 


Skip Distance 


1.123 


Waveband of Visible Light 




1.7 


Sound Signal 


1.5 


Wipe 




1.97 


— Supervisor 


1.88 








— Transmitter 


1.102 et seq. 


Zoomar 




1.58, 1.95 



BASIC TELEVISION 

Part 2 




A Basic Training Manual developed by 

H. A. COLE, CEng., M.I.E.R.E., 

working in conjunction with 

the Editorial and Art Staff of the Publishers. 




OXFORD 

THE TECHNICAL PRESS LTD 



NEW YORK 
THE BROLET PRESS 



Copyright © 1972 by 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 

New York, U.S.A. 

All rights reserved 



First published 1972 
Reprinted 1976 



The words "COMMON-CORE", with device and without device, 
are trade-marks of the Copyright owners 



SBN 291 39573 2 



Made and printed by offset in Great Britain by 
William Clowes & Sons, Limited, London, Beccles and Colchester 



PREFACE 

.the aim of this Series on BASIC TELEVISION is to explain in simple language the 
physical principles which make television possible and the way in which a typical 
television system works — from the generation of the signal in the TV camera to the 
final presentation of the picture on the screen by your own fireside. The Series is based 
on the two TV systems working in Great Britain today — the very-high frequency (VHF) 
one working on 405 lines per picture and the ultra-high frequency (UHF) one working 
on 625 lines per picture. The receiver considered in Parts 2 and 3 is the British Dual- 
Standard Receiver which is capable, on operation of the "Standard Selection" control, 
of receiving programmes on either of these two considerably different systems. 

Two decisions of particular importance had to be made in planning the Series. 
The first was to describe the working of the TV receiver almost wholly in terms of 
valves, even though in many of the latest single-standard and colour receivers the 
thermionic valve is being progressively replaced by semiconductor devices. This 
decision was made on two grounds. The first was that a large majority of the 
millions of receivers operational in Britain in the second half of 1971 are wholly or 
mainly valve-operated rather than transistorized and that, for technical and economic 
reasons which are more fully discussed in the final Section of Part 3 "trends in tv 
receiver design", the valve will in all probability continue to play an important part 
in TV receivers, especially in those built on the Dual-Standard principle, for a signi- 
ficant number of years to come. The second reason was that, since the COMMON- 
CORE Series as it exists at present is planned on the basis of explaining the working 
of electronic devices in terms of current flow through a valve, it was desirable to 
keep this account of the basic principles on which television works compatible with 
the foundation COMMON-CORE volumes in their present form. 

The other major decision in planning BASIC TELEVISION was to cover black-and- 
white ("monochrome") transmission and reception only, in the interest of keeping 
the descriptions of the various stages in the studio camera, the transmitter and the 
receiver relatively simple and relatively short. With the basic principles involved thus 
established (it is hoped) in the reader's mind, a further Series on Basic Colour TV, 
fully transistorized to reflect modern progress, is currently planned. 

Most of the measurements given in the Series have been expressed (or in Part 1, 
which was first published in 1967, re-expressed) in SI Metric units. In particular, 
"Hertz" and "MHz" have been used in place of "cycles per second" and "Mc/s" 
throughout. But certain measurements either familiar to the viewer (e.g., the sizes 
of picture tube) or else representative of orders of magnitude rather than of precise 
distances have been left in inches, miles, etc., as being more likely in that form to give 
the ordinary reader a clear picture of the point being made. 

The Series has been written and illustrated to take its place in the growing 
COMMON-CORE Series of Illustrated Training Manuals on subjects connected with 
electricity and electronics. Originated in the United States by the distinguished 
New York firm of technical education consultants and graphiological engineers, 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 



\ 
the twenty-one Manuals of which the COMMON-CORE Series now consists have 
already sold over 1,500,000 copies in their British and Commonwealth editions. Six 
of the Manuals have been wholly conceived, written and illustrated in the United 
Kingdom; while all the remainder have been extensively rewritten to conform with 
British terminology and notation. 

The BASIC TELEVISION Manuals presuppose in the reader a working know- 
ledge of the contents of the foundation volumes of the COMMON-CORE Series, 
principally the five Parts of BASIC ELECTRICITY and the six Parts of BASIC 
ELECTRONICS. Prior acquaintance with the two-part series BASIC ELEC- 
TRONIC CIRCUITS will also prove useful when the operation of the TV receiver 
is studied in Parts 2 and 3. 

The BASIC TELEVISION Series has been written, in conjunction with the editorial 
staff of the Publishers, by Mr. H. A. Cole, a Senior Scientific Officer in the Elec- 
tronics and Applied Physics Division of the Atomic Energy Research Establishment 
at Harwell. Mr. Cole is a Chartered Engineer, and a Member of the Institution 
of Electronic and Radio Engineers. All illustrations of a technical nature have been 
drawn by Mr. Cole himself, with the Art Department of the technical press 
responsible for their "decoration" and captioning. 



TABLE OF CONTENTS 

Section Page 

9 Introducing the TV Receiver 2.1 

10 The Receiving Aerial 2.17 

11 The Tuner 2.59 

12 The IF Amplifier 2.83 

13 The Sound Signals 2.110 

14 The Video Detector 2.116 

15 The Video Amplifier 2.122 

16 Coupling the Video Signal to the Picture Tube 2.136 




The COMMON VW\ CORE Series 



of Basic Training Manuals 
embraces so far the following titles : 

BASIC ELECTRICITY 

BASIC ELECTRONICS 

BASIC SYNCHROS AND SERVOMECHANISMS 

BASIC ELECTRONIC CIRCUITS 

BASIC RADAR 

BASIC INDUSTRIAL ELECTRICITY 

BASIC TELEVISION 



Foreword on International 
TV Systems 



The television set round which this Series has been written is the so-called British 
Dual-Standard Set, which is capable of receiving signals on two distinct line-systems — 
the 405-line and the British 625-line systems. 

If you wonder at the emphasis placed on the word "British" in that phrase, "the 
British 625-line system", the reason for it is that it has regrettably not yet been possible 
to secure international agreement on all the technical details of any standard 625-line 
system. 

For some time past, it has been the aim of the CC I R (the Comite" Consultatif 
International des Radio, or International Radio Consultative Committee) to persuade 
all the countries of the world to adopt a common TV system, on the grounds that it 
would be of great benefit to everyone from the point of view of convenience, ease of 
programme exchange, and manufacturing economy. Although complete agreement 
is still a long way off, progress has certainly been made over the past few years. 

There are at present seven major TV systems in the world : the American 525-line, 
the French 625-line, the French 819-line, the West European 625-line, the East 
European 625-line, the British 405-line, and the British 625-line systems. The 
British 405-line system is due to be gradually discontinued over the next few years 
and will eventually be replaced by a 625-line system. 

Unfortunately, not all European countries — even the Western ones — agree on the 
technical details of a standard 625-line system. It is true that they agree on such 
important features as aspect ratio, scanning sequence, method of interlacing and a 
few others; but differences still exist over (for example) the choice of vision bandwidth, 
channel spacing, sound-to-vision carrier spacing, and the degree of modulation which 
shall correspond to black level. These differences, though not very great, can some- 
times prevent satisfactory exchange of two 625-line programmes. For example, the 
625-line system employed by Belgium and France uses amplitude modulation for the 
sound carrier, whereas all other European countries use frequency modulation. 
Similar differences exist elsewhere in Europe over the relative spacing of the sound 
and vision carriers. 

The Western European and Eastern European systems differ mainly in the values 
chosen for channel width and vision bandwidth. The Western European system uses 
a 5 MHz vision bandwidth and 7 MHz channel spacing, whereas the Eastern European 
system uses a 6 MHz vision bandwidth and 8 MHz channel spacing. 

The British 625-line system differs from both European systems in that it uses 
a 5-5 MHz vision bandwidth and 8 MHz channel spacing. Other differences concern 
the width of the vestigial sideband and the setting of the black level. 



§9: INTRODUCING THE TV RECEIVER 



2.1 



You learnt in Part 1 of Basic Television the general principles on which TV works. 
You saw how the sound and vision signals are produced in the studio or processed 
there after an outside broadcast, and you learnt about the various stages through which 
these signals must pass before they are radiated by the transmitter. At the end of 
Part 1, you left the sound and vision signals quite literally "in the air" between 
transmitter and receiver. 

You must now see how these signals are picked up by the aerial, and how they are 
thereafter resolved in the receiver into the picture you wish to view and into the 
accompanying sounds which, with the picture itself, will reproduce by your fireside the 
complete scene enacted in the distant studio. 

Reduced to its essentials, the purpose of a television receiver is to select from the 
many signals which are always present in the air that one signal which carries the 
desired train of information, and then to decode and process this signal in such a way 
that it will cause to be reproduced an audible and visible image of the studio scene. 
To achieve that purpose, the receiver must be capable of performing certain basic 
functions. 



THE BASIC REQUIREMENTS 
of a TV Receiver 



First, as has been said, the appropriate signal in the required channel must be 
selected from the other signals present at the aerial, and then amplified to a usable 
level. 

Second, the sound and vision carriers must be separated from one another, and 
passed to their respective sound and vision circuits for demodulation — namely, the 
extraction from the carriers of the audio and video modulations which contain the 
required intelligence. The resulting audio signal, after further amplification, is taken 
to the loudspeaker and made to modulate it so that it will reproduce audibly the 
audible content of the studio scene. The video signal, also after further amplification, 
is used to modulate the intensity of an electron beam scanning the raster of the 
picture tube, and so to reproduce visibly the visible part of the studio scene. 

Third, sync pulses extracted from the video signal must be made to synchronize both 
the operating frequencies and the starting times of the line and field scanning circuits 
which produce the raster of the picture tube. 

Lastly, separate power supplies must be connected to produce both the very high 
voltage needed on the picture tube, and the much lower HT voltage required by all the 
other circuits in the receiver. 

The large illustration across the next two pages shows in block diagram form these 
various essential functions of the TV receiver. Note that the illustration is in no sense 
a circuit diagram. You will find that the circuits and blocks of circuits which have 
been developed to enable the various functions of the receiver to be performed differ in 
some important respects from the layout shown overleaf. 



2.2 

The Essential Functions of a TV Receiver 



AERIAL 



SOUND 
CARRIER 




CHANNEL 

SIGNAL 

SELECTION 



SOUND/VISION 

CARRIER 

SEPARATION 



VISION 
CARRIER 



Wm 



{ESSE NTIAL FUNC TIONS 
\of a TV Receiver 1 




SOUND 

CARRIER 

AMPLIFICATION 




VISION 

CARRIER 

AMPLIFICATION 



1 1 1 1 


H.T. AND L.T. 
SUPPLY TO 
ALL STAGES 



[§9 



The broad principles of operation of a TV receiver are similar to those of the 
amplitude-modulated radio receiver you learnt about in Basic Electronics. Both make 
use of the superheterodyne principle for achieving high gain and good selectivity for 
both the sound and the vision signals. Both use similar circuits for such purposes as 
signal detection, suppression of noise, and automatic gain control. 

The essential differences between them are that the TV receiver normally operates at 
frequencies not used for the domestic radio receiver, and that it contains additional 
circuits devoted to reproducing the picture on the picture tube, to generating timebases 
for the line and field scans, and to synchronizing these timebases with those used in the 
TV cameras in the studio. 

Note that the illustration above shows only the essential links in the process of 
achieving audible and visible reproduction of the studio scene. Many and great 
advances in receiver design have been achieved in the last 30 years. You will be 
learning about many of these improvements later on, and you will see that the perform- 
ance of a TV receiver reduced to the essentials shown above would fall far below 
present-day standards of viewing. 

Without these essentials, however, there would be no sound or picture at all. 

Two different systems of television exist, as you know, side-by-side in Britain today. 



§9] 

The Essential Functions of a TV Receiver 



2.3 



AUDIO 
DEMODULATION 



VIDEO 
DEMODULATION 



nnjir. 



SYNC 
PULSES 



(YV) 



w-*- 




PICTURE/SYNC. 
SEPARATION 



LINE AND FIELD 

SCAN WAVEFORM 

GENERATORS 



AUDIO ^ 

SIGNAL 



LOUDSPEAKER 



SUPPLY 




PICTURE 
TUBE 



^SCANNING 
WAVEFORMS 



The 405-line system operating in the VHF band of frequencies carries the programmes 
radiated by BBC 1 in Band 1, and by all stations of ITV in Band 3. It employs 
positive amplitude modulation of the carrier to produce its vision signal, and amplitude 
modulation for its sound carrier. 

The 625-line system operating in the UHF frequency band carries the programmes 
put out by the BBC and the Independent Television companies in Bands 4 and 5. It 
employs negative amplitude modulation to produce its vision signal, and frequency 
modulation for its sound carrier. 

Since it appears likely that both of these systems will co-exist in the British air for a 
good many years to come, the type of receiver studied in this book will be the so-called 
Dual-Standard Set. These sets consist effectively of a complete 405-line receiver 
linked to a complete 625-line receiver inside the same set, but with both receivers using 
common circuits and components wherever possible. Since the two receivers work on 
different line structures and employ different modulation techniques, a complex 
switching arrangement is involved every time the viewer changes from one system to 
the other. 

The principles on which the 405-line system operates follow on directly from what 
you learnt in Basic Electronics, so the circuit arrangements of that system will generally 
be treated first in the pages which follow. 



2.4 



[§» 



The Pattern of the Series 

In the remainder of this Section, the circuit arrangements of a dual-standard TV 
receiver will be built up in block schematic form. These blocks will then be put 
together to make up the complete receiver shown (still in block diagram form) in the 
large double-page illustration on pages 2.14 and 2.15. 

Later Sections will consider each of these blocks in greater detail; and the series will 
conclude with a short Section on fault-finding in TV receivers. 

Essential Components of the TV Receiver — The Aerial 

The starting point in the reproduction process is, of course, the aerial ; for it is here 
that the wanted signal is collected from the air before being applied to the receiver 
proper. 

You already know that aerials designed to pick up signals lying within different 
wavebands are of completely different shapes. For some 20 years or so past, the urban 
skyline of Britain has been festooned with the familiar designs erected to pick up the 
Band 1 programmes of the BBC. 



X-type 




AERIAL SHAPES 

for Reception of 

Signals in BAND 1 




H-type 



Such aerials as these would have a poor response, however, to signals whose 
frequencies lay (for instance) in Band 4, between 470 and 582 MHz. That is why 
aerials designed to pick up Band 4 and Band 5 programmes are typically shaped 
very differently. 





AERIAL SHAPES 
for Reception of Signals 
in BAND 4 

Aerial arrays can be designed to respond much better when they are pointed in one 
direction rather than in another. When such a directional array is properly mounted 
on the viewer's roof, its degree of selectivity towards the wanted signal will be much 
improved. You will learn more about the properties of aerials in Section 10. 

All these different aerial shapes have been worked out over the years, both theoretic- 
ally and by a continuing process of trial and error. It is an interesting fact, however, 
that whatever their shape, all aerials contain (for a given signal strength) almost the 
same total length of metal tubing exposed to the air. Thus although a Band 4 aerial 
looks much smaller and more compact than a Band 1 aerial, both are in fact presenting 
about the same overall length of sensitive antenna to the incoming signal. 



§9] 



2.5 



Essential Components of the TV Receiver — The Tuner 

The next stage in the handling of the signal detected by the aerial is the so-called 
tuner section, which is physically situated inside the receiver itself. Signals reach the 
tuner from the aerial through a coaxial cable. The internal impedances of aerial and 
tuner need to be carefully matched in order to keep attenuation of signal strength to 
a minimum. 

The function of the tuner is to pick out from the range of channels selected by the 
aerial the particular sound and vision signals required by the viewer, and to amplify 
them to a usable level. 

Much further amplification will be required in later stages, however, and signals of 
the very high frequencies used in TV cannot in practice be satisfactorily amplified to 
the necessary levels without endangering signal stability. The second function of the 
tuner is therefore to lower the frequencies of the received sound and vision signals to 
intermediate frequencies, exactly as you saw being done to radio signals in Part 5 of 
Basic Electronics. The same superheterodyne principle is used as you there studied. 
The received signals are fed into a mixer stage together with signals from a local 
oscillator, whose frequency is constant and somewhat above the frequency of the 
received signals. There emerge two different and still separate frequencies (vision and 
sound respectively) as the joint output of the tuner section. 

The very high frequencies handled in the tuner section create some special problems 
which make it necessary for the section to be carefully screened, both from the aerial 
and from the rest of the circuits in the receiver. It is even necessary for the several 
circuits inside the tuner itself to be screened from one another. You will see how 
this screening is done in Section 11. 

The viewer selects the channel whose programme he wants to watch by operating a 
channel selection knob situated on the outside of the receiver casing, either on its face 
or on one of its side panels. Operation of this knob alters the settings (and con- 
sequently the values) of certain components contained in the r.f. amplifier. It also 
varies the frequency of the local oscillator so that the tuner as a whole shall respond 
to the frequency of a signal in a different channel. 

The basic layout of the tuner section is shown below. 




2.6 [§ 9 

Essential Components of the TV Receiver — The IF Amplifiers 

On leaving the tuner, the sound and vision i.f. signals pass on together towards 
intermediate-frequency amplification. The function of this i.f. amplifier section is to 
amplify the signals fed in from the tuner until they are large enough to be handled 
effectively in later sections. 

At this point, there arises an immediate difference between the two systems which 
are combined in the Dual-Standard Set. This difference is so important that the 
opposite page has been devoted to a highly non-technical cartoon which may help you 
to get the essence of the matter firmly into your memory. 

In this illustration the briefcases which the little matchstick-men are carrying to the 
appropriate i.f. amplifiers represent the respective i.f. carriers, and contain within them 
the modulations representing the sound and video messages. When the briefcases 
emerge, considerably enlarged, from the i.f. amplifiers, they are taken to the appro- 
priate sound and vision detectors, where they are opened up and their contents are 
inspected. 

If (as is possible) you have a mind which does not react happily to that sort of 
pictorial stimulus, here is the nub of the distinction in words. 



Of 7HB VHF (405-Utte) SYSTEM 



There are separate amplifiers for the sound and vision i.f. signals. It is therefore 
necessary to separate the two signals before the full i.f. amplification is applied to 
either of them. 



m THB UHf (625-Lm£) SYSTBM 



Both sound and vision i.f. signals are put through a common i.f. amplifier, before 
being separated at a later stage in the receiver. 



Manufacturers' practice varies about the way in which the sound and vision signals 
in the 405-line system are separated from one another before being fed to their 
respective i.f. amplifiers. Sometimes the signals are separated as soon as they leave 
the tuner, by being passed through & filter. This consists generally of a series-tuned 
circuit offering a much lower impedance to one of the signals than it does to the other, 
so enabling the two signals to be routed in different directions. Sometimes separation 
of the signals is accomplished in this way only after some degree of i.f. amplification 
has been applied to both of them. 

Amplification of both signals in the 405-line system is achieved in accordance with 
the general principles you learnt about in Part 5 of Basic Electronics. The separated 
sound and vision signals reach their respective i.f. amplifiers, where two things happen 
to them. Each is passed through a number of consecutive amplifying valves — 
generally not fewer than three of them. The anode circuit of each of these valves is 
tuned to resonate to the i.f. frequency of the signal it is handling— but much more 
sharply so tuned than the circuits in the tuner itself could be. A second function of 
the i.f. amplifier in the 405-line system is thus to increase the selectivity of the receiver 
as a whole. 

In the UHF (625-line) system, the sound and vision signals are amplified together 
in a series of similar stages, and are passed on (still unseparated) to the next section. 



§9] 

Routing the Sound and Vision Signals 



2.7 



t 

I' 

m 





fgUJ 

Jjoa 
loo 


■ 






$ I 

I* 
I 



i 

m 






?~ 


Q> o 


"a a) 


> 0) 


Q 








£37 




2.8 



[§9 



Essential Components of the TV Receiver— The Video Section 

The video section of all TV receivers consists essentially of three stages— the video 
detector, the video amplifier and an automatic gain control (AGC) circuit. 



The I.F. AMPLIFIERS and VIDEO Sections 



■m 



1 


I.F. Arr 

To ^_ 


lplif ier Section 


i i 
i , 

i ■ 


Video Section 


~I ^_ ' 


Gain-controlling j Voltage 


Automatic 
Gain Control 






Tuner 


i 


r 


■ i 
i i 




i 


1 J Picture 1 




i i 
i i 










i 


\i 


L Video \ I 






Vision and 
Sound I.F. 
Amplifiers 


i i 
i i 


Video 
Detector 




Video 
Amplifier 


Signal \ § 






i*i 
i i 






1 




i i 
j i_ 










J ^ ^^ Contrast 
r Kjr Control 

chronising £00. 
ction .*$®1|IIk 


i 

I.F. Si 
from " 


jnals 
'uner 






6 MHz In 
Signal to 
Sound Sc 


' 1 
tercarrier To Syn 
625-line Se 
action 



The task of the video detector is to remove from the vision carrier the video modula- 
tion carrying the picture information, and to pass this information to the video 
amplifier. After amplification in that stage, the video signal is applied to the cathode 
of the picture tube, where it is used to modulate the intensity of the scanning beam in 
such a way as to build up the desired picture on the screen. 

In the 405-line system, in which the sound signal never reaches the video stage at all, 
this process is quite easy to follow; but a complication arises in the 625-line system, in 
which (you will remember) the sound and vision signals are at this stage not yet 
separated. You must now pause a moment to see how this separation is achieved. 

The UHF vision signal, you will recall, is amplitude-modulated and has a steady 
frequency; the sound signal is frequency-modulated. When the two signals arrive 
together in the video detector, they are made to beat together in a mixing action, with 
the constant frequency of the vision signal acting as a kind of local oscillator beating 
with the varying frequency of the sound signal. The output is a difference frequency 
which you can simply calculate as 39-5 MHz (the frequency of the vision i.f. signal), 
minus 33-5 MHz (the frequency of the sound i.f. signal), plus and minus the compara- 
tively small (75 kHz maximum) frequency variation of the sound signal. The 
difference frequency thus has a steady mean frequency of 6 MHz, which is being 
continuously modulated either side of 6 MHz by the frequency-modulated sound 
signal. 



§9] 



2.9 



Essential Components of the TV Receiver— The Video Section (continued) 

The 6 MHz beat-frequency signal you learnt about on the last page is commonly 
called the intercarrier signal. It needs now to be applied to the sound section of the 
receiver to be demodulated for extraction of the sound signal. 

It is fed to this section through the now-idle sound i.f. amplifier stage of the 405-line 
system, and dealt with thereafter in the sound section of the receiver in a way you will 
read about in a moment. Note at this point that it is a feature of the British Dual- 
Standard set for stages in the system to which the receiver is not at a given moment 
switched to be used in the circuitry of the system to which the receiver at that moment 
is switched. 

So much, for the present, for the video detector. From the video amplifier there 
are taken two other outputs in addition to the video signal. The first serves, in both 
systems, as input to the first stage of the synchronizing section, of which more below. 
The second is a voltage applied to an AGC circuit, which works somewhat differently 
in the two systems. In the 405-line system, the AGC voltage is fed back both to the 
vision i.f. amplifier and to the VHF tuner. In the 625-line system, it is applied to the 
vision i.f. amplifier only. Wherever it is applied, its purpose is the same— to try to 
maintain at a constant level the overall gain of the vision circuits of the receiver, 
whatever the strength of the received signal. The object is to prevent the frequent 
sudden and inevitable increases and decreases in signal strength from showing up as 
irritating brightenings and dimmings of the picture on the screen. 

The AGC circuit does its job by producing a negative feedback voltage of appropri- 
ate value, which varies the bias conditions (and therefore the gain) of the valves to 
which it is applied. 

Only one other feature of the video section remains to be mentioned at this stage. 
The contrast control knob is situated on the outside of the receiver (on its front or on a 
side panel), and is manipulated by the viewer to vary the amplitude of the video signal 
applied to the cathode of the picture tube, and so the degree of picture contrast (that is, 
the ratio of "blackness" to "whiteness" on the screen). 

Essential Components of the TV Receiver— The Sound Section 

The only complication in the sound section of the British Dual-Standard receiver 
arises from the fact that the VHF (405-line) system uses amplitude modulation to 
produce its sound signal, while the UHF (625-line system) employs frequency 
modulation. 



Volume Control 



405-line 
Sound I.F. 



Voltage 



Automatic 

Volume Control 

(A.V.C.) 



me mm mtim *r the 

TV Receiver , — J 



V 



I.F. Amplifier 



6 MHz Intercarrier 
Signal 




A.M. Detector 



F.M. Discriminator 




I Volume 
Control Knob 



2.10 [§9 

Essential Components of the TV Receiver— The Sound Section {continued) 

The process of AM detection was described in Part 5 of Basic Electronics, that of 
FM detection in Part 6 of the same Series. There is no point in repeating the two 
accounts in detail here. The VHF signal arrives from the VHF tuner and the sound 
i.f. amplifiers direct. In the AM detector block it is demodulated, in exactly the 
same way as it was in the superhet radio receiver you studied in Part 5. The 
Automatic Gain Control circuit was described in Basic Electronics, and performs 
the function of keeping the receiver output reasonably constant despite variations in 
the strengh of the received signal. 

In the UHF system, the 6 MHz intercarrier from the vision detector arrives in the 
sound section via the sound i.f. amplifiers of the VHF system (which, you will recall, 
the UHF system in the Dual-Standard Receiver "borrows" for the occasion). This 
intercarrier, however, is carrying both the amplitude modulation which is required in 
the vision section and the frequency modulation which is required in the sound section. 
It is therefore passed first through a limiter circuit (such as you can study in detail in 
Part 1 of Basic Electronic Circuits) where it is "clipped" of all variations in amplitude 
so that the signal emerging is one varying in frequency only. 

The FM output of the limiter is then converted to an audio signal in a discriminator 
circuit such as you read about in Section 3 of Basic Electronics, Part 6. The frequency 
of this audio signal is the frequency at which the deviation of the FM signal occurs. 
Its amplitude varies with the magnitude of this frequency deviation. 

Both audio signals, VHF and UHF alike, are then amplified in a sound power 
amplification section common to both systems, and are routed to the diaphragm of the 
loudspeaker. The audio level of the loudspeaker output is controlled in the usual way 
by means of a volume control knob such as you have manipulated many times on the 
front panels of your own TV or radio sets. 

Essential Components of the TV Receiver— The Sync Section 

You will recall that the video amplifier in the video section of the receiver has a 
third output (in addition to its outputs to the AGC circuits of both systems and to the 
cathode of the picture tube). 

This third output is applied to the synchronising section — and you will be relieved 
to learn that from this point on there are no differences of principle in the way in 
which the two systems in the Dual-Standard Set operate. The signal entering the 
sync section is no different from the video signal applied to the cathode of the picture 
tube. This latter signal is simply divided up so as to flow, in unequal proportions, to 
the two sections — rather as if it were water flowing down a pipe to a tap connection 
having two outlets of different sizes. 

Field Sync 

Video Signal 

from Video — 

Amplifier 



';,Sync Pulse-; 
§i& Separator s 



Pulses 
;>j*Sync Pulse 
11, Shaper 1 

:iUy^it. ff. :" \ ^ fc Line Sync 
Pulses 



The SYNC SECTION 

The function of the sync section is to separate out the sync pulses from the picture- 
signal content of the video signal. It then sorts out the line sync pulses from the field 
sync pulses, and converts them into waveforms capable of synchronising the oscillators 
of the line and field scan generators. 



§9] 



2.11 



Essential Components of the TV Receiver — The Sync Section (continued) 

The separation of the sync pulses from the picture signal is achieved with the aid of a 
simple limiter circuit (Basic Electronic Circuits, page 1 .48). Separation of the line and 
field sync pulses from one another calls for the use of integrating and differentiating 
circuits of the type you will most easily recall by referring to Basic Electronic Circuits, 
page 1.44. 

After they have been separated, both line and field sync pulses are put through 
separate amplitude-shaping circuits (Basic Electronic Circuits, Section 11) whose 
function is to "square up", or make more nearly vertical, the leading edges of the two 
pulse trains. The line and field scans both need to be triggered off at very exact 
moments of time, and this can only be done with sufficient accuracy if the trigger 
pulses have sharply-defined leading edges. 

Essential Components of the TV Receiver— The Field Scan Section 

In both systems in the Dual-Standard Set, the function of this section is to produce 
a waveform which is capable of causing the scanning beam of the picture tube to 
complete one vertical scan of the tube fifty times in every second. 

The scanning waveform itself is a linear rise or fall of voltage or current, such as a 
sawtooth (Basic Electronic Circuits, Part 2). Such a waveform, you there learnt, is 
capable of producing on the screen an accurate timebase on to which the information 
contained in the picture signal can be superimposed. Nearly all TV picture tubes 
employ the electromagnetic method of deflecting the electron beam down and across 
the face of the tube. This necessitates (Basic Electronic Circuits, Part 2 again) the 
application of a linearly rising or falling current to the deflection coils of the tube. 

The scanning waveform typically starts life as a reasonably square wave of voltage 
generated by an oscillator such as a multivibrator, oscillating at exactly 50 cycles per 
second (50 Hz). This is, as you know, the field scanning frequency. 

The oscillation is kept precisely at 50 Hz by the field sync pulse which is applied to 
it from the preceding sync section. You will remember that this sync pulse was itself 
applied to the transmitted signal by a similar oscillator working in the camera circuits 
many miles away. Remembering the enormous speed at which radio waves travel, 
you can now see how the apparent miracle is achieved of exactly synchronising the 
appearance of the televised picture on the screen of your set with the scan of the scene 
in the distant studio. 



^^m0Mp^#> 



Field Scanning 
Coils 



^*^&y#^*$$g 


.'ife-f 












r| 


Field Scan 
Oscillator 




Field Scan 
Shaper 


Field Scan 

Power 
Amplifier 




Matching 
Transformer 


. 1 








/ 

Field Sc 




, Field Sync 
Pulses 



Waveform 



The 50 Hz square wave of voltage is next passed through a shaping circuit, which 
re-shapes it as a sawtooth having a good linear rise (or fall). It is then sent on its way 
through the field scan power amplifier towards the field scan coils fitted round the 
neck of the picture tube. 



2.12 



t§9 



Essential Components of the TV Receiver— The Field Scan Section (continued) 

A difficulty arises, however, because the output of the field scan amplifier valve has 
a high impedance, while the field scanning coils of the tube have a very low one. In 
order to match these two Z's and so to ensure maximum transfer of energy to the coil, 
a matching transformer is inserted between the amplifier and the coil. 

This transformer is of the step-down variety, having many turns in its primary and 
few turns in its secondary. It is therefore capable of stepping down the voltage, and 
stepping up the current, to the desired level. (Exactly the same type of transformer 
is needed, as you will recall from Basic Electronics, Part 5, between the final amplifier 
of the ordinary radio receiver and the loudspeaker.) 

Essential Components of the TV Receiver — The Line Scan Section 

The function of this section is to produce a waveform which is capable of causing 
the scanning beam of the picture tube to move across the screen at the rate appropriate 
to the system. In the 405-line system, this rate is 202| horizontal scans during the 
period of every field scan. (Remember the effects of interlacing, with every line of the 
tube scanned once in every second field scan. Half 405 is 202Jk) 

Similarly in the UHF system, the rate of line scan needs to be half 625, or 312£, 
horizontal line scans during the period of every field scan. If you bear in mind that 
"the period of every field scan" is only one-fiftieth of a second, and that two or three 
hundred-odd line scans have to be accomplished within this period, you will get some 
idea of the enormous speed at which the scanning beam has to move. It can only 
hope to scan at this speed because (you will remember) an electron beam has virtually 
no momentum. 

The line scan waveform, like its field scan counterpart, is generated in an oscillator 
of one kind or another (different makers differ in their choice); but the frequency of 
oscillation must be enormously more rapid than it was in the field scan section. If the 
VHF scanning beam has to complete 202^ line scans in one-fiftieth of a second, its 
frequency needs to be 202^-^ cycles per second, or 10-125 kHz. By a similar 
calculation, the frequency of the 625-line oscillator needs to be 15-625 kHz. 

In exactly the same way as in the field scan section, the line scan oscillator is 
synchronized with the corresponding oscillator working in the distant camera circuits. 
The synchronized waveform is then passed in turn through an amplifier to "beef it up" 
and through a matching transformer to "step it down", before being applied to the 
line scan coils fitted round the neck of the picture tube. 



m im scut see e/M 



Hn 



Line Scanning 
Coils 





Line Scan 
Oscillator 




Line Scan 
Shaper 




Line Scan 

Power 
Amplifier 




Matching 












Transformer 


Line Scan 




— Line Sv 


nc 












Waveform 
^ E.H.T. (for F 



Pulses 




to E.H.T. Rectifier 



§»] 



2.13 



Essential Components of the TV Receiver — Power Supplies 

Both the VHF and UHF systems in the British Dual-Standard Receiver have need of 
two different kinds of power supply. 

(a) The EHT Section 

The job of this section is to produce the very high voltage needed at the final anode of 
the picture tube to draw the electron beam from the cathode of the tube to the screen. 
In a 23-inch tube, this voltage needs to be of the order of 18 to 20 kilovolts. 

Although the current carried is only a small one, a voltage of this order is obviously 
dangerous; so remember to keep your fingers away from the final anode connection of the 
picture tube whenever your TV set is switched on. 

The voltage required is generated in the matching transformer in the line scan section 
of the receiver. This transformer is actually an auto-transformer consisting of 
thousands of turns of wire. It is tapped at various points for connection to the line 
scan power amplifier and to the scanning coils. When the current to the picture tube 
falls away very rapidly during the fast fly-back period of the scan, a very high back- 
e.m.f. is induced across the auto-transformer. 



Iff 



Line Scan 
Waveform 



Line Scan 

Matching 

Transformer 




The a.c. voltage induced in the transformer during flyback is first applied to an 
ordinary half-wave rectifier circuit to turn it into d.c. The d.c. is then applied as an 
electron accelerator to the final anode of the picture tube. 

This method is used in all TV receivers, and is known as the flyback-derived type of 
EHT generation. 

(b) The HT Section 

All the valves in a TV receiver need a heater supply for their cathodes, and the usual 
HT supply to enable them to operate. 

This HT is derived from a section of the receiver which has been omitted from the 
block schematic of the Dual-Standard Set on the next two pages in order not to over- 
complicate the diagram. It is taken direct (i.e., without transformer action) from the 
mains supply, and (after rectification) is normally of the order of + 200 volts. It is 
then passed through the simple fine of circuitry shown below. 



The H.T. SECTION 



200/250v > 
A.C./D.C. Mains 



Half -Wave 
Rectifier 



H.T. Filter 



+ 200v D.C. 

to all Valve Sections 



2.14 



[§» 



Band 1 



Bands 4& 5 " 



Band 3 



■H 




405/625 Standard 
Selection Switch 



VHF Tuner 



UHF Tuner 



405/ 



625 



Gain Controlling / 
Voltage ' 



Gain Controlling 



Sound 
I.F. Amplifiers 




Vision I.F. 
Amplifiers 



>• V 



(Circuit change) 
625 • • 405 



Sync Pulse 
Shaper 



Sync 
Separator 




Field Scan 
Oscillator 



5 V e 

fc(Ci 



405 # #625 

I (Circuit change) 



Line Scan 
Oscillator 



§9] 



2.15 






AVC Circuit 




Voltage 






* 






A.M. Sound 
Detector 








405 




A 






i 

• 


Sound 
Amplifier 


, m A 




w *— 

625 


■^ 




F.M. Sound 
Discriminator 








. Loudspeaker 




















■ A.G.C. Circuit 










Scanning A 1 












I 






Vision 
Detector 




Video 
Amplifier 


i 


IfPicture] 










BBHubeJl 






# (Circuit ( 
625 • • 405 


:hange 


) 


625 • 


Circuit change) 
fc 405 ' ' 




Field Scan 

Output 

Amplifier 






i 


\ Final 






' 


Anode 


405* #625 
^ (Circuit 


change) 


, 


Line Scan 

Output 

Amplifier 
















1 


' 




















E.H.T. 






Power 


Supply 

















2.16 



[§9 



Essential Components of the TV Receiver — Power Supplies (continued) 

In the HT section, the a.c. from the mains is first rectified by being passed through a 
normal half-wave rectifier circuit. The d.c. output from this circuit is then smoothed 
(that is to say, it has the ripples taken out of it) by being applied to an HT filter circuit, 
which uses either the LC or (more usually) the RC smoothing techniques you learnt 
about in Section 6 of Basic Electronics, Part 1. 

WARNINC 

Remember always that the HT supply at any point in a TV set is capable of delivering a 
much higher current than is the EHT supply, despite the latter 's much higher voltage. So 
whereas a shock from the EHT can give you a nasty burn, current from the HT supply can 
kill you. For this reason, it requires considerable experience before you can safely fiddle 
about anywhere in the circuitry of a TV set once it is switched on. YO U HA VE BEEN 
WARNED! 

Whatever the degree of your experience, the following rules for exploring the inside 
of a live TV set should always be observed : 

Q Never do it with wet hands. Water conducts current far better than does dry skin. 

Q Always see that the handle of any tool you intend to use is properly insulated, and not 
broken or cracked. To patch up a split handle with insulating tape may work all right 
for a long time, but — is it really worth the risk ? 

© Rubber is a useful insulator. It is safer to wear rubber-soled shoes, and to stand on a 
dry rubber mat. Even if you haven't got such a mat handy, do at least see that the 
surface you stand on is dry. 

© Keep your left hand in your trouser pocket all the time you are poking about inside the 
set with your right hand. If you do this, any current you may pick up will flow to 
earth down the right-hand side of your body. Most people keep their heart on the left. 

© The most hazardous posture you can possibly adopt is in some respects the most con- 
venient. But if you are probing away with your right hand in the set, with your left 
hand resting anywhere on the chassis, you are guaranteeing that any current you may 
pick up will take the most dangerous possible path on its way to earth through your body. 




§10: THE RECEIVING AERIAL 



2.17 



In a technically advanced country like Britain, the air is everywhere filled — indoors 
and out-of-doors, in bedrooms, bathrooms, churches, trains and in the depths of the 
countryside — with thousands upon thousands of man-made electromagnetic waves of 
varied origin and many different kinds. 

All of these waves (which can be neither seen, felt nor heard by an unaided human 
being) have been propagated in order to convey intelligence of one sort or another 
between two or more distant points. These messages are all present in the air 
together, some of the signals being very strong but many so faint as to be barely capable 
of detection. The carrier waves on which the signals travel vary greatly in wavelength. 

The import of the intelligence carried varies no less widely than the amplitude of the 
signals and the frequency of the carriers — ranging from landing instructions for a jet 
airliner coming in from Sydney or Capetown to the return echo of a radar beam 
bounced off the Moon, a Z-car message to a police patrol, or the sound-and-vision 
record of an athletics meeting in Russia travelling to New York via a satellite 
permanently stationed 2,000 miles above the middle of the Atlantic. 

Aircraft, shipping, military vehicles, radar, spacecraft, navigation aids, radio and 
television broadcasting — all contribute to the vast communications network which 
now encircles the globe. But of all these signals the most numerous and the most 
powerful are those which emanate from the various national radio and TV broadcast- 
ing stations ; and it is they alone which are the concern of this book. 




Spacecraft 

MM \ 
Shipping 



Radar 



Aircraft 

TCUVISIOH 

Taxis 





Designed to reach millions of listeners and viewers in their country of origin (and 
often in other countries as well), most radio and TV signals are extremely powerful. 
Yet there are a great many of them; they travel on very different frequencies; and even 
a strong signal can fade to low power in an area in which it may be important for it to 
be picked up. The receiving aerial must therefore be capable of very accurate selection 
among the multitude of signals impinging on it. It must be able to reject (or, rather, 
to remain indifferent to) the vast majority of them; but it must respond at once to the 
wanted signal, however faint it may be at the point of reception. 

You will not be surprised to learn that, with this sort of task to perform, the aerial 
of a TV receiving set is a more complicated object than a glance at the serried ranks of 
them rising above any street in urban Britain might lead you to suppose. 



2.18 



[§10 



The Requirements of a TV Aerial 

The task of a TV receiving aerial is to extract from the sound and vision signals of 
the desired transmission the amount of energy required by the receiver to reproduce 
adequately the sounds and the visual scenes created in the studio. Since the strength 
of the wanted signals varies so widely in different parts of the country, large and 
complicated aerials are needed to perform this function in areas of poor signal 
strength, while quite simple ones suffice in areas of good signal strength. 

The aerial performs its task by responding to signal frequency. It responds well to 
the frequencies of the two wanted (sound and vision) frequencies, but very poorly 
to signals of other frequencies. To be rather more precise, the aerial must be made to 
respond well to a narrow range of frequencies in a given channel in a given Band. 

In Britain, as you know, the sound and vision signals of a given transmission are 
contained in frequency channels either 5 MHz wide (in the 405-line system) or 8 MHz 
wide (in the 625-line system). The ideal aerial is therefore one whose frequency 
response, or bandwidth, is just wide enough to accept the range of frequencies con- 
tained in a single channel, and to reject all other frequencies. In practice, however, 
as you will see, it is sometimes necessary for the aerial to be made to respond to a 
group of channels rather than to a single channel only. 

The principle on which any receiving aerial works is that, when the electromagnetic 
waves radiated from the transmitter cut across the aerial, they generate in it a small 
voltage. This voltage causes to flow in the aerial system a weak current having the 
same frequency, and carrying the same modulations, as the current in the transmitter 
which caused the waves to be radiated in the first place. The problem is to design an 
aerial in which this induced current will be as large as possible when a signal in the 
desired frequency range cuts across it, and as low as possible when a signal of any other 
frequency arrives. 

You learnt in Basic Electricity, Part 4, that when an alternating voltage is applied 
to a series circuit containing both inductance and capacitance (in addition, of course, 
to the normal R which any circuit contains), current flow is greatest when the inductive 
reactance (X L ) of the circuit is equal to its capacitive reactance (X c ), so that their 
effects cancel out. Circuit impedance is then minimum (equal to R only), and the 
circuit is said to be "at resonance". The frequency of the signal which causes a series 

circuit containing given values of L and C to resonate is 
the "resonant frequency" (/,) of that circuit. 




1000 1500 
Frequency 



2000 



2500 



§10] 2.19 

The Requirements of a TV Aerial (continued) 

What is needed for an efficient aerial, therefore, is something equivalent to a series 
resonant circuit in which the values of L and C are such that X L and X c will be equal 
when an alternating voltage of the frequency of the desired signal is applied to the 
circuit. 

Such an equivalent is found in a length of hollow metal rod called a dipole, whose 
dimensions have been very carefully calculated. 

The values of inductive and capacitive reactance which a dipole offers to an incom- 
ing signal of alternating frequency depend on several factors — notably the length and 
diameter of the rod, and (to a much lesser extent) the material of which it is made. 

Aerial material is generally a compromise between good conductivity and reasonable 
cost, and the diameter of the aerial is largely dictated by another factor which you will 
learn about later on. This means that the amount of L and C in an aerial are 
principally dictated by its length. 

How long, then, must an aerial be to do its job with maximum efficiency ? 

Imagine that you have a length of aerial rod to which you have connected an 
instrument capable of measuring the strength of a collected signal, and that this aerial 
is positioned, well clear of the ground and of other obstructions, in the field of a passing 
signal. Imagine also that you can slowly adjust the length of this aerial without 
disturbing its position. 

You would find that as the aerial is slowly increased in length, so the signal collected 
by it increases correspondingly in strength. This is much what you would have 
expected, on the ground that the more there is of the aerial exposed to the passing 
wave, the more of that wave will it collect. 

You would also find, however, that when the length of the aerial becomes equal to 
one half-wavelength of the signal being collected, the strength of the signal greatly 
increases; but then for a time actually declines as you slowly increase the length of the 
aerial still further. When aerial length approaches a further half-wavelength (making 
one complete wavelength in all), the collected signal again climbs to a maximum 
value, this time greater than it was with aerial length at half- wavelength. This 
sequence would, if other factors played no part, repeat itself for each additional 
half-wavelength increase in aerial length. 

What is happening to the aerial to cause these fluctuating values of signal strength 
is that, whenever the aerial reaches the critical length of one half-wavelength (or any 
multiple of one half-wavelength) of the signal being received, it suddenly starts behav- 
ing like a series-resonant tuned circuit. X c and X L cancel out; minimum impedance 
is offered to the signal applied to the circuit; and the current flowing through the 
circuit rises to maximum value. 

Now fix your expandable aerial to a length equal to one half-wavelength of a given 
signal, so that it resonates to that signal. As signals of other frequencies close to, 
but not at, the resonant frequency strike the aerial, X c and X L in the aerial begin to 
move out of balance with one another; and either inductive or capacitive reactance is 
added to the circuit as the frequency of the applied signal rises respectively above or 
below the/ r of the aerial. Current flow becomes smaller and smaller; and soon the 
impedance of the aerial circuit rises to a point at which incoming signals whose 
frequency is too far removed from the/ r of the aerial have great difficulty in getting 
through at all. 



2.20 [§10 

The Requirements of a TV Aerial (continued) 

You have just seen that, provided its length remains an exact multiple of one half- 
wavelength of the required signal, the longer an aerial is, the better it is able to pick 
out this signal from the electromagnetic waves radiated by the transmitter. In 
practice, however, there are two good reasons why TV (and radio) aerials are seldom 
made longer than one half-wavelength of the required signal. 

The first is a matter of practical convenience. You know that the wavelength, in 
metres, corresponding to any frequency is 300 divided by the frequency in question, 
expressed in megaHertz (MHz). The lowest frequency used in British TV is 41 -5 MHz, 
which is the sound signal frequency in Channel 1. This corresponds to a wavelength 
of nearly 7 metres, or some 23 feet. An aerial of that order of size would be very 
difficult to handle, and to keep securely fixed on a wind-swept roof. Even at the top 
end of the VHF range in Band 3, the MHz divisor in the formula is still quite small, 
and all wavelengths in that range are still some metres long. 

This practical limitation on aerial size does not, of course, arise in respect of trans- 
missions radiated in the UHF bands. The frequencies there are so high that at the 
highest channel frequency (854 MHz) one wavelength is little more than 300 mm 
(12 inches) long. It would obviously be possible to construct and mount an aerial 
whose length is equal to quite a few of such wavelengths without serious difficulty. 
But now the second consideration arises. 

The polar diagram of a simple half-wave dipole mounted in free space is of such a 
shape that maximum reception occurs around its centre. That is to say, the reception 
pattern of a vertical dipole is predominantly horizontal. It consists of two lobes in 
the shape of the figure 8, as shown in the illustration below. 



Half -Wave Dipole 




For a full-wave dipole, however, the polar diagram is quite different. It consists of 
four lobes, none of them predominantly horizontal in orientation. 



Full-Wave Dipole 




Maximum receptivity is obviously impossible with such an aerial, for the received 
signal can never be arriving from the direction in which all four lobes are aligned, at 
one and the same time. Even if one lobe could be pointed directly at the distant 
transmitter (which would seldom be very easy) the receptive power of the other three 
lobes would be entirely wasted. 



§10] 2.21 

The Requirements of a TV Aerial (continued) 

The polar diagram of a two-wavelength dipole produces no improvement in 
receptivity. Although it has still more lobes, none of them is orientated towards the 
site of any likely transmitter, and again performance is apt to be poor. 



Two-Wavelength 
Dipole 




These lobes are really standing waves of received energy, and the reason why they 
make their appearance in the polar diagrams of the full-wave and two-wavelength 
dipole aerials is that part of the energy received by one half of the dipole is cancelled 
by energy of opposite phase present in the other half. This mutual cancellation of 
energy causes gaps where (in the half- wave dipole) there are lobes of energy, and lobes 
where there are gaps. 

In certain applications (such as radar, for instance) it is often desirable to have 
energy radiated in a semi-vertical direction; and in these applications the full-wave or 
two-wavelength dipole is often used. For television purposes, however, the more 
horizontal polar diagram of the half-wave dipole is much more suitable. 

Since, therefore, (a) the half- wave is the minimum length of dipole which will behave 
as a series-resonant circuit (and therefore the minimum length in which the first peak 
in the graph of collection efficiency will occur) ; and since (b) the half- wave is also the 
maximum length which presents an acceptable polar diagram, the great majority of TV 
receiving aerials are constructed of a length equal to one-half of the wavelength of the 
signal which they are designed to receive. 

Two Practical Consequences. 

Since aerial length plays so important a part in determining the range of frequencies 
which an aerial will efficiently accept, and since the length of a fixed aerial cannot 
easily be altered once it has been made, it follows that a given aerial will not necessarily 
be capable of accepting the same programme when it is taken to a different area of the 
same country. In the Reading area, for instance, the programmes of BBC 1 are 
broadcast on wavelengths of 7-23 and 6-6 metres in Channel 1 (that is to say, at sound 
and vision frequencies of, respectively, 41-5 and 45 MHz). In Birmingham, the same 
programmes are broadcast on wavelengths of 5-15 and 4-86 metres in Channel 4 
(i.e., at sound and vision frequencies of 58-25 and 61-75 MHz respectively. 

A person moving house from Reading to Birmingham would therefore not be able 
to use his old Band 1 aerial in the new area, and might just as well leave it behind for 
the buyer of the house he is leaving. 

Another consequence of the inability of a given aerial to respond to more than a 
narrow range of frequencies is that aerials capable of receiving ITV programmes in 
Band 3, or BBC 2 programmes in Bands 4 and 5, need to be quite different from 
Band 1 aerials, and call for separate arrangements in the receiver itself to accept their 
very different signals. 



2.22 



[§10 



Properties of the Half-Wave Dipole 

A great deal of what you learnt about the properties of the half-wave dipole in 
Part 1 are equally true of the dipole when it is used in a receiving role. The main 
difference is that the transmitting dipole receives the signal from the transmitter in the 
form of a flow of current and radiates it as an electromagnetic wave, while in the 
receiving dipole an alternating voltage is induced by the arrival of the same electro- 
magnetic wave. The principle is the same as that operating in a simple transformer, 
when a voltage is induced in the secondary winding (here, the dipole) by a voltage 
(here, the signal) generated in the primary. 

The current which is made to flow in a half-wave dipole by the voltage induced in it 
by the signal varies at different points along the dipole's length because of the way in 
which the self-capacitance of the dipole is distributed. This capacitance is maximum 
at the centre of the dipole, so that a large charging current flows at that point, and 
minimum at the ends, where only a small charging current flows. 

This distribution of current and voltage along a dipole remains constant irrespective 
of the strength of the signal. The distribution pattern shown in the illustration below 
is thus permanently valid. The heavy vertical line to the left represents a vertical 
dipole aerial. The voltage pattern along the aerial is shown in a thin unbroken black 
line marked V Wave, and the current pattern in a dotted line marked / Wave. 

Vwave 



2000 a 




The Otetritartfen of V, I and Z along the Baff-Wawi Oipote 



The first important point to note about this pattern chart is that at the ends of the 
dipole voltage is maximum and current minimum; while at the centre of the dipole 
current is maximum and voltage minimum. Since the object to be achieved is that the 
wanted signal shall be the one which causes the largest current to flow into the aerial 
terminals, it is usual for an aerial to be tapped for its energy at its centre point. 



§«0] 



2.23 



Properties of the Half-Wave Dipole (continued) 

The second important point to note about the illustration on the last page is that, 
since a half-wave dipole is a form of series-tuned circuit in which current flows under 
the impulse of alternating voltages, it must possess impedance. 

This impedance varies at every point along the dipole. It is very high (several 
thousand ohms) at either end of the dipole, but falls to a minimum value (at about 
73 ohms) at its centre. It can be calculated at any point along the length of the dipole 
by applying Ohm's Law to the instantaneous values of voltage and current at the 
desired point. That is how the curve of Z in the illustration on the last page was 
calculated. 

Provided that the ratio of dipole diameter to dipole length is kept very small, the 
value of impedance remains at the approximate value of 73 ohms at the centre of any 
half- wave dipole. This fact is of great importance, for it has led to the standardization 
of the characteristic impedance (Z ) of much of the feeder cable which is used to take 
the signal from the aerial to the receiver. 

You know from your work in Basic Electronics, page 4.48, that to obtain maximum 
transfer of signal strength this cable must have a Z as nearly as possible equal to that 
which exists at the point of its junction with the dipole. Since most dipoles are 
centre-fed, the standard feeder cable used in Britain is manufactured with a Z of 
some 73-75 ohms. 

As you will see in the next few pages, there exist many devices for improving the 
reception efficiency of a dipole aerial; but most of them have the disadvantage of 
altering the impedance at its centre point. It is always expensive and sometimes 
inconvenient to use non-standardized cable. That is why efforts are always made, 
when any of these devices are used, to counteract their effect on the centre Z of the 
dipole, and to bring it back to the approximate value of 75 ohms. 



The TV Aerial— Dipole Length 

You have seen that the overall length of a half-wave dipole, measured from tip to 
tip, needs to be equal to one-half of the wavelength of the frequency to which the 
dipole behaves as a resonant circuit. At that length, the inductive and capacitive 
reactances of the dipole cancel out when the wanted signal arrives. Dipole resistance 
becomes purely resistive and therefore a minimum — at about 73 ohms. 



J 
1 



X = 




2.24 [§I0 

The TV Aerial — Dipole Length (continued) 

The statement that the overall length of a dipole needs to be equal to half the wave- 
length of the wanted signal must now be modified in two respects. Neither modifica- 
tion is of first importance in itself; but both help to show the complexities which 
exist in aerial design, and the need there is for compromise in solving them. 

The first modification arises from the fact that all signals tend to travel slightly more 
slowly through the material of which an aerial is composed than they do through the 
atmosphere. This means that, to achieve perfect resonance, the correct length of the 
dipole needs to be slightly less than an exact half-wavelength of the required signal. 

In practice, therefore, the length of a "half-wave" dipole is made about 0-48 times 
the wavelength of the signal to which it is to resonate. In more exact formula, the 
length of a dipole, in feet, can be calculated by dividing the number 468 by the 
frequency of the desired signal, in MHz. 

The second modification arises out of the question, "What is the signal to whose 
frequency you want your aerial to resonate?" The TV aerial has to accept two 
signals, sound and vision, whose frequencies are some distance apart. The dipole 
cannot 'therefore be built to resonate exactly to both of them. 

Some manufacturers solve this dilemma by cutting dipole length so that the aerial 
resonates exactly to the vision signal, as being the more important of the two ; and 
are content to accept some attenuation of the sound signal in consequence. 

Others adjust dipole length so that the aerial resonates to what is called a centre 
frequency. This is the geometric mean between the frequencies of the sound and 
vision signals. In the case of the BBC 1 transmissions in Channel 1, this centre 
frequency is about 43 MHz. 



Geometric Mean of 
Sound and Vision = 
Frequencies 












£ 43 MHz 


V/i 


x/2 = 


V41-5 


X 


45 s 



A frequency of 43 MHz corresponds to a wavelength of almost exactly 7 metres. 
Tip-to-tip length of this dipole is thus a little under 3-5 metres, and each of its two 
A/4 sections measures nearly T75 metres, or about 5£ feet. 

A similar calculation for the ITV transmissions on Channel 9 of Band 3, whose 
centre frequency is about 193 MHz, shows that each quarter- wave section needs to be 
some 0-4 metres, or about 16 inches, long. 

You will gather from these figures that the overall length of dipole required decreases 
sharply as the operating frequency rises. This obviously has its inconveniences, in that 
the higher-frequency signals tend to be weaker (other things being equal) than the 
lower-frequency ones, yet need to be detected by a smaller aerial. You will learn 
about some of the devices used to meet these inconveniences later on in this Section. 

Meanwhile, it is of passing interest to note that if a half-wave dipole had to be 
designed to receive the BBC Light programme (sound only), which is broadcast at 
200 kHz on a wavelength of 1,500 metres, the overall length of the dipole would need 
to be almost half-a-mile ! That is why the use of dipole receiving aerials is usually 
confined to the h.f. and higher frequency bands — save in cases where convenience of 
installation must take second place to maximum efficiency. 



§10] 



2.25 



The Frequency Response of a TV Aerial 

You have seen that no dipole can be built to resonate exactly to the frequency of 
every signal it needs to pick up. What happens to the signal-collection efficiency of 
the aerial when it is "just off" resonance to the frequency of the desired signal ? 

You already know that the frequency response of a receiving aerial expresses the 
degree to which the aerial will respond to signals of equal strength but varying 
frequency. It is generally displayed in the form of a graph. To a signal frequency 
corresponding to the resonant frequency of the aerial, response will always be maxi- 
mum, and maximum current will be available at the aerial terminals to flow through 
the feeder cable to the first stage of the receiver itself. 

Ideally, the frequency response of a TV aerial designated to operate in a single 
channel would be as you see it opposite — I / I ^^^""""""""""m Max 

high and perfectly level to a range of signal / ^-Vision Carrier 

frequencies covering the sound and vision 
signals and their two "outer" sidebands, 
zero to all frequencies lying outside these _ 
limits. 



Sound _ 
Carrier 

Channel Width - 



:\: 



An acceptance bandwidth of this shape would ensure uniform receiver response to 
both sound and vision carriers and to their respective information-bearing sidebands, 
and would also ensure perfect rejection of all unwanted signals. 

Unfortunately, a frequency response of this shape is not attainable in practice; and 
the response of a typical half-wave dipole is more like that shown below. 



Actual 
Curve 



5-5 




Frequency Response 

of a 

Single Half-Wave Dipole 



Channel 5 

Limits (63-25-66-75 MHz) 




Channel 5 Signal Frequency (MHz) 

This frequency response is reasonably flat at the peak of the curve, where it covers 
the range of frequencies contained in the channel, but the ability to reject unwanted 
signals outside that channel is not good. You will shortly see how the frequency 
response of a dipole can be materially sharpened by the addition of further elements 
to the aerial array. 

It is also possible to broaden the frequency response of an aerial to cover several 
channels. This is not often desired at VHF, but one way of achieving it would be to 
increase the diameter of the rods of which the dipole is constructed. (Note, though, 
that this method affects the distribution of the internal impedances of the dipole, and 
so gives rise to matching difficulties with the feeder cable to the receiver.) 

In UHF aerials, however, a broader frequency response is generally required. In 
Britain, these aerials are built to operate over all four of the channels on which it is 
planned eventually to radiate 625-line programmes. Since these channels are not 
contiguous, UHF aerials need to cover a frequency range of some 88 MHz. You will 
see how this is done by means of special aerial designs later on. 



2.26 [§10 

VHF Aerials— The Reflector 

An ordinary dipole aerial, mounted vertically by itself in free space, would be very 
inefficient at its required task of detecting a particular signal coming from a known 
transmitter. The reason is that, mounted thus, it is what is called omni-directional, 
which means that it will receive signals with equal efficiency from any direction in the 
plane which lies at right angles to its own. It would therefore be liable to pick up 
interference signals of many kinds. 

Ideally, of course, the receiving efficiency of an aerial would be zero in all directions 
save that from which the desired signal is arriving, and 100% in that direction alone. 
Although such an ideal cannot be realised in practice, excellent directional properties 
can be given to an aerial by positioning behind, and in front of, the dipole additional 
elements known respectively as reflectors and directors. 

A reflector is a length of rod, usually a metal tube of the same diameter as the dipole 
but about 5% longer (so that it incidentally conforms rather closely to the exact half- 
wavelength which the dipole should theoretically measure). It is mounted in the 
same plane as the dipole, parallel to it, and behind it with respect to the incoming signal. 
The simplest example is the H-aerial with whose appearance you are very familiar. 

The reflector is not connected electrically 
with the dipole, and is mounted at dis- 
tances behind it which vary between one- 
tenth and one-quarter of the signal wave- 
length. This spacing is important, as you I B | Dipole PIllS 
will see in a moment. 




A Half-Wave 

Vipole Plus 

Reflector 



of Signal 



The effect of a reflector is to re-radiate 
some of the energy of the wanted signal 
back to the dipole in such a way that it 

arrives there in phase with the signal itself, | fl | -^ D ! r ^ ction , 

and thus adds to it. At the same time, any 

signal arriving from the rear of the dipole- 

reflector combination (i.e. , from a direction ■ V 

180° away from the direction of the signal \ x Di P° le 

source) will be re-radiated by the reflector e ec or 

in such a way that the re-radiation reaches 

the dipole in anti-phase with the unwanted signal, and therefore diminishes its strength. 

In other words, the presence of a reflector distorts the circular response pattern (or 
radiation acceptance pattern) of a solitary dipole into a pear-shaped lobe pointing 
towards the signal source, with a much smaller lobe pointing in the opposite direction 
(see illustration on page 2.28). It thus greatly improves the directivity of the dipole in 
the required (or forward) direction. 

The addition of a reflector also narrows the frequency response of a half-wave dipole 
so that the combination tunes more sharply to the wanted signal. (This is equivalent 
to improving the selectivity of the resonant tuned circuit which is what the dipole 
aerial essentially is.) A disadvantage of adding a reflector to a dipole is that it tends, 
as you will see, to reduce the centre impedance of the dipole. 



§10] 2.27 

VHF Aerials — The Reflector (continued) 

The degree of accentuation of a wanted signal reaching the dipole by means of an 
in-phase reflection from the reflector is termed the power gain, or forward gain, of the 
dipole-reflector combination. It is expressed in decibels, and is typically of the order 
of 3 to 5 dB. 

The power gain of a dipole-reflector combination is much affected by the spacing 
between the two uprights of the "H". It is maximum with close spacing of about 
0-1 wavelength between dipole and reflector, and falls gradually to about 3 dB with 
quarter-wavelength spacing. 

In the 

arm asr/ai 

of Signal 




Dipole/Reflector Spacing 



supporting Determines Forward Gain 

Mast 

It would seem from this that the closer the spacing, the better. Two other factors, 
however, complicate the problem. If the spacing of the reflector from the dipole is for 
any reason reduced below 0-1 wavelength, power gain suddenly drops very sharply, and 
there is a sudden large drop in the strength of the signal fed to the receiver. Such a 
reduction in dipole/reflector spacing could often occur in a roof-mounted aerial, and 
severe flutter in the picture received would result whenever a strong wind blew in gusts. 
Although power gain is less when dipole/reflector spacing is eased out to between 
0-1 5 A and 0-25 A, there is no such sudden drop when the spacing is momentarily altered, 
and the signal developed at the receiver input is much more stable. 

The second objection to over-close spacing of reflector and dipole is that, while the 
addition of any reflector to a dipole reduces the centre impedance of the dipole and so 
endangers the all-important impedance match with the feeder cable, the closer the 
spacing, the sharper this reduction in centre Z becomes. At quarter-wave spacing, 
the reduction is only from 73 to about 60 ohms, which is still tolerable from the 
matching point of view; but at a tenth-wavelength spacing, impedance at the centre of 
the dipole can drop to as little as 15 ohms. 

Though there are (as you will see in a moment) means of counteracting the effects 
of a centre Z even as unacceptably low as that, you will gather that the mutual spacing 
of dipole and reflector is one of the many factors which have to be taken into account 
when the perpetual compromise involved in achieving a good aerial design is being 
worked out. 

One other technical term is in common use in connection with the simple dipole- 
reflector combination. The front-to-back ratio of the combination provides a measure 
of the accentuated response of the dipole-plus-reflector in a forward direction (i.e., in a 
direction pointing towards the signal source), compared with its response in a back- 
ward direction. An "H" aerial can often provide a front-to-back ratio of 9 or 10 dB — 
high enough to be very useful in discriminating against unwanted signals arriving from 
sources other than the transmitter itself. 



2.28 



[§10 



VHF Aerials— The Reflector (continued) 

Be careful not to confuse forward gain with front-to-back ratio. In the illustration 
below, the response of a dipole mounted by itself is shown as a dotted circle super- 
imposed on the pear-shaped response of a dipole-reflector combination, drawn in 
continuous heavy outline. The dipole alone is, as you know, omni-directional, and 
its value is given (for purposes of the illustration) as unity. 



Response of 
Dipole only 

V-- 

' Reflecto 


\ 


Response of 
Dipole plus 
Reflector 


r ^ \ 


/ Jf /Dipole \ \ 
1 # « 




V 1 




. / 






V 


- 


S 









2 ( 


) 1 

, Forw 
Gai 


1 

arri , 
n 


8 



FORWARD CAIH 



Signal 



and 



FR0HT-T0-8ACK 
RATIO 



The response curve of the dipole-reflector combination is also given for purposes of 
illustration only. A combination could be designed to give comparative figures of this 
order of magnitude, but so could lots of others giving different ones. 

Say, then, that the response of the combination is 1-8 in a forward direction from the 
aerial, and 0-2 in a backward direction. The forward gain is, of course, only 1-8:1, 
and the addition of the reflector has improved the response of the dipole in that ratio. 

The front-to-back ratio, on the other hand, is 1 -8/0-2, or 9 : 1 ; and this ratio expresses 
the relative efficiency of the combination in a forward direction as compared with its 
efficiency in a backward direction. 

The Shape of VHF Aerials— The Director 

It is possible to enhance still further the directive properties of the dipole-reflector 
combination, and to sharpen still more its frequency response, by adding to it in front 
of the dipole with respect to the wanted signal other lengths of continuous rod known as 
directors. 

Directors are similar in appearance to the reflector, and are always fixed in the same 
plane as the dipole-reflector combination and parallel to it. In length, the first 
director (i.e., the one immediately in front of the dipole) is normally made about 5% 
shorter than the dipole itself; and each succeeding director is made about 5% shorter 
than its predecessor nearer the dipole. 

The simplest form of dipole-director combination is the X-aerial of which you see so 
many on the roof-tops of Britain today. This type of aerial is described on the next 
page. 



§10] 2.29 

VHF Aerials — The Director (continued) 

The X-aerial consists of a half-wave dipole plus a director, both bent outwards at an 
angle of 90° and both rigidly supported at their centre (which is also the approximate 
centre of gravity of the array) at a common point. To this common junction is fixed 
the aerial support, down which runs the feeder to the receiver. 

Note that in an X-aerial the dipole is the element furthest away from the signal 
source, whereas in the H-type jt is the element nearest to it. In both aerials, dipole 
size is the same — slightly less than half the wavelength which the aerial is intended to 
receive. 

Pick-up efficiency of the X-aerial is about the same as that of the H-type, but it 
provides a slightly better front-to-back ratio. It is also somewhat the smaller of the 
two, because a director is always some 10% shorter than a reflector, and because the 
bending of the elements makes the array more compact. 

The smaller size, superior rigidity and better mechanical balance of the X-aerial 
make it the easier of the two to erect, and provide a further marginal advantage over 
the H-type. 

Multi-Element Arrays 

When an aerial contains a dipole, a reflector and a number of directors, it is known 
as a multi-element array. A theoretical form of such an array, containing four 
elements only — dipole, reflector and two directors — is illustrated below. (In practice, 
as you will see on the next page, the dipole in a multi-element array is always of a 
different shape.) 

«*««' Dipole Director Ojr , elor ^ 

FOOR'€UMCHT 



I 



Signal 



4rr*y 



0-5A 0'47A 042X 0-38A 

The addition of directors) increases both the forward gain and the front-to-back 
ratio of an H-type aerial by amounts which depend primarily on the number of directors 
added, and on the degree of spacing between them. In theory, the optimum gain 
would be achieved by spacing the directors about a tenth of the wavelength apart; but 
better all-round results are achieved in practice by increasing this spacing to between 
one seventh and one-eighth of the wavelength (0-14A to 0-125A). 

The number of directors used depends on the location of the receiving aerial in 
relation to the transmitter. In areas where the signal is strong and interference low, 
no director at all will be needed ; and the ordinary H-type or X-type aerial will suffice 
to pick up even signals in Band 3 with acceptable efficiency. But as distance from the 
transmitter increases and sources of interference signals multiply, so the number of 
directors needed grows (especially in Band 3); until in fringe areas double aerial 
arrays sometimes containing as many as 12 elements may be required. 



2.30 



[§10 



The FOCDSD 
DtPOU 




VHF Aerials— The Folded Dipole 

The action of a director array is similar to that of the reflector. Energy from the 
incoming signal is extracted by the director(s) and re-radiated so that it arrives at the 
dipole in phase with the signal itself. 

Unfortunately, however, the addition of directors produces a most undesirable 
effect on the impedance of the dipole to which they are attached, reducing the centre 
impedance of the dipole to a level far below an acceptable match with the feeder cable. 

One of the most common ways of overcoming this mismatching (especially in Band 3, 
where the number of directors needed to pick up an adequate signal is often high) is to 
fold over the dipole so that it looks like a 
continuous half-wave element connected in 
parallel at its ends with an ordinary half- 
wave dipole, which in turn is cut and tapped 
to the feeder at its centre point. Another 
way of looking at the folded dipole is to 
regard it as a full-wave element folded back 
on itself with its two sections only some 
50 mm (2") apart, to form as nearly as may 
be a centre-fed dipole. 

The principal advantage of the folded 
dipole is that the division of the current be- 
tween its two sections increases the centre 
impedance of the dipole by a factor of four as compared with the centre Z of an 
ordinary dipole. With a "natural" centre Z of some 300 ohms, the folded dipole can 
therefore be reinforced by a reflector and several directors without this impedance 
being reduced to a level which would present a serious mismatch with the feeder cable. 

For this reason, nearly all multi-element arrays require a dipole of the folded type. 
Its signal-detecting ability is about the same as that of the ordinary dipole, but its 
frequency response tends to be rather broader. 

Multi-element arrays containing folded dipoles are often called Yagi aerials, from 
the name of the man who first introduced 
them. Band 3 signals in fringe areas often 
require multi-element double- Yagi arrays, 
such as that illustrated opposite, to pick 
them up satisfactorily. In such an array, 
two separate aerials are supported by a 
common mast, each aerial complete with a 
reflector, a folded dipole, and 3 or 4 
directors. The aerials are mounted one 
half-wavelength apart, side by side when 
the polarization of the signal calls for 
vertical mounting, and one above the other 
when polarization is horizontal. 

It is very important in double Yagi's that the cables connecting each aerial to the 
common junction should be exactly equal in length. Any significant difference will 
cause the signals from the two aerials to arrive slightly out of phase with one another, 
with resulting loss of receiver efficiency. In the worst case, with the signals reaching 
the junction in full anti-phase with one another, they would entirely cancel out; and no 
signal at all would get through to the receiver. 




§10] 



2.31 



VHF Aerials — Polarization 

You saw in Part 1 that the sound and vision signals used in TV are always trans- 
mitted with a certain polarization, and that this polarization depends on the plane in 
which the electric field of the electromagnetic wave is made to travel outward from the 
transmitter. It is open to the broadcasting authority to arrange for the polarization of 
the transmitted wave to be either vertical or horizontal, depending on the technical 
characteristics of the area to be served. 

Rather more than half the TV programmes broadcast in Great Britain in the VHF 
Bands 1 and 3 are today transmitted with vertical polarization; the remainder with 
horizontal polarization. All UHF transmitters (with the exception of a few "fill-in" 
stations) are scheduled to radiate waves having horizontal polarization. 

The efficiency of a receiving aerial of the dipole type is greatly affected by its 
positioning with regard to the polarization of the signals to which it is required to 
respond. Its efficiency will only be maximum if its orientation corresponds to the polar- 
ization of the signal to be received. In other words, an aerial must always be mounted 
vertically for best reception of a vertically-polarized signal, and horizontally for best 
reception of a horizontally-polarized signal. 

A dipole must also be mounted so that it presents itself broadside on to the direction 
from which the signal is arriving. It thus offers the whole of its length to the 
magnetic field of the oncoming signal, and so allows the maximum number of the lines 
of force in this field to cut it. (You will recall from Basic Electricity, page 1.34, that 
an electric current is caused to flow in a conductor when a magnetic field is moved 
across the conductor.) 



Electro-Magnetic Field 
Surrounding a 
Half-Wave Dipole 




Half-Wave 
Dipole 




Magnetic 'H' Field 



Electric 'E' Field 



VHF Aerials— The Slot Aerial 

You should now learn a little about another type of VHF receiving aerial which 
appears to break the rule about needing to be mounted with a polarization correspond- 
ing to that of the signal it is intended to pick up. It is called the slot aerial. You will 
not often see such an aerial mounted on a roof-top; but it has advantages for the 
"do-it-yourself" enthusiast who wishes to erect a receiving aerial inside a loft, for 
example, or in the attic of his house. 



2.32 



[§10 



VHF Aerials— The Slot Aerial {continued) 

Take a sheet of conducting material and cut out of its centre a rectangular slot 
running horizontally across the sheet. With the proper electrical connections, this 
slot can be made to behave as though it were a vertically-mounted rod-type dipole, 
and can be used either for transmitting or for receiving signals having vertical 
polarization. 

Now turn the sheet through 90° so that the slot lies in the vertical plane, and it will 
behave as if it were a horizontally-mounted dipole. How does this "inverse orienta- 
tion" property of the slot aerial come about, and can it be put to any use ? 

You know that when a correctly oriented half-wave dipole is accepting a signal to 
which it is resonant, a voltage difference is built up between its ends and the current 
flowing in the dipole is a maximum at its centre. It is as if the dipole were a conductor 
surrounded by a non-conducting medium, with an imaginary electrical generator 
connected at its centre. 

Now imagine that your sheet of conducting material has had cut out of its centre a 
slot whose length is equal to that of the dipole {i.e., approximately half the wavelength 
of the signal which it is intended to receive), and whose width is small in proportion 
to its length. Imagine also that this slot, too, has an electrical generator connected 
across it at its centre point. 

If the current created by this "generator" is to flow between its two "terminals", the 
current path must be up one side of the slot, across the top, down the other side, across 
the bottom and back up the other side to where it started. Since the currents on either 
side of the slot are flowing in opposite directions, there must be a potential difference 
between the two sides— which is only what you would have expected, since the 
terminals of the generator are on opposite sides of the slot and will naturally be of 
opposite polarity. 



Imaginary 
Generator 



i 

- I 



"Ti, 



' 2 

i 

i 

/ 

/I 

/ 
/ 

' ;- 




The path of least resistance across the slot obviously exists at its two ends, where 
there is a good conducting connection; and it is at these points that the current reaches 
a maximum value. Maximum resistance, and therefore minimum current flow, is 
found across the centre of the slot. 

Conversely, the voltage distribution across the slot will be a maximum at the centre, 
and minimum at the two ends. 



§10] 



2.33 



VHF Aerials— The Slot Aerial (continued) 

It follows from what you read on the last page that the distribution of current and 
voltage along the half-wave slot cut in the conducting medium is exactly opposite to 
the distribution of current and voltage along the half-wave dipole situated in the non- 
conducting medium. So if a slot aerial is to respond with maximum efficiency to a 
horizontally-polarized wave, it must be mounted vertically if the electrical field of the 
wave is to be able to set up a voltage difference across the slot and the magnetic field 
to create a current along the slot. And the reverse applies when a vertically-polarized 
wave is to be received. 

Because the slot aerial has to be aligned parallel with the magnetic field of the received 
wave (whereas a dipole has always to be aligned with the electric field), the slot is 
sometimes referred to as a magnetic dipole. 

One snag about the slot-type aerial is that the impedance measured at the centre of 
the slot is about 500 ohms, which obviously creates matching difficulties with the 
feeder. One of the simplest ways of creating a good match is as follows. 

The illustration below shows a slot aerial constructed from ordinary chicken wire, 
with dimensions suitable for receiving a signal in Channel 1 . The width of the cut-out 
slot affects the bandwidth of the aerial in much the same way as does the diameter of a 
dipole rod. A slot width of 220 to 300 mm (9" to 12") is generally suitable for 
TV reception. 

Wire Netting 
12mm. Mesh 




The feeder is connected to the slot by stripping off its outer covering of PVC for a 
length equal to about half that of the slot, and then soldering the exposed braiding to 
one side of the slot at short intervals. The centre conductor is then extended and 
soldered to one end of a stiff rod, slightly shorter in length than half the length of the 
slot, which is placed inside the slot so that it runs down its centre parallel to its sides. 
The rod is there fixed firmly in position by means of insulated spacers (typically, thin 
strips of perspex). 

The whole aerial assembly can then be supported from its four corners by strong 
cords attached to nearby rafters or beams. It should be positioned broadside on to 
the transmitter for best reception (though a certain amount of offset may sometimes 
be necessary to reject outside interference or one of the "TV ghosts" which you will 
learn about later in this Section). 



2.34 



[§10 



VHF Aerials — The Slot Aerial (continued) 

In the earlier slot aerials, the conducting "sheet" into which the slot was cut often 
consisted of ordinary 12 mm (£") wire netting, with a thicker piece of wire joining up 
the ragged edges of cut wire to form the outline of the slot itself. This was cheap and 
worked well, but more efficient slots are generally used nowadays. 

The principal use of the slot-type aerial is indoors, where maintenance is easier and 
problems of weatherproofing do not arise — generally in a loft or attic in an area of 
good signal strength. Because it is smaller and needs to be placed horizontally to 
receive a vertically polarized signal, the slot is much more convenient in such a location 
than are the big H or X aerials used to receive signals in Band 1. 

Moreover, the slot aerial has a better (i.e., more directional) polar diagram than has 
the half-dipole, and so better selectivity. This selectivity can be further improved by 
the addition of a reflector element, in just the same way as can the performance of a 
dipole. 

The slot aerial reflector takes one of two forms. It can either be the normal rod- 
type element positioned the usual distance behind the aerial, or it can be another slot 
similarly positioned. The main difference between the two types lies in the way they 
are mounted. The rod-type reflector, being of opposite polarization to the slot, needs 
to be mounted vertically beind a horizontal slot; whereas the slot reflectors, having the 
same polarization as the slot, must be mounted parallel with it. 

The dimensions of the slot reflector, and its spacing from the slot itself, are exactly 
the same as those of the ordinary dipole reflector. 

One big disadvantage of the slot reflector is that it increases the impedance of the 
slot, whereas the addition of a reflector to a dipole, as you know, reduces its centre 
impedance. The addition of a reflector to a slot aerial thus increases the difficulties 
of matching the aerial to the feeder cable running to the receiver. One way of over- 
coming them is to use an aerial known as the folded slot. 







The 
I FOtDiD 
SLOT 



You know that when an ordinary dipole is folded, its impedance is increased by a 
factor of four. When a slot aerial is folded, exactly the opposite happens — its centre 
impedance is reduced by a factor of four. This is most convenient, for the 125-ohm 
impedance of the folded slot is far easier to match to the 75-ohm impedance of the 
feeder than is the 500-ohm Z of the normal slot. Indeed, the loss of signal will not be 
more than 1 or 2 dB if a folded slot aerial is connected directly to a 75-ohm feeder. 

The folded slot is made by placing a second sheet of conducting material inside the 
original slot itself, holding it in position by means of insulated spacers. The method 
of connecting a coaxial feeder to a folded slot is similar to that of connecting it to an 
ordinary slot, but (as you can see in the illustration) a good deal simpler. 



§"0] 



2.35 



The VHF Aerial— The Skeleton Slot Aerial 

Recent experiments on the slot aerial have shown that almost all the conducting 
material surrounding the slot can be cut away — leaving only the skeleton outline of 
the slot itself — without affecting the properties of the original aerial. The logical 
development of this discovery has been the construction of aerials made of ordinary 
dipole-type half-inch rod having the same dimensions and outline as the original slot. 

Such skeleton slot aerials, as they are called, are now commercially available for 
reception in all television Bands. Some are designed with bandwidths wide enough to 
allow the aerial to operate over a complete Band of frequencies; others to operate over 
three adjacent channels only. Bandwidth is varied by altering the width of the slot. 
The naturally high directivity of the slot gives good gain and good rejection of un- 
wanted interference signals. 

One British manufacturer has combined the advantages of the Yagi with those of the 
skeleton slot to produce a composite array having very high gain. An array of this 
type uses the high impedance of the slot to reinforce the sharply-lowered impedance of 
the dipole which results when a reflector and a number of directors are added to it. 



a Double Yagi 

AERIAL WITH 

Skeleton Slot 



Skeleton 
Slot 



Reflectors 




Feeder to 
Receiver 



The illustration pictures an aerial array of this type — one designed to operate over 
three adjacent channels in Band 3. You will observe that the horizontal skeleton slot 
is teamed up with two vertical Yagi's having a total of two directors and eight reflectors 
between them, to receive a vertically polarized wave. Directivity is very high, the 
beam width of the polar diagram being only some 60°. 

The coaxial feeder is matched to the slot at its centre (through a so-called delta 
match which you will be learning about soon). With the high Z of the slot being 
reduced by the low Z of the two Yagi's, an excellent match with the 75-ohm Z of the 
feeder is achieved. It is this good match which gives the array its very high overall 
gain of some 16 dB. 



2.36 



[§10 



The Slotted Cylinder Transmitting Aerial 

It is perhaps worth while to digress a bit at this point, and to return for a moment to 
the transmitting aerial. 

As you learnt on page 1.129, the principle of the slot is used also in transmitting 
aerials, one of its most useful applications taking the form of the so-called slotted 
cylinder transmitting aerial. You may remember the description applied to a trans- 
mitter of this type when the section of the TV mast housing the slotted cylinder aerials 
was said to look "rather like a number of the small corner turrets found in mediaeval 
castles, set one on top of another without their pepperpot roofs". 

Here is what the aerial section of such a mast looks like when seen at shorter range. 



Metal 



SSl^T Cap 



Slot 




the Slotted Cylinder 



TRANSMITTING AERIAL 



Supporting 
Mast 






Coaxial 
Feeder 



The slotted cylinder aerial works just like the other slot aerials already described, 
save that the conducting material surrounding the slot consists of a section of a metal 
cylinder instead of a flat sheet. It is widely used by the BBC for the transmission of 
their VHF/fm sound radio programmes in Band 2, the same aerial being used to trans- 
mit all three programmes in the Band. 

The metal cylinder of a broadcasting aerial of this type forms a natural extension of 
the supporting mast, and has a large number of slots (typically 32) arranged at 90° 
intervals round the cylinder in sets of four placed one above the other. This arrange- 
ment ensures that transmissions can be beamed with equal efficiency to all points of the 
compass. 

Note that the vertical configuration of the slots ensures that the polarization of the 
radiated signal shall be horizontal. The length (or depth, if you prefer it) of all the 
slots is adjusted to half the wavelength of the mid-band frequency in Band 2. 

The comparatively wide-band characteristic of the slot aerial permits the use of a 
single transmitting aerial to radiate the complete band of frequencies (from 88 to 100 
MHz) lying in Band 2. Thus all Band 2 signals can be transmitted from the same 
point, so that a single receiving aerial can pick them all up. But for this, a viewer 
would be put to the expense of having to mount several aerials, all pointing in different 
directions, to pick up all the frequencies radiated in the Band. 



§10] 



2.37 



The UHF Aerial 

Signals in the UHF Bands travel, as you know, on much shorter wavelengths than 
do signals in the VHF range. They therefore require shorter aerial elements to detect 
them. In order to produce a signal voltage across its terminals equivalent to that of a 
VHF aerial immersed in the same field strength, a UHF aerial must therefore be built 
with a higher power gain to make up for its smaller collection area. 

There are at least two other reasons why a UHF aerial system needs to possess a 
relatively higher power gain than does a VHF aerial. The first is that the signal, 
though generally no less strong than the VHF signal when it is transmitted, and though 
it is capable of being transmitted with considerably better directivity, is nevertheless 
much more easily attenuated by passage through the atmosphere. It therefore 
weakens with distance from the transmitter more quickly than does a VHF signal of 
the same original strength. 

The second reason why a UHF aerial system needs relatively higher gain is that the 
aerial needs to deliver to the receiver a signal which is actually stronger than the signal 
which would be acceptable at VHF. All valves, as you know, produce noise when 
they are used to amplify a signal; and the higher the frequency of the signal to be 
amplified, the worse this noise tends to become. So with noise at UHF tending to be 
high, the signal reaching the r.f. amplifier stage in the tuner needs to be about three 
times as high at UHF as it does at VHF. 

With aerial elements which need to be short having to amplify an often weaker signal 
to a higher power, it is obvious that a UHF aerial array needs to be more elaborate 
than its VHF counterpart. The principles on which it works, however, are exactly the 
same; and the extra complexity of structure arises only from the addition of more 
directors, and from the use of a reflector of different and more efficient design. 

By reason of the high frequencies to which it has to resonate, a UHF dipole is only 
about one-tenth of the length of a dipole tuned to a frequency in Band 1 . To take an 
example about midway through the UHF range, a dipole tuned to a frequency in 
Channel 39 (614-622 MHz) needs to be about 230 mm (9") in length, its reflector 
240 mm (9^"), and its first director a little under 220 mm (8|")- long Subsequent 
directors still need to diminish in length by about 5% compared with their immediate 
predecessor nearer the dipole. 

Spacing between reflector and dipole needs to be about 115 mm (4£") at these 
frequencies; that between dipole and first director only about 90 mm (3£"). 



2%y °» . 




J^.v 



4 Typ/c*/ 
UHF ACRIAL 



You will see that, despite their greater complexity, UHF aerials are not physically 
large; and it is quite easy to design arrays which are neither too heavy nor too cumber- 
some to be fixed securely on an ordinary roof-top. 



2.38 [§I0 

The UHF Aerial— The Reflector 

Because of the poor signal-collection efficiency of the short individual elements in a 
UHF aerial array, and because a relatively stronger signal is nevertheless needed by 
the UHF receiver, the problem of gain is of first importance in UHF aerials. Any- 
thing which will give more gain is generally worth adding to the array, even if it brings 
difficulties with it. (These difficulties, of course, mainly affect the correct matching of 
the aerial itself to the feeder cable running to the receiver.) 

The reflector is the element on which chief reliance is placed to achieve more gain in 
a UHF aerial. You will note that in the illustration below the reflector has become a 
four-element affair erected with the same polarisation (horizontal) as a six-element 
director array. This four-element reflector is placed a carefully calculated distance 
behind the aerial, the object being (as with all reflectors) to ensure that the re-radiated 
signal from the reflector should reach the aerial in phase with the signal itself. 



Multiple 
Reflector 




UHF Aerial 

WITH REFLECTOR 



Folded 
Dipole 



Note that the dipole used is of the folded type. An aerial having a high "natural" 
centre Z of its own is always required in a UHF array to offset the impedance-lowering 
properties of the multiple reflector-and-director arrays which are needed to give the 
aerial adequate gain. 

Another type of reflector which is sometimes used to give added gain to a UHF 
aerial is the corner reflector. Two types are shown below — the horizontally and the 
vertically polarised, with differing angles of inclination. 




Folded 
Dipole 



Reflector 



UHF Aerials 

with 

CORNER REFLECTORS 



Reflector 




Folded 
Dipole 



Note that the two halves of the multiple reflector are bent inwards towards the 
folded dipole aerial, which has no directors. The angle of inclination between the two 
halves of the reflector can vary between 60° and 90°. 

The advantage of the corner reflector is that it gives high forward gain — typically 
between 9 and 1 1 dB. 



§10] 



2.39 



The UHF Aerial— The Wire Mesh Reflector 

A more common type of reflector used with the UHF aerial is the wire mesh type. 
Theoretically, the ideal reflector would consist of a flat sheet of metal, of infinite 
dimensions ; for re-radiation of the transmitted electromagnetic waves would, from 
such a surface correctly positioned behind the aerial, be maximum. But the cost of 
such a sheet, and the practical difficulties of mounting it securely on an ordinary roof- 
top, would be great; and it has been found that results almost 90% as good can be 
obtained from a quite small rectangular grid consisting of closely-spaced wire rods, or 
of heavy wire mesh, erected at the correct spacing behind the folded dipole. 

The important point is that the spacing between the rods (or between the vertical/ 
horizontal wires in the mesh, depending on its polarisation) shall not exceed one-tenth 
of the wavelength (0-1 A) of the signal. 



Multiple 
Reflector " 




The 

WiKS MiSH 

Reflector 



The UHF Aerial — Acceptance Bandwidth 

All Britain's TV Broadcasting at UHF has been planned in advance to allow an 
eventual choice of four programmes in every viewing area. At present, only three 
of these programmes are being broadcast; but the eventual pattern will be two 
BBC and two ITA programmes available in most parts of the country. 

In order to save money, it is planned to radiate all four of these programmes from a 
single transmitting station in each viewing area (the technique is called co-siting). 
Frequencies must obviously be chosen for these programmes which will not interfere 
with one another. Channels in each area have therefore been allotted on a wide 
separation basis. In the London area, for instance, the channels allotted are 23, 26, 
30 and 33; and this formula — n, n + 3,n + 7,n+10, where n is the channel number of 
the lowest frequency to be broadcast — will hold good, with small variations, over the 
rest of the country. 

London's four channels thus cover a frequency range of 486 MHz to 574 MHz; and 
this total frequency range of 88 MHz will also be approximately the rule elsewhere in 
Britain. 

It is obviously desirable that a single receiving aerial shall be capable of receiving 
efficiently over all this frequency range. The alternative would be four UHF aerials 
on every roof, and a complicated and "lossy" sharing network to couple all four of 
them into the single input terminal of the receiver. All British UHF aerials are there- 
fore designed to have a more or less even frequency response over the wide bandwidth 
of 88 MHz. Their acceptance bandwidth is therefore broad. 



2.40 



[§10 



The UHF Aerial — Acceptance Bandwidth {continued) 

This broadening of the acceptance bandwidth is generally achieved by increasing the 
ratio of element diameter to element length. Since UHF aerial dipoles are not very 
long anyway (less than 300 mm, or 12", in all cases), this extra thickening results in 
their acquiring a generally "tubbier" appearance than that of their VHF counterparts. 

Two consequences of thickening a dipole should be noted. The first, as you already 
know, is to lower the centre impedance. When the diameter of a 12 mm Q") dipole is 
increased to 60 mm (2^"), to give a typical example, centre Z is reduced from the 
neighbourhood of 73 ohms to about 40 ohms. 

The second consequence of a thicker dipole is that the velocity of the electro- 
magnetic wave travelling through it is still further reduced in comparison with the 
velocity of the same wave when travelling in space. You already know that this loss 
of velocity needs to be compensated by reducing the length of the "half-wave" VHF 
aerial of 12 mm diameter by about 5%. At UHF the corresponding reduction needs 
to be 10%. The "half-wave" UHF dipole of 60 mm diameter thus needs to be cut to a 
length of about 0-45A. 

Some UHF aerials, particularly in the United States, achieve the required broadened 
bandwidth by changing their shape. Typical of these is the so-called conical dipole (it 
should strictly be called the "bi-conical dipole", as you will see from the illustration 
below; but it seldom is). The two cones have a comparatively large diameter at their 
ends (which gives them the broadened bandwidth), but taper to a very small diameter 
at their inner ends (which helps to keep their centre Z high enough to offset the diluting 
effect of directors). 

I A 

r t 





The Conical Dipole 

The conical dipole may either be constructed of hollow cones of sheet metal, or the 
cones may be simulated by a suitable arrangement of thin metal rods so as to offer less 
resistance to wind. 

Another similar shape of dipole, also to be seen in Britain, is the so-called bat-wing 
aerial. With its wings made of flat sheet metal or wire mesh shaped into two elon- 
gated triangles, it has the same properties of wide acceptance bandwidth and ade- 
quately high impedance at the point where it is tapped by the feeder. 



The Bat-Wing Dipole 




§10] 2.41 

The UHF Aerial — Positioning and Alignment 

UHF aerials are, as you have seen, highly directional. Their accurate positioning 
and alignment are therefore of great importance. 

In general, the best position for the UHF aerial is as high above ground as you can 
get it. It should also be positioned well away from all other aerials, for their proxi- 
mity can alter the centre impedance of the UHF dipole, resulting in what looks like a 
reduction of the bandwidth over which the dipole will accept signals. 

Yet here, as always with aerial design and mounting, there is need for compromise. 
For if you fix your UHF aerial as far away from other aerials as you possibly can — say, 
to a second chimney — a greater length of feeder cable will be needed to connect the 
aerial to the receiver. As you will see shortly, all feeder cables tend to attenuate sig- 
nals passing along them; and the longer the feeder, the worse the attenuation. You 
know how important aerial gain is at UHF; so it is important not to impair it by 
putting into the aerial circuit more feeder than is absolutely necessary. 

Exact orientation of the UHF aerial is best done by a professional rigger armed with 
a piece of equipment called a UHF signal strength meter. This is connected to the 
feeder, and the aerial is then slowly rotated until a maximum reading on the meter is 
achieved. 

The service area of a UHF transmitter tends to be more sharply defined at the edges 
than is that of a VHF station; yet conditions for reception at points within the service 
area are far less predictable. 

The very high frequency of the electromagnetic waves carrying the UHF signal cause 
them to move in a more or less straight line, almost as if they were waves of light. 
These direct waves travel in a straight line to the edges of the service area, and then 
continue on into space. Very little of them is diffracted over the edge of the line-of- 
sight horizon to receiving aerials situated below it. 

Within the service area, different factors come into play. The UHF signal, with its 
much higher frequency, is far more easily blocked by large objects — hills, gas-holders, 
large buildings and even the foliage of trees — than is the VHF signal. Aerials situated 
directly behind such objects with respect to the transmitter tend to be shielded from the 
signal by them, and to receive the signal either much attenuated or not at all. 

For the same reason, the walls of a house, and its roof, often prevent the use of an 
indoor aerial for UHF reception, even when the transmitter mast is close enough to be 
seen through the kitchen window. 

Irregular ground within the service area, moreover, often gives rise to standing wave 
patterns (alternate regions of high and low signal strengths), and the vertical or hori- 
zontal adjustment of a UHF aerial by a few decimetres or a few degrees can some- 
times make all the difference between a good picture and a poor one. 

The reason why a high-frequency wave suffers greater attenuation when passing 
through a physical obstruction than does a lower-frequency wave passing through the 
same obstruction is that the wavelength of the shorter wave approaches more closely to 
the electrical length of the molecules constituting the obstructing material. You 
know that when the electrical length of a conductor approaches one-half of the wave- 
length of a signal applied to it, the conductor starts behaving as a series-tuned resonant 
circuit and absorbs energy from the signal. Even the shortest UHF wave is, of course, 
much longer than the molecular length of the obstructing material; but it is a great deal 
closer to that critical dimension than is any VHF wave, and so loses more of its energy. 



2-42 [§ I0 

Television Ghosts 

In addition to undergoing greater attenuation than the VHF wave, the UHF signal 
is also more liable to be reflected off objects (particularly metal ones) in the service 
area, in such a way that the reflected wave interferes with proper reception by an aerial 
to which the signal is also travelling direct. Such reflection is always more pro- 
nounced from objects whose length is equal to, or greater than, A/2 of the signal, so 
that there are many more potential reflectors of the shorter UHF signal than there are 
of the longer VHF one. 

In theory, therefore, the problem of interference from what are known as TV ghosts 
should be much worse at UHF than it is at VHF; and it is worth a brief discussion of 
the ghost phenomenon before you see why in practice this is not true. 

A television ghost is the name given to a second, sometimes watery-looking, dupli- 
cate of the wanted picture which appears on the screen some little way to the right of 
the picture itself. It is produced when the wanted signal reaches the receiving aerial 
twice — that is to say, by two routes, one longer than the other. 

When the difference in length between the two routes is large, the ghost picture 
appears completely separated from the main picture, and it is possible to see two 
entirely separate pictures of the scene side by side. When the difference between the 
routes is small, the ghost picture occurs only slightly to the right of the main picture 
and gives it a fuzzy, un-focused outline. 

In the illustration below, the receiving aerial is situated 18 miles (29 km) from the 
transmitter, and the large metal gas-holder rather further away from it at 20-1 miles 
(32 km). Distance from receiving aerial to gas-holder is 2-83 miles (4-5 km) ; and both 
have an uninterrupted view of the transmitting mast and of each other. 




Transmitter 



Signal £0' 





Reflected 
Signal 



Gasholder 



§10] 2.43 

Television Ghosts (continued) 

The signal radiated from the transmitting mast travels outwards as a broad beam, 
reaching the receiving aerial first and appearing on the picture screen in the normal 
way. A little later, the signal also reaches the gas-holder. Being large and made of 
metal, this reflects the signal — some of it in the direction of the aerial. The aerial, 
having been tuned to accept a signal of this wavelength, obediently picks it up; and in 
due course the reflected signal also is displayed on the screen. 

Since the two signals were received at different moments of time, they will be dis- 
played at different points on the screen. You know that the raster on the picture tube 
is traced out by the scanning beam moving from left to right of the screen, and that the 
received signal is used to modulate this beam. So if two identical signals are applied 
to the picture tube, one rather later than the other, the later picture will be displayed 
on the screen by the side of the first one, but to its right as you look at it from your 
armchair. 

Now for some simple sums, all done in Imperial rather than metric measures 

because that is how they were originally worked out and the principle is the same 

whichever system is used. The signal received direct from the transmitter had to 

travel 18 miles. The signal reflected from the gas-holder had to travel 20-1 +2-83 = 

22-93 miles— a difference of 4-93 miles. Since radio waves travel at 186,000 miles per 

second, the time taken for the signal to travel this extra 4-93 miles must be: 

4-93 4-93 x 10 6 

seconds = . Q , nm (xs = 26-5 microseconds 



186,000 *"•"*"" 186,000 

Now consider the picture tube. The scanning spot traces out one line of the raster 
in a period equal to about 57 microseconds in the British 625-line system, this being the 
effective time-duration of the line after allowance has been made for the blanking 
period. On a 19-inch picture tube, the actual width of the screen is about 16 inches. 
The distance travelled by the scanning beam in a period of 57 microseconds is thus 
16 inches, and the speed of the scanning beam is 16 inches divided by 57 microseconds, 
or 0-281 microseconds per inch. 

With the reflected signal being received 26-5 microseconds after the real signal, the 
scanning beam will travel 26-5x0-281 = 7-5 inches during this delay period. This 
means that the ghost will appear 7-5 inches along the scanning line after (and so to the 
right of) the point at which the main signal is displayed. 

Working the same calculations backwards, you can establish that a 16-inch- wide 
picture tube represents a distance of (186,000 x 57 x 10" 6 =) 10-6 miles. With this 
knowledge it is possible to estimate fairly closely the distance from the receiving aerial 
of an object causing a ghost picture. Identification of the object is thus made much 
easier. 

Ghost signals are of two main kinds, positive and negative. When the signal 
reflected from an object appears at the receiving aerial in phase with the main signal, it 
will add to the main signal carrier and will produce a positive ghost. This means that 
the tonal content of the ghost picture will be more pronounced than that of the main 
picture itself (we are still talking of the British 625-line system, which uses negative 
modulation on the vision carrier). 

If, on the other hand, the ghost reflection arrives at the aerial in anti-phase with the 
main signal, a negative ghost picture will be produced whose tonal content will be less 
pronounced than that of the main picture. 



2.44 



[§10 



The TV Aerial — Acceptance Angle 

You will see that the liability of an aerial to pick up reflected signals is proportional 
to the degree to which it is prone to pick up signals reaching it from the side (from the 
side, that is to say, by reference to the direction of the transmitting aerial). The more 
highly directional the aerial, the less its liability to pick up ghosts. 

You know that a multi-element array has good directional properties. As more 
elements are added to the bare dipole, so the polar diagram of the aerial becomes 
narrower and narrower, and its liability to pick up signals from an unwanted direction 
becomes less and less. The acceptance angle of an array is the angle between the two 
points on the polar diagram (one on either side of the axis) at which signal strength has 
dropped by 50%. The illustration below shows this angle superimposed on the polar 
diagrams of (on the left) a dipole-and-reflector-only array and (on the right) of a 10- 
element Yagi. 





Acceptance Angle 



The narrower beam of the 10-element Yagi is typically some 35°. Aerials used in 
radar and in radio astronomy have beamwidths of a few degrees only. 

A way in which the narrow acceptance angle of a multi-element array can be utilized 
in the rejection of unwanted signals is shown in the diagram below. The solid-outline 
polar diagram (or lobe) is that of the array when the aerial is pointing straight at the 
transmitter and so is receiving maximum signal. Unfortunately, when it does so it 
also picks up strong interference from another source nearly in line with the trans- 
mitter, and the picture on the screen is impaired. 



Reduction in Wanted Signal 



Aerial 




New Alignment 
— Wanted Signal 




— Unwanted Interference 

Reduction in 
Unwanted Interference 

If the array be now made to point a little away from the direction of the transmitter, 
its polar diagram will be that shown in dotted outline above. There will be, as you 
can see, some reduction in the strength of the wanted signal picked up, but a much 
larger reduction in the strength of the unwanted signal. The resulting improvement in 
the signal-to-noise ratio of the picture displayed will more than compensate for the 
small loss of signal strength picked up. 

The "exorcism" of a TV ghost can often be achieved in this way — by turning the 
aerial slightly away from the transmitter but in a direction unfavourable to the ghost 
signal. This technique of adjusting aerial orientation can often be usefully employed 
in the vertical plane, as well as in the horizontal. 



§"0] 



2.45 



Stacked Aerial Arrays 

You have just seen that the directivity of an aerial is increased by the addition of 
more director elements, and that this improvement in directivity carries important 
advantages. Yet you also know that, beyond a certain point, the addition of more 
directors to an array defeats the primary object of the addition — which is to increase 
aerial gain. This happens because the resonant impedance of the dipole is reduced 
more and more as each extra director is added, until it becomes so low that the addi- 
tional gain given by one more director is cancelled out by a greater loss of gain caused 
by increased aerial-to-feeder mismatch. 

There are several techniques used to compensate for this reduction in resonant 
impedance (the use of the high-Z folded dipole is, as you have seen, one of them). A 
rather costly but very efficient one, which has the added advantage of giving excellent 
directivity, is to mount two similar but quite separate aerials side-by-side (or one above 
the other if the polarization of the wanted signal is horizontal) on the same mast. 

Such an assembly is known as a stacked aerial array. A typical UHF variety, such 
as will often be found in fringe reception areas, is illustrated below. 




The two halves of a stacked aerial array are normally mounted one half-wavelength 
apart. Great care must be taken to see that the length of feeder cable running from 
each aerial to the common junction point is exactly equal, so that the signals picked up 
by the aerials shall arrive at the junction correctly in phase with one another. 

If this is done, and if correct matching between the aerials and their respective 
lengths of feeder is achieved, two identical aerials stacked in this way will yield twice 
the gain of either aerial singly, and will also usefully concentrate the acceptance angle 
of the array in the desired plane. 

An aerial array can be stacked with its two halves either one above the other, or side 
by side. The side-by-side arrangement shown in the illustration is much the more 
common of the two because it almost eliminates interference signals coming from 
either side — directions from which most interference signals must obviously originate. 



2.46 [§10 

Aerial Feeder Cables 

The signals collected by the aerial are, as you know, transferred to the receiver by 
means of a length of special cable called a transmission line, or feeder. Many types of 
feeder cable are made, but in Britain only the coaxial type is used for television. 

A coaxial feeder consists of a copper conductor of small diameter running through 
the centre of, but insulated from, a braided copper sheath. It is the coaxial arrange- 
ment of these two conducting elements which gives the cable its name. The sheath has 
a protective covering, usually of PVC, to make the cable waterproof and to protect it 
from abrasion. 

The insulating material used between the two conductors (the core and the braided 
copper sheath) varies. The reason is that no feeder can pass a signal without attenuat- 
ing it to some extent, but that some types of insulation cause less attenuation than 
others. These latter types tend, however, to be more expensive, and are therefore only 
used when signal attenuation needs to be kept to a minimum. 

Signal losses increase, of course, the further they have to travel through a feeder; so 
it is customary to measure the loss in decibels per 100 foot of cable run, and to grade 
different makes of cable accordingly. 

Cable losses also increase with the frequency of the signal being passed, at a rate 
approximately equal to the square root of the frequency. They are therefore more 
serious at UHF than at VHF— yet it is at UHF, you will recall, that loss of signal can 
least be afforded. Cable insulation at UHF needs therefore to be better than it does at 
VHF. 

In Band 1, it is possible to use as insulating material a quarter-inch-thick solid 
insulating sheath of polythene. This is cheap, but the polythene absorbs part of the 
signal. 

Braided Copper Sheath 
llllllllllllllllll (Outer Conductor) 

PVC Covering 




Copper 
Centre 
Conductor ^^^s*®^" "\ Polythene 

Insulation fflllillllllillllllll 

In cables used at higher frequencies, it is usual to keep the two conductors insulated 
from one another by means of a thin strip of polythene wound spirally round the 
centre conductor. Such a cable is more expensive to manufacture, but it contains less 
polythene so that less of the signal is absorbed. A spongy type of synthetic material, 
of cellular construction, is sometimes used as an insulator in UHF feeder cables. 

Cable losses tend to decrease when the thickness of the conductors used is increased. 
In areas of poor signal strength it is therefore often useful (though always more expen- 
sive) to use cables of considerably larger diameter. 

The fact that cable losses increase with the frequency of the signal means that a 
feeder good enough to pass VHF signals is seldom good enough to pass UHF without 
unacceptable attenuation. A joint VHF-UHF downlead, provided the cable is of 
good quality, is sometimes enough in areas where the UHF signal is very strong; but it 
is usual for British Dual-Standard receivers to have two input sockets, with separate 
feeder cables (the UHF one of superior quality) running down into them from the 
respective aerials. 



§10] 



2.47 



Aerial Feeder Cables (continued) 

UHF signal strength can be increased in areas of poor reception by connecting into 
the aerial circuit special transistorised UHF amplifiers. These are capable of stepping 
up a weak signal, and so improving the signal-to-noise ratio in the first stage of the 
receiver. They are usually fitted near the receiver end of the feeder, where they can be 
conveniently powered either from the mains or by a small battery of their own. 

You already know how important the characteristic impedance (Z ) of the feeder 
cable is in aerial construction. In a coaxial cable, the Z is determined by the ratio of 
the inside diameter of the copper braiding to the diameter of the centre conductor. It 
is, as you have seen, standard practice for these conductors to be so made that the Z 
of all feeders approximates to 75 ohms. The impedance of the aerial itself is then, by 
one means or another, made to conform to this standard impedance. 

At the receiver end, the 75-ohm impedance of the feeder is matched to the relatively 
high input impedance of the r.f. amplifier valve in the tuner, usually by means of a 
small coupling transformer. 

The feeder is normally connected to the centre-fed aerial dipole in the manner shown 
in the illustration below, one quarter-wave section of the dipole being fed to the centre 
core, the other to the copper braiding forming the outer conductor. The joint is made 
rigid and weatherproof by being enclosed in a container sometimes called a gland. 
This is a small junction box whose lid is fitted with a rubber gasket. 



Gland 



X, 



Dipole 
Aerial 




To the TV Receiver 



Two other methods of connecting the feeder to the dipole are sometimes used. 
They are closely similar, and are called respectively T-matching and delta-matching. 
The object of both techniques is to overcome the mismatch which occurs between the 
centre Z of the dipole in a multi-element array and the standard impedance of the 
cable. 

You know that the impedance distribution along a dipole rises progressively from a 
low figure at the centre (it can be as low as 15 ohms in a multi-element array) to a 
figure of several thousand ohms at its ends. Somewhere along each arm of the dipole 
there will be a point at which the impedance is exactly equal to the Z of the feeder. If 
the two wires leading to the conducting elements of the cable are connected to these 
points, a perfect match will result. 

The connection is usually made through a pair of rods fixed to the dipole at the 
correct matching points. The shape of the resulting connection resembles either the 
letter T (see left-hand sketch below) or the Greek letter delta. 



E 



T MATCHING 



|75n 



c 



j-KT' 



DELTA MATCHING 



75 n 



2.48 



[§10 



The Unipole, or End-Fed Dipole 

Another method of overcoming the problem of securing a good match between 
dipole and feeder has been successfully tried by at least one British manufacturer. 

In this method, the dipole no longer consists of two quarter-wavelengths of rod 
tapped to the feeder at their centre point, but of a single length of rod, half a wave- 
length long, tapped to the feeder at one of its ends. The method of operation of such 
an end-fed dipole, or unipole, can be followed in the diagram below. 



Dipole 



\ 



Director 



n 



Reflector 



Insulator 



Aerial 
Mast 




4- Coaxial Stub 



To Receiver 



Short-circuit 

between Inner and Outer 

Conductors of Stub 



THE UHIPOU or £M>-M> DIPOU 



The end of the dipole (where the impedance, you will recall, is several thousand 
ohms) is connected to the centre conductor of a special piece of coaxial cable whose 
length is exactly equal to one quarter- wavelength of the desired signal, and whose inner 
and outer conductors are connected (and therefore short-circuited) at the end furthest 
from the dipole. 

A A/4 feeder so connected behaves like a A/4 stub; and (as yoiHearnt on page 4.55 of 
Basic Electronics and in the first Part of this present Series) it is a characteristic of such 
a stub that it has a very high value of impedance at the open-circuited end. This end 
of the stub therefore presents a good match to the high impedance at the end of the 
dipole. 

At the other (short-circuited) end of the stub, impedance is of course zero ; so you 
have a situation in which Z rises along the length of the stub from zero to a high value. 
If, therefore, the feeder cable is tapped into the stub at the exact point along its length 
at which a perfect impedance match is available, maximum transfer of signal will be 
achieved. 

Tapping is done by removing the PVC covering, the outer conductor and the poly- 
thene insulation from the stub at the desired point, and connecting the centre conduc- 
tor of the feeder to the exposed centre conductor of the stub. The severed ends of the 
outer conductor of the stub are re-connected across the gap; and the outer conductor 
of the feeder is connected to the centre conductor of the stub at its short-circuited end. 

The points at which the stub is connected to the end of the dipole, and the feeder to 
the desired point along the stub, are usually encased in insulated weatherproof glands 
to give protection and to ensure maintenance of electrical connection. 



§10] 



2.49 



The Diplexer 

The British Dual-Standard TV receiver has two input sockets only. One of them is 
reserved for the feeder coming from the UHF aerial. The other has to accept all 
VHF signals, in Band 3 as well as in Band 1. Since separate aerials are used to detect 
the VHF signals arriving in these two Bands, some arrangement is needed for com- 
bining the two signals into a common feeder leading into the receiver. 

Combination can be effected either at the aerial end of a common downlead from 
both aerials, or at the receiver ends of two separate downleads. In practice, it is 
generally less expensive to combine the signals near the aerials. 

You might think that the leads from two separate aerials could be combined any- 
where, and without difficulty, simply by connecting the leads in parallel to the receiver. 
The trouble about this solution is that the signals from one aerial would always be 
getting into the aerial circuit of the other, upsetting the characteristic impedances of 
both circuits and impairing reception in both Bands. Combining must be done in 
such a way that each aerial circuit presents a very high impedance to signals from the 
other. 

This is the job of the diplexer — a device which is also sometimes known as a cross- 
over network or (for obvious reasons) as a combining unit. 



Band 1 (+- 
Input V 
Socket 




(•) Band 3 
^ Input 
Socket 



the wpuxcr 

CIRCUIT DIAGRAM 



Diplexer units commonly take the form of fully shielded metal boxes slightly larger 
than a match-box. Each has three connecting points for coaxial feeder cables. 
These points may be used (as you will see in a moment) either as two inputs and an 
output, or as one input and two outputs. 

Diplexers each contain essentially two inductors and two capacitors connected in 
two LC pairs in such a way as to form two frequency-selective filters, one low-pass to 
let the lower-frequency Band 1 signal through to the output socket, the other high-pass 
to let the Band 3 signal through. The best way of understanding how the niters work 
is to read what follows in conjunction with the double illustration on the next page. 
This shows, at the top, the effective circuit of the diplexer when a Band 1 signal arrives; 
and, below, the effective circuit when a Band 3 signal arrives. 

Take the Band 1 signal first. As it leaves its input, it "looks into" two possible 
paths. One is through C 1; connected in parallel ; the other is through L lf connected in 
series. The values of these components are so chosen that L x offers a lqw reactance to 
the Band 1 signal and C x a high reactance. The signal, a's always, takes the path of 
least resistance, and passes through 1^. 

On emerging, it again "looks into" two possible paths — C 2 and the common output 
socket. The value of C 2 is chosen to give it a high reactance to a signal of Band 1 
frequency, so the signal tends to choose the easier path to the output socket. Any part 
of it which does manage to struggle through C 2 is kept out of the Band 3 input by 
giving L 2 such a value as to offer very low reactance to a Band 1 signal. The remain- 
der signal thus chooses this path, and is "shorted" to earth. 



2.50 



[§10 



The Diplexer (continued) 

Now consider the Band 3 signal. On leaving its input socket, it confronts two com- 
ponents, C 2 in series and L 2 in parallel. But the value of C 2 , which offered high 
reactance to the Band 1 signal, now (by careful selection of its value of capacitance) 
offers low reactance to a signal of Band 3 frequency; and L 2 , previously so ready to 
pass the remnant of the low-frequency Band 1 signal, now (by equally careful selection) 
strongly opposes the passage of a Band 3 signal. 

The signal therefore takes the easy route through C 2 , and is again confronted with 
two possible paths — an easy one to the common output socket, and a much harder one 
through a now-high-reactance L x backed up by a now-low-reactance C x waiting to 
pass any surviving signal to earth. 

In short, C x and L x function as a low-pass filter, and C 2 -L 2 as a high-pass filter, to 
all signals of Band 1 frequency; while for signals in Band 3 the roles of the two LC 
pairs are reversed. 



Band 
Input 
Socket 



-Signal Path 




0) Band 3 
Input 
Socket 



EFFECTIVE DIPLEXER CIRCUIT 
when a Band 1 signal arrives 



Band 1 

Input 

Socket 



<£ 



C1 

(= Low 
Resistance) 



L1 



-AAAAr 

(- High 
Resistance) 



/" 



Signal Path 



(*> 



r 



*-C2 

(=Low 
Resistance) 



Common 

Output 

Socket 



$ 



»L2 

(= High 
Resistance) 



Band 3 

input 

Socket 



EFFECTIVE DIPLEXER CIRCUIT 
when a Band 3 signal arrives 



Go back now to that bit on the last page in which you read that a diplexer could also 
be used as a single input feeding a signal of two different frequencies to two appro- 
priate outputs. How could the diplexer be used in this role ? 

In the older types of receiver designed for VHF only, two input sockets were some- 
times provided — one for Band 1 signals, the other for Band 3. Owners had a choice 
of leading two separate downleads from the appropriate aerial into the appropriate 
socket; or they could combine the signals near the aerials, pass them through a single 
downlead, and then split them again near (or inside) the receiver itself. 



§10] 



2.51 



The Diplexer (continued) 

The splitting of the signals mentioned at the foot of the last page was achieved by 
means of another diplexer, having the role of one input and two outputs. This 
diplexer was mounted either outside the receiver with the signals being fed after split- 
ting to two separate sockets in the receiver, or inside the receiver itself with the still- 
combined signals arriving through a single socket. 

Work out for yourself the path of a Band 1 or Band 3 signal arriving at the Common 
Output Point in the illustration on page 2.49, and making its way through the appro- 
priate low-pass or high-pass filter to its exit at one of the Input Sockets. You will see 
that it is again a question of choosing values of L and C which will offer the correct 
amounts of reactance or impedance to the signals of different frequency. 

Note, by the way, that though a single downlead has obvious economic and aesthe- 
tic advantages over a double downlead, and is easier to maintain, it will lead to some 
attenuation of the signal by reason of the small losses which are inevitable in the com- 
bining and re-separating processes. 

The Triplexer 

This is a rather more complicated form of diplexer used for passing the signals 
derived from three aerials into a single feeder, or vice versa. It is used in TV receivers 
which have separate input sockets for Band 1 and Band 3 signals, and which can also 
receive the BBC's sound radio transmissions at VHF in Band 2. 

The circuit diagram of a typical triplexer is shown below. The essential addition, 
you will see, is a band-pass filter. A filter of this type offers minimum opposition to 
frequencies lying within a certain band of frequencies only. To all other frequencies, 
whether higher or lower than this favoured band, the opposition it offers is very high. 

["High" Pass | 



Band 3 
Input 



£j) Band 1 




Tl>€ TRIPUX6R 

CIRCUIT DIAGRAM 



£#) Band 2 
^ Input 



The band-pass filter components in the triplexer will be given values such that the 
filter will offer minimum impedance to all frequencies in Band 2 (87-5 MHz to 100 
MHz), and maximum impedance to all frequencies outside this range. The high-pass 
filter and the low-pass filter behave just as they do in the diplexer; and it is not difficult 
to see how the three filter circuits can be used to isolate the three aerial circuits from 
one another and yet allow their individual signals to pass to a common output. 



2.52 



[§10 



Home-Made Aerials 

Home-made aerials are best erected indoors, in a loft or attic, where maintenance is 
easy and problems of weatherproofing do not arise, and where there is no need for 
extra-rigid elements or mounting fixtures to combat the effect of high wind-pressures. 
Even the take-off thrust of a starling can have disastrous effects on an insecurely- 
mounted out-door aerial; and it is no fun (and rather humiliating) having to clamber 
over brittle tiles on a borrowed ladder to repair an aerial dislodged by so trivial an 
accident! 

So put up your aerial indoors whenever you can. Given adequate lighting to show 
you what you are trying to do, you'll be able to erect it and maintain it in warmth and 
comfort for ever after. . . . 

There is often no need for rigid aerial elements when you are putting up an indoor 
aerial. Appropriate lengths of coaxial cable stretched between two rafters and correctly 
orientated to the wanted signal will often serve as excellent dipoles, reflectors and 
directors; and the spacing between them can be brought close to the ideal 0-1 A without 
fear of picture flutter being caused by wind. One essential precaution is to see that 
the rafters used are really dry, so that they can give adequate insulation. 

If you decide to use rigid aerial elements after all, they can be mounted on a simple 
wooden boom, which will not rot or distort since it is not going to be exposed to the 
weather; and this boom is then easy to orientate towards the signal. 



Dipole 




A DO-IT-YOURSELF 



BAND / A6RIAL 



Wooden Support 



75-ohm 
Feeder 



Of course, if you happen to have a horizontal rafter running in the right direction, 
you can fix your aerial elements to that and dispense with even the boom. 

By and large, you can say that the stronger the signal from the transmitter reaching 
your attic, the less elaborate will your home-made aerial need to be. Never forget, 
though, that it is much better to have an aerial with too much gain than one with too 
little; so do not shrink from adding to your aerial further correctly-spaced elements of 
the correct length if you find that the picture you are getting is less good than it should 
be. 



§10] 



2.53 



Home-Made Aerials (continued) 

Pictured below is a simple home-constructed aerial which could be used to pick up 
transmissions in Band 3 and even (in particularly favourable circumstances) at the 
lower-frequency end of Band 4. 



Folded 
Dipole 




Reflector 



A HOMEMADE AERIAL: 

Bands 3&4 



■ 75-ohm Feeder 



All the elements which affect the performance of an outdoor aerial — signal fre- 
quency, distance from transmitter, height of aerial above ground, the presence or 
absence of hills, big buildings or large sheets of open water and so on — all these affect 
equally the performance of an aerial erected indoors. Additionally, the indoor aerial 
can be affected by the nature of the material forming the roof, and by metal guttering, 
water-pipes, storage tanks and chimney-stacks situated in the "wrong" direction from 
the aerial with respect to the path of the incoming signal. 

Generally speaking, the comparatively long-wave transmissions in Band 1 are least 
affected by structural details of the house itself, but more care is needed in the siting of 
a Band 3 array. Try always to ensure that your aerial has an uninterrupted view to- 
wards the distant transmitter — assuming, as you normally can at VHF frequencies, 
that year roof itself is "radio-transparent". 

At UHF frequencies, however, this last assumption is much less true; and it is only 
rarely that very favourable conditions make the erection of an indoor UHF aerial 
worth while. Luckily, the much smaller size and weight of these aerials make their 
erection out-of-doors a comparatively simple operation for the keen "do-it-yourself-er". 

Remember that final adjustments to the mounting position and attitude of any 
aerial, indoor or outdoor, carried out on a trial-and-error basis, will often give you a 
better picture than the one you had to begin with. It will frequently happen that the 
aerial alignment giving you the strongest signal will not also give you the best signal, 
because the theoretically correct alignment may be the very one which makes the aerial 
most liable to pick up a strong interference signal. 

Remember also, once you have decided on the correct alignment for your aerial, to 
try tilting the array up and down slightly to see if a more or less vertical (or horizontal, 
as the case may be) inclination will bring about a better picture. 



2.54 [§10 

Home-Made Aerials (continued) 

If you are making your own aerial elements, they will need to be cut to the correct 
lengths. The formulae which follow will help you to do this. In them f(MHz) is the 
frequency of the desired signal in megaHertz; but this frequency can be either the fre- 
quency of the vision signal or the geometric mean of the frequencies of the sound and 
vision signals depending on the degree of importance you attach to maximum clarity 
of picture signal on the one hand, or to a good balance of sound and vision on the 
other. 



Q Length of Dipole (in feet) = 

Q Length of Reflector (in feet) = 

@ Length of First Director (in feet) = 



468 
f(MHz) 

498 
f(MHz) 

450 
f(MHz) 



Subsequent directors, as you know, should diminish in length by 5% of the length of 
their immediate neighbour nearer the dipole. 

In the case of a folded dipole, dipole length is measured from the middle of the cur- 
vature at one end, round the unbroken side, to the middle of the curvature at the other 
end. 

Last Words on Aerials 

Aerials of the set-top type, which extend out of the top of a receiver like the an- 
tennae of a very large insect, are only useful in localities of high signal strength; for 
they can only be approximations to an accurately built and properly erected array. 
The advantage of receivers equipped with them is, of course, mobility, for there is no 
need to plug them into any feeder cable. But the price paid in impairment of picture 
signal is generally a high one. 

Do not be surprised if you meet aerials whose performance in practice contradicts a 
certain amount of what you have learnt in this Section. There is, for instance, a Band 
3 aerial erected on high ground near Tunbridge Wells which gives an acceptable picture 
of a BBC 2 programme without there being a UHF aerial of any kind on the roof. 
The Band 3 aerial is of quite the wrong electrical length to resonate to the UHF signal, 
and much of the latter's strength is inevitably dissipated by the standing waves set up 
along the dipole when the signal is received. 

But the strength of the signal reaching the aerial is so great in this particular locality 
that it can produce a fair picture on the screen even after severe impairment. ("The 
signal we get from the Crystal Palace is so strong", says the owner with grateful over- 
statement, "that you could pick it up with a piece of wet string!") 

Aerial performance, in fact, varies with a good many factors, but by far the most 
important of them is signal strength at the point of erection. You have seen how 
widely this strength can vary at different points within the service area, especially at 
UHF. Do not be afraid, therefore, to experiment boldly if the officially "correct" 
rules fail to give you the results you hoped for when you are mounting a new aerial in a 
strange neighbourhood. 



§10] 



2.55 



REVIEW of the Receiving Aerial 

The task of a TV receiving aerial is to extract from the sound and vision signals of the 
desired transmission the energy required by the receiver to enable it to reproduce audibly 
the sounds uttered in the studio, and to display on the screen a satisfactory picture of the 
action taking place in the studio or elsewhere. 

The aerial performs this task by responding to signal frequency. It responds well to 
all signals whose frequencies lie within the narrow band separating the sound and vision 
signals, but poorly to signals of all other frequencies. 



An aerial is essentially a series-resonant 
circuit containing R, L and C. The current 
flow induced by a signal striking such a cir- 
cuit is a maximum when the signal is at the 
resonant frequency of the circuit. Aerials 
can thus be made to resonate to signals of 
different frequency by changing the amounts 
of R, L and C in the aerial circuit. 






The essential part of a TV aerial designed to pick up signals in the VHF range is a 
hollow metal rod some 12 or 13 mm in diameter whose length is equal to a little less than 
one-half of the wavelength of the desired signal. Such an element is known as a half- 
wave dipole. 



The distribution pattern of current and 
voltage along a dipole remains constant 
irrespective of the strength of the signal At 
either end of the dipole voltage is maximum 
but current minimum; and at the centre of 
the dipole current is maximum and voltage 
minimum. It is therefore usual for a dipole 
to be tapped for its energy at its centre point. 



Two-Wavelength 
Dipole 




Since a half-wave dipole is essentially a series-tuned circuit in which current flows 
under the impulse of alternating voltages, it must possess impedance. This impedance 
varies at every point along the dipole. It falls to an approximate value of 73 ohms at the 
centre of any dipole provided that the ratio of dipole diameter to dipole length is kept very 
small. 



2.56 

REVIEW of the Receiving Aerial (continued) 



[§10 



The receiving efficiency of a half-wave dipole is increased by the addition of a reflector 
and/or of one or more directors. These elements increase the ability of the dipole to pick 
up signals coming from a desired direction, so improving its directional properties. 



I 



A reflector is a length of rod, usually a 
metal tube of the same diameter as the dipole 
but about 5% longer. Fixed between 0-15 
and 0-25 of the wavelength behind the dipole \ 

with respect to the direction of the signal, it 

re-radiates some of the energy of the signal back to the dipole so that it arrives there in 
phase with the signal itself, and so adds to it. The degree to which the wanted signal is 
amplified in this way is termed the power gain, or forward gain, of the dipole-reflector 
combination. 

The addition of a reflector also diminishes the response of the dipole to signals arriving 
from behind it with respect to the direction of the signal, and the front-to-back ratio of 
the dipole-reflector combination provides a measure of its improved response us a for- 
ward direction compared with its response in a backward direction. 



The director is a length of rod about 5% shorter than the dipole, placed between one- 
seventh and one-eighth of the wavelength in front of it with respect to the direction of the 
signal. Its effect is to increase the directional properties of the dipole and to sharpen its 
frequency response. 

Further directors may be added in front of R ., lector . 

J Ketiector ni „„| D Director „. . A 

the dipole to improve its performance still 
more. Each should be some 5% shorter 
than its predecessor. An aerial array con- 
taining a dipole, a reflector and a number of 
directors is called a multi-element array. 



nvitcuMem 

Signal 



The addition of a reflector and/or directors to an aerial diminishes its centre impedance 
and so risks a mismatch with the feeder cable to the receiver, with its Z of about 75 
ohms. One way of reducing this danger is to fold the dipole over on to itself, and to tap 
it at the point where its ends nearly meet. The centre Z of a folded dipole is considerably 
higher than 75 ohms and so can stand reduction by the addition to the array of more 
aerial elements. 

A multi-element array containing a folded dipole is often called a Yagi aerial. 



The efficiency of an aerial will only be maximum if its orientation corresponds to the 
polarization of the wanted signal. An aerial must always be mounted vertically for best 
reception of a vertically-polarized signal, and horizontally for best reception of a hori- 
zontally-polarized signal. 



§10] 



2.57 



REVIEW of the Receiving Aerial (continued) 

Other types of aerial used at VHF are the 
slot, the folded slot and the skeleton slot. 
All present problems of good matching to the 
characteristic impedance of the feeder cable; 
but once these have been overcome, slot-type 
aerials provide good forward gain combined 
with manageable size. A multi-element 
array consisting of a double Yagi and a 
skeleton slot aerial, designed to operate over 
three adjacent channels in Band 3, has ex- 
cellent directivity and an overall power gain 
of the order of 16 dB. 



a Double Yagi 

urim. wmi 

Skeleton Slot 




UHF aerials need to produce a higher power gain than do VHF aerials because their 
A/2 elements are shorter and so have lower signal-collection efficiency, because the signal 
reaching the aerial is often weaker, and because a stronger signal is nevertheless needed 
in the receiver to overcome the higher noise produced by valves operating at ultra-high 
frequencies. UHF aerials therefore need to be more elaborate; but the small size of 
their elements makes them easy to mount and maintain. 



The need for high gain in UHF aerials is 
often met by adding several directors to spe- 
cially designed reflectors. Examples of the 
latter are four-element reflectors and wire 
mesh reflectors. 

Another type, the corner reflector aerial, makes use of a folded dipole with no direc- 
tors. It gives good forward gain. 




All British UHF aerials are designed to have a nearly even frequency response over the 
wide bandwidth of 88 MHz, to make them capable of receiving all the four UHF pro- 
grammes planned for every viewing area. 

Methods of giving them the broader acceptance bandwidth made necessary by this 
decision include increasing their diameter relative to their length, and the conical, simu- 
lated conical and bat-wing types of dipole. 



2.58 

REVIEW of the Receiving Aerial {continued) 



[§io 



Television ghosts are seen on the picture 
screen when the signal is received by the 
aerial twice — once direct from the trans- 
mitter and again, a few microseconds later, 
after being reflected from a large object like 
a gas-holder. If these two signals arrive in 
phase with one another, the ghost will be a 
positive one; if they arrive out of phase, it 
will be negative. 




UHF aerials, with their greater directivity (narrower acceptance angle), are less liable 
to pick up ghosts than are VHF aerials, provided they are mounted rigidly and securely. 
Ghosts can often be "exorcised" by turning the aerial array slightly away from the trans- 
mitter in a direction unfavourable to the ghost signal. 



Aerial feeder cables are usually of the coaxial type, and are made with a characteristic 
impedance of about 75 ohms. It is essential to ensure that the impedance of the dipole 
at the point where it is tapped to the feeder is also about 75 ohms, for a mismatch will 
result in part of the signal being lost in the feeder on its way to the receiver. 

Methods of preventing a bad mismatch include the T-matching and delta-matching 
techniques, and the tapping of a unipole at one of its ends with the aid of a quarter-wave- 
length stub. 



The diplexer is a device for feeding into a common socket signals of different frequency 
while keeping each signal out of the aerial circuit of the other and so upsetting its charac- 
teristic impedance. It consists essentially of a high-pass filter plus a low-pass filter. 

By the addition of a band-pass filter, the device becomes a triplexer, which can be used 
for passing the signals derived from three aerials into a single feeder, or vice versa. 




EFFECTIVE MH.EXER CIRCUIT 
hen a Band 1 signal arrives 



Band 1 
Socket 



$" 




EFFECTIVE DHHEXER CIRCUIT 
when a Band 5 signal arrives 



§11: THE TUNER 



2.59 



The tuner is the first major component which the incoming signal from the aerial 
encounters after it enters the receiver. 

It is also the most important component in the entire receiver. Practically any 
other stage in the receiver can be working "below par", and you will probably get a 
picture of sorts on your screen. But if the tuner is wrong, you will get nothing 
intelligible at all. 

For reasons which you will see, the tuner has to be kept well away from the other 
components in the receiver. In the Dual-Standard Set, the oblong metal boxes con- 
taining the two tuners (VHF and UHF) are situated in the receiver chassis directly 
behind the Channel Selection knob on the front panel. The box containing the VHF 
tuner is always the bigger of the two. 

Because of the different order of electronic problems raised by frequencies in the 
UHF range, the two tuners need to be quite different; but there is some use of circuits 
in the VHF tuner to help out with special difficulties encountered in the UHF tuner. 

The function of the tuner is, of course, common to both systems. It is to select the 
sound-and- vision signal required by the viewer from the group of such signals present 
in the channel covered by the aerial; to raise the amplitude of this signal to a usable 
level well above the inherent noise-level of the receiver; and to convert the sound and 
vision frequencies of the signal into the intermediate sound and vision frequencies of 
the receiver. 

(The tuner section of a TV receiver, by the way, also has an unofficial name. It is 
often known simply as "the front end".) 

Begin, now, with the VHF Tuner used in the British Dual-Standard Set, and look at 
a simple schematic diagram of the blocks which go to make it up. 



Aerial 




Tuned 
Circuit 




Tuned 
Circuit 




Tuned 
Circuit 




\ 


/ 


















Step-up 
Trans- 
former 




RF Amplifier 




Mixer 


Outpu 




















i 


i 




, 


i 






AGC Feedback from 
Video Amplifier 


Local 
Oscillator 





BLOCK SCHEMATIC *f the VHF WHi* 






2.60 [§M 

The VHF Tuner 

The first stage in the VHF tuner is a small step-up transformer whose function is to 
raise to a more usable level the voltage of the very faint signal coming through the 
coaxial cable from the aerial. The stepped-up signal is then passed through a tuned 
circuit to the second, more important stage, the r.f. amplifier. 

The function of the r.f. amplifier stage is to raise the amplitude of the wanted signal 
to a level high enough to make it usable after it has been mixed with the signal which 
will be applied to it from a local oscillator in the next stage. What sort of valve can be 
used for this amplification ? 

The obvious type to try would seem to be a pentode, whose amplification factor you 
know to be high and whose anode-to-grid capacitance is very small. Unfortunately, 
however, the number of electrodes contained within a pentode cause it, at the high 
frequencies used in TV, to generate an amount of electrical noise which is often great 
enough to obliterate the wanted signal altogether. 

A triode, which has a much lower noise factor, has to be used instead ; but it too has 
disadvantages. Not only is the gain of a triode at high frequencies poor. It also has 
an anode-to-grid capacitance'which, when the valve is operating at high signal frequen- 
cies, can become large enough to stop the valve acting as an amplifier at all, making it 
burst into oscillation instead. 

For a fuller explanation of the Miller Effect by which this phenomenon is caused, 
you are referred to pages 2.43 and 2.44 of Basic Electronic Circuits. What happens, 
briefly, is this. You know (Basic Electronics, Part 2) that the anode-to-grid capaci- 
tance — C aB — of a triode causes feedback of voltage from the anode to the grid, and so 
makes it appear as if the anode load is affecting the input impedance of the valve. 
This change in input impedance is much increased once the triode starts to conduct — 
the effective value of the C ag of the valve becoming 1 + A times its physical value, 
where A is the voltage gain of the valve. 



When the Triode 

is CONDUCTING 

Apparent' 8 c * 9 (*+*) 



HT 



o— 
o— 


\ 

r 


1 

cfa 
c-aC 

T APPARENT 
* 




V 


OUT 


-o 



With a multiplier of this order at work, the amount of voltage fed back to the grid 
soon becomes so large that the valve bursts into oscillation, and ceases to amplify at all. 



§11] 



2.61 



The VHF Tuner— The Cascode Amplifier 

The range of frequencies over which a triode can be used as an amplifier can be 
largely extended by earthing its grid, and by applying the signal to its cathode instead. 
The grid now acts as an earthed screen between anode and cathode. Capacitive 
coupling between anode and grid, and therefore the risk of instability, is thus almost 
completely removed. 

The illustration below shows the different connections of a normal triode amplifier 
with its cathode taken to earth, and of the so-called grounded-grid configuration 
described above. 



H.T. (+) 




-r- <$ ^ Outp ut 



From 
Aerial 



From Aerial 




H.T. (+) 



Signal I — 
Input Circuit 



The GROUNDEDCATHODE Amplifier The GROUNDED GRID Amplifier 



Here again, however, an apparent remedy develops its own disease, and turns out to 
be only a step on the way to a cure. When a valve is connected as a grounded-grid 
amplifier, its anode current must flow through the signal source, since this is connected 
in series with the cathode. The result is a very low input impedance, which causes 
serious damping of the signal and loss of sharpness of tuning. Both the frequency 
response and the gain of the valve are reduced, and selectivity becomes poor. 

A solution has been found by using two valves in the r.f. amplifier stage of the VHF 
tuner, connected in such a way that the resulting circuit combines a high input impe- 
dance with the ability to amplify signals having a high operating frequency. Such a 
circuit configuration is called a cascode. Until quite recently, cascode amplifiers have 
dominated the design of VHF tuner circuits ; and they are still widely used in sets of 
British design. 



2.62 



[§H 



The VHF Tuner — The Cascode Amplifier {continued) 

The cascode amplifier consists essentially of an ordinary (grounded-cathode) triode 
— having another triode connected, in the grounded-grid configuration, as its anode 
load. In the circuit below, the grounded-cathode valve is V^ V 2 , its anode load, has 
its grid effectively earthed (as far as the signal is concerned) because the reactance of 
the capacitor C 2 is very low. — r h.t. (+) 



The BASIC CASCODE CIRCUIT 



r 



Output 




Aeria 

Feeder 

Cable __ 

The circuit arrangement of V x gives it a high input impedance, thus matching it with 
the signal reaching it from the step-up transformer T ± connected between the coaxial 
feeder from the aerial and the valve. The secondary of T,. is tuned (by Q and the 
input capacitance of Vi) to the frequency of the desired signal, and therefore behaves 
as a parallel resonant circuit. An input circuit of such a type gives an increase in 
signal amplitude, often of the order of 1 : 5. 

You know that the gain of an amplifier is increased as its anode load is increased. 
Here, the anode load of V! is the input impedance of V 2 , which is always low and 
usually less than 200 ohms. The gain of V x is therefore low— but because of this low 
gain, the valve remains stable. A higher gain would give a correspondingly high value 
to the "Miller" coupling between its anode and its grid, and the valve would cease to 
amplify and would burst into oscillation instead. 

The function of N y in the cascode circuit is therefore not to provide gain in the nor- 
mal way of an amplifier, but to act as an impedance multiplier. This high impedance 
makes it possible to connect the cascode direct to T t (which itself, as you have seen, 
provides a 1 :5 step-up in signal strength at Vj. input). 

A further advantage of the high impedance offered by Vj is that it improves the 
selectivity of the cascode by lessening the damping effect on the input signal of the low 
impedance of V 2 . 

V 2 in the cascode acts as a normal grounded-grid amplifier, amplifying the signals 
fed to its cathode from Vj. Its anode load, which is always made high to give it good 
gain, is a parallel-tuned circuit designed to resonate at the same frequency as the 
tuned circuit (Tj secondary-CO connected to the grid of V x . This arrangement en- 
sures that the impedances of the two tuned circuits, and therefore the overall gain of 
the cascode, are maximum only at the signal frequency to which the two circuits are 
made to resonate. 



§11] 



2.63 



The VHF Tuner — A Practical Cascode Amplifier 

A practical cascode amplifier contains a number of additional components com- 
pared with the basic circuit you have just learnt about. Pick them out on the circuit 
diagram below as you read the account of their functions which follows. 



A PRACTICAL 

CASCODE AMPLIFIER 

CIRCUIT 




H.T. (+) 



Output 
Signal 



AGC Voltage 



Q L 5 is a coil connected in the anode circuit of V 1; and acts as a peaking coil. It is 
put in to compensate for the progressive loss of gain which an amplifier suffers as 
frequency rises beyond a certain point. In conjunction with the input capacitance of 
V 2 and the stray capacitances which are always present at this point in the circuit, L 5 
forms an effective parallel-tuned circuit. 

The frequency to which this tuned circuit is resonant is slightly higher than the 
highest frequency of the signal to which the two transformers (L!-L 2 and L 3 -L 4 ) are 
tuned. This arrangement serves to improve the frequency response of the circuit as a 
whole. 

Q C 2 , in conjunction with C lt acts as a neutralizing capacitor, and is inserted in 
order to nullify the dangerous C as of Vj. Connected between Vj. anode and the 
lower end of the coil L 2 in the tuned circuit of V x grid, it feeds back to the grid a signal 
equal in magnitude, but opposite in phase, to the signal fed-back through the C aK . 
The two signals thus cancel one another out, and the stability of V x is greatly increased. 



2.64 [§l I 

The VHF Tuner — A Practical Cascode Amplifier {continued) 

^f R 2 and C 3 are the normal components used to deliver bias voltage to the cathode 
of Vi. Bias for V 2 is derived from the potential divider circuit R 3 -R 4 , which is con- 
nected across the HT supply. The values of R 3 and R 4 are so chosen that the positive 
voltage developed across R 4 will always be less positive than the voltage at V x anode. 
This latter voltage is always positive, and is always the same as the voltage at V 2 
cathode. Thus the less positive voltage applied from the biasing circuit to the grid of 
V 2 keeps the grid biased effectively negative with respect to its cathode. 

C 4 is the grounding connection for earthing the a.c. signal on the grid of V 2 . 
Q The negative AGC voltage is, as you know, fed back to the VHF tuner from a 
later stage in order to keep constant the overall gain of the receiver whatever changes 
occur in the input signal. This voltage is applied to the grid of V x through the resistor 
R x . The capacitor Q serves as a blocking capacitor to isolate the AGC voltage from 
earth, as well as forming part of the tuned circuit of V t grid and of the neutralising 
circuit with C 2 described above. 

Q The output of the cascode is taken from the transformer tuned circuit (L 3 -L 4 ) 
in the anode of V 2 . This circuit is tuned to resonate at the same frequency as the 
transformer tuned circuit (Li-L a ) in the grid of V x . L 2 , L 3 and L 4 have adjustable 
metal cores so that the two tuned circuits can be correctly set up during initial align- 
ment by the manufacturer. 

Brass or iron dust is used for the cores of inductors handling frequencies in Band 1 . 
Brass is usual for the handling of frequencies in Band 3. 



Summing Up the Cascode 



The purpose of the cascode amplifier is, first and foremost, to remain stable, while 
raising the amplitude of a high-frequency signal to a level which makes it usable in the 
mixer stage which immediately follows it, without equally raising the level of its accom- 
panying noise. High-frequency signals are not easy to amplify at all, and the gain of a 
cascode is typically only about ten times. Real amplification of the signal is only 
attempted after it has been reduced to a much lower intermediate frequency. 

Note that the cascode has been shown in the circuit diagram in this Section as two 
separate valves. In practice, these two valves are always contained within a single glass 
envelope. 

Frame Grid Valve Construction 

It was primarily the limitations of the conventionally-built cascode, and of other 
types of valve also tried out in the tuner stage of TV receivers, which led to the intro- 
duction of a type of valve construction somewhat different from those you have met so 
far. The new type has proved so valuable that nowadays almost every valve intended 
for use in HF applications, including pentodes and the latest types of UHF triode, is 
built in this way. 

Valves built on the frame grid principle, as it is called, look from the outside much 
like any other valve, and they function in exactly the same way. The construction of 
one of the electrodes in them is, however, different; and you must now see how it is 
done. 



§11] 



2.65 



Frame Grid Valve Construction {continued) 

The problem which needs to be overcome is this. The overall amplifying ability of 
any valve depends, as you know (Basic Electronics, page 2.21), largely on its mutual 
conductance (G m ). The higher the G m , the greater the amplification of the valve. 

It is possible to get high values of G m in a valve by positioning the control grid very 
close to the cathode, so that it has maximum influence on the electron stream. Unfor- 
tunately, the closer the grid and the cathode are brought together, the greater the 
capacitance (C gk ) between them— and a high C gk means a low input impedance to the 
incoming signal. The result is poor amplification — and back you are where you 
started. 

In the frame-grid type of valve, the advantages of close grid-to-cathode spacing with- 
out the disadvantages of increased C gk are obtained by constructing the grid of 
extremely thin wire — no more than 10 microns in diameter. (A micron is one 
thousandth of a millimetre, or about l/25th of one thousandth of an inch.) The grid 
is then positioned only 50 microns away from the cathode. 

The fact that the grid is not a continuous plate but rather an array of very thin wires 
separated from one another by an air gap greatly diminishes the capacitance between 
grid and cathode, despite the fact that the latter is positioned only some 0-05 mm. 
away. 

With such close spacing, however, it is vital to keep the grid wire structure rigid. 
Otherwise, small variations in the spacing (caused, e.g., by temperature changes or 
mechanical shock) could bring about large changes in anode current. 

Rigidity is achieved by winding the grid wire under tension in a spiral round a thick 
and sturdy frame. It is this frame which gives the type of valve its name. 

Rigid 
Frame 




Grid ii 

(very fine wire wound |& 
in tight spiral on former) :§ 



frame grid 

Valve 

CD traetton 



Grids constructed in this way can be substituted for the normal control grid in any 
type of valve — triode, tetrode or pentrode — and greatly increase their amplifying 
efficiency. For example, a TV receiver having frame-grid valves in the cascode and 
frequency-changing stages of the tuner, and serving as the first valve in the i.f. ampli- 
fier, will have an overall gain factor eight times that of a receiver using conventionally 
constructed valves. A typical value for the G m of a frame-grid triode used in a VHF 
tuner is 12-5 mA/V, compared with only 6 mA/V for conventionally constructed triode. 

The increased gain factors made possible by frame-grid construction have done 
much to improve the performance of TV receivers in areas of poor reception. 



2.66 [§|| 

The VHF Tuner— Channel Selection 

The next problem arises when the viewer wishes to connect his set to an aerial in a 
part of the country where the programme he wants to watch is radiated on a frequency 
different from that on which he has watched it hitherto. How can he switch his set to 
receive signals in a different frequency channel without having to alter important 
physical components inside his receiver ? 

The difficulty arises from the fact that, every time the viewer switches to a different 
channel, the resonant frequencies of three important components in the tuner need to 
be changed in order to conform to the different resonant frequency of the new aerial. 
(Remember that an aerial of different length is required for good signal reception in 
every different channel, because every channel carries a different range of wavelengths 
and because aerial length must always be close to one-half of the wavelength of the 
signal which the aerial is to receive. The three components needing to be changed are 
the tuned circuits in the grid and in the anode of the cascode amplifier, and the local 
oscillator in the mixer stage (which you will be learning about shortly). 

There are a number of ways of bringing about the required changes of resonant 
frequencies. Most British makers prefer a technique whereby a different set of induc- 
tor coils, each pre-aligned for one of the 13 different resonant frequencies required, is 
physically connected into the circuitry of the tuner every time the viewer switches to a 
different channel. The technique by which this is done is known as turret tuning. 

In turret tuning the complete set of coils required for each separate channel is 
mounted on the inside of a piece of insulated material shaped as a one-thirteenth seg- 
ment of a hollow drum. Each of these segments is individually known as a biscuit, 
and can be plugged into its place on the periphery of the drum. 



BISCUITS 

in place on 
Tuner DRUM 



Stud-type 
Contacts 



Channel 

Selection 

Knob 




Tuning Coils and 
Coil Former 



UNDERSIDE 
OF BISCUIT 



Connections to 
Stud Contacts 



The connecting leads to every set of coils are taken through the insulated material of 
the biscuit on which they are mounted to a pair of contact studs on its outer surface. 
Rotation of the drum is effected when the viewer operates the Channel Selection knob 
on the face of his TV receiver. Every click of this knob moves a different biscuit into 
position inside the tuner, and thirteen clicks make up one complete rotation of the 
knob. The turret tuner thus works on the same principle as does the bullet-holding 
chamber of a revolver. 

When a particular biscuit is moved into the working position in the tuner, the con- 
tact studs on its outside surface are connected into the circuitry of the tuner through a 
set of wiper contacts of the spring-leaf type fixed on the tuner chassis, which "make" 
with the contact studs of the biscuit. There being only one set of wiper contacts, only 
one set of coils can be connected into the tuner circuits at any one time. 



§11] 2.67 

VHF Tuners — Channel Selection (continued) 

Another method of changing the resonant frequencies in a VHF tuner is called 
incremental tuning. Though less used than the turret type, it is exclusively employed 
by some British manufacturers. 

The incremental tuner is a variable inductor in which a rotary switch is moved to 
short out unwanted parts of the total inductance. A number of coils, wired together 
in series, are mounted round the periphery of flat disks called wafers or banks. There 
are a number of these banks fitted on top of one another like a layer-cake, one for 
every section of the tuner (aerial, r.f. amplifier, oscillator, etc., circuits) which needs 
adjustment during the tuning process. All the banks are mechanically linked to the 
operating shaft of the switch so that when the Channel Selection knob is rotated by the 
viewer, all move round together. 



J 8 




One Bank of a 

VHF INCREMEKTAL TUNING SWITCH 



cu ta ! in9 v „ ^^-U^ Disk 

Shorting-hnk ^~- 

Switch Shaft 



Insulated when Set is switched to Channel 10 

— Coils 1-9 are shorted -out 

— Coils 10-13 are in-circuit 



The inductance value of every coil mounted round every bank of the switch is chosen 
with care. When the tuner is required to operate on the lowest channel frequency 
(Channel 1, 45 MHz vision), the value of inductance required in the various stages 
needing adjustment is high. All the coils in every bank are used, and their individual 
values of inductance are added together and connected in series to a larger inductor, 
of fixed value, which is not affected by the switch and which provides the greater part 
of the total inductance required. There is a separate fixed inductor, of course, for 
every bank of the switch. 

When a lower value of total inductance is required (say when the tuner is being set to 
operate on the highest channel frequency — Channel 13, about 215 MHz vision), all the 
coils save one on every bank are shorted-out, so that very little is added to the value of 
the fixed inductor for that setting. 

At the low-frequency side of every wafer, the inductors are small coils of wire thick 
enough to need no other support. Towards the high-frequency side, straight (or only 
slightly bent) pieces of wire connected between the switch tags are sufficient to provide 
the small values of extra inductance required. 

Although the incremental-inductance type of tuner is simpler in construction than 
the turret, and so generally cheaper, it has a number of disadvantages. One is the 
large number of switch contacts required, especially in the higher frequency channels 
where many unwanted inductors need to be shorted out. Dirt on the surfaces of any 
of these contacts can cause trouble. 

Another disadvantage is the difficulty of alignment. Individual coils cannot be 
adjusted for inductance value without affecting the rest of the inductors, since all are 
connected in series. This means that the coils have to be carefully manufactured so as 
to have precisely the required values of inductance. Once they have been fitted, 
further adjustment to them is a matter for expert attention. 



2.68 



[§H 



The VHF Tuner — Lowering the Frequency of the Wanted Signals 

Once the channel has been selected in this way, the sound and vision signals in it are 
amplified to a usable level by the cascode valve in the r.f. amplifier, and then fed to the 
frequency changer for conversion to a lower frequency. 

The frequency changer consists of two valves — a local oscillator which is a triode, 
and a mixer valve which is a pentode. In the majority of British VHF tuners, the two 
are contained within a single glass envelope. 

Mixing is done according to the normal superheterodyne principle you learnt about 
in Part 5 of Basic Electronics. The output is two still-separate sound and vision sig- 
nals each carrying the amplitude modulation bearing the desired information, but both 
on intermediate frequencies low enough to make possible all the amplification needed 
in later stages of the receiver. 

The circuit diagram of a typical VHF frequency changer stage looks like this: 

The FRtMiNCy CHANGCR Circuit |jjj 

in the ¥M Tuner I 



Output to 
I.F. Amplifiers 



H.T. (+) 




The local oscillator in the circuit above is the triode V 4 . This functions as a Col- 
pitts oscillator of the type you learnt about on Basic Electronics, page 3.63, and makes 
use of its own interelectrode capacitances to resonate with the variable inductor L 8 . 
HT is supplied to V 4 through R 8 , being prevented from reaching the grid of the valve 
by the capacitor C 9 . The valve is self-biased by the components R 9 and C 9 — the 
latter thus performing a second useful function. 

Since it is necessary for the mixer stage to produce the same intermediate-frequency 
signals whatever the frequency channel over which the viewer wishes the tuner to 
respond, the operating frequency of the local oscillator needs to be altered every time 
the channel is changed. (Remember that the intermediate frequency required is the 
resultant when the sound or vision signal frequency is deducted from the frequency of 
the local oscillator. The L.O. frequency must therefore be kept above signal fre- 
quency by a constant amount.) 



§11] 2.69 

The VHF Tuner— Lowering the Frequency of the Wanted Signals (continued) 

The frequency of the local oscillator is changed at the same time as the tuning coils 
are changed, by the viewer manipulating the Channel Selection knob on the outside of 
his receiver. If incremental tuning is being used, the change of frequency is achieved 
by varying the value of the inductance L 8 . With the turret tuner, it is achieved by 
physically changing this inductor for one of a different value. 

This gives coarse control of L.O. frequency. Fine control is achieved when the 
viewer rotates another knob on the face of his receiver, generally mounted with its 
shaft concentric with that operated by the Channel Selection control so that the two 
appear to operate through the same shaft. 

This Fine Tune control is achieved by adjusting the value of the variable capacitor 
Cio, which is connected in parallel with V 4 . This capacitor, which is of very small 
capacitance, is effectively in parallel with the C ak of V 4 and thus forms part of the 
tuned circuit of the oscillator. 

Fine tune control can, of course, be used by the viewer to correct for frequency drift 
within a given channel, as well as for final tuning after the channel is changed. 

The signal produced by the local oscillator is generally injected on to the control grid 
of the pentode mixer valve V 3 through the low-value capacitor C 6 (though you will 
come across other arrangements which rely either on the self-capacitances between the 
two valves, or on the mutual inductance between the inductors L 8 and L 4 when placed 
physically close to one another in the tuner). 

The channel signal reaches the control grid of V 3 through C 5 , and superheterodyne 
action follows. R 7 -C 8 form the normal cathode-biasing arrangement for V 3 , and R 5 
is its grid leak. The anode load of V 3 is the transformer stage C 7 -R 6 -L 6 -L 7 . 

Among the many intermediate frequencies produced by the mixing process in V 3 are 
the two frequencies which the viewer wants — namely, the i.f.'s of the sound and vision 
signals he is looking for. The resonant circuit L 6 -L 7 is broadly tuned to these i.f.'s, 
and ensures that the degree of amplification which the mixer valve additionally pro- 
duces is applied only to the two wanted frequencies. This broad tuning is achieved 
with the aid of the damping resistor R 6 , connected in parallel with the coil, which 
broadens the frequency response of the tuned circuit by introducing a resistive loss. 

The two intermediate frequencies required are fed to the main i.f. amplifier stages in 
the next section of the receiver through coil L 7 , which is loosely coupled to L 6 in such a 
way as not to alter either the resonant frequency or the loading of the latter. 

The intermediate frequencies at present used for 405-line reception in Bands 1 and 3 
are 34-65 MHz for the vision signal and 38- 15 MHz for the sound signal. With the 
local oscillator frequency kept above both these frequencies so that the two i.f.'s are 
obtained by deduction from it, the 45 MHz frequency of the vision signal in Channel 1 
of Band 1 calls for an L.O. frequency of (45 + 34-65 = ) 79-65 MHz. This oscillator 
frequency produces an i.f. of 38-15 MHz from the sound carrier by deduction from it 
of the sound carrier frequency in Channel 1 of 41-5 MHz. 

Note that the two i.f.'s (34-65 MHz for vision and 38-15 MHz for sound) are still 
3-5 MHz apart, but that their relative magnitudes have been inverted. The sound 
frequency is now the higher of the two, whereas before mixing the vision carrier fre- 
quency (45 MHz) was 3-5 MHz above the sound carrier frequency (41-5 MHz). This 
inversion always occurs when the local oscillator is operated above signal frequency. 



2.70 




The 


Complete VHF Tuner 




o 




0) U) 
-Q 0) 
U ~ 

_ Q. 

.5 E 
ou: 



[§ll 




Q. 

E 
< 



4C±> 



-vQ&b- 



o 



-^WA- 



AAAAr 



-S-g 



I 



it 






u 
5 



o 
> 

o 
o 

< 



H 



§11] 



2.71 



The UHF Tuner 

The function of the UHF tuner is exactly the same as that of its VHF counterpart: 
namely, to select the required signal from the many other signals contained in the 
channel covered by the aerial, and to convert the sound and vision frequencies of this 
signal into the two intermediate frequencies needed by later stages in the receiver. 
Because of the extremely high frequencies over which the UHF tuner has to operate, 
however, the VHF techniques which you learnt about earlier in this Section will not 
work; and something quite different has to be adopted instead. 

The main problem which arises in UHF tuning is that at ultra-high frequencies the 
ordinary LC tuned circuit will not work. For reasons which you will have to accept 
on faith until you have done a good deal of work on microwave theory, at UHF most 
inductors and most capacitors develop a distressing tendency to behave like their 
opposite numbers (capacitors behaving as inductors, and vice versa). Such com- 
ponents would work properly at these frequencies only if they were so minutely small 
as to be practically incapable of manufacture. 

Before you consider how to set about overcoming these difficulties, it is worth taking 
a brief look at the nature of the practical problem which has to be solved. 

Bands 4 and 5, over which Britain's 625-line programmes are transmitted at UHF, 
are divided into 48 channels numbered from 21 to 68, each having a channel width of 
8 MHz. Band 4 contains 14 channels (Nos. 21 to 34), and extends over a frequency 
range from 470 to 582 MHz. Band 5 contains 30 channels (Nos. 39 to 68), extending 
over a frequency range from 614 to 854 MHz. Channels 35 to 38 are reserved for 
non-TV use. 

Every transmitting station is allotted four channels, not consecutive but generally 
spaced three or four channels apart. No station at present uses all of its four chan- 
nels, but all will eventually do so as additional 625-line programmes are introduced. 

If a TV receiver is to remain in a single area of the country all its working life, its 
UHF tuner will therefore never be required to operate over more than four channels— 
and with very few stations will the maximum frequency range it has to cover exceed 
88 MHz. For this reason, a good many UHF tuners are of the push-button type, 
covering four channels only. This makes them easy to tune by the non-expert; but if 
their owner takes them to another part of the country served by different channel 
frequencies, they will not work in the new area until all their channel coils have been 
replaced. 

Another type of UHF tuner is continuously variable, and is capable of tuning over 
all the available UHF channels. The required channel is "tuned in" by operation of 
a slow-motion control, which is carefully rotated until the number of the channel 
sought appears on a numbered dial. 



A Receiver with 
PUSH-BUTTON TYPE 
Tuning 




2.72 [§|| 

The UHF Tuner — Resonant Lines 

You have seen that the ordinary LC tuned circuit will not work properly at UHF 
because at these frequencies most inductors and most capacitors cease to behave as 
they do at lower frequencies. What can be found to take their place? 

Recall at this juncture what you learnt about resonance in mis-matched transmission 
lines in Section 5 of Basic Electronics, Part 4. A resonant line, you found, is a section 
of a transmission line consisting of two parallel wires, which can be made to behave 
either as a series resonant circuit (having low resistance) or as a parallel resonant cir- 
cuit (having high R) depending on its length and on the nature of its termination. 
Given a section of line whose electrical length is exactly one-half of the wavelength of 
the transmission being passed through it, the section will behave as a series-resonant 
circuit when its termination is a short circuit, and as a parallel-resonant circuit when its 
termination is an open circuit. 



fc= 



Short-circuit 
Termination 



A SHORT-CIRCUITED Termination behaves like 
A SERIES-RESONANT LC Circuit f$| £g|^ 




Open-circuit 
Termination ■ 





(large)/ 


— 





An OPEN-CIRCUITED Termination behaves like 
A PARALLEL-RESONANT LC Circuit 

Now the grid and anode circuits of the cascode amplifier are both parallel-resonant, 
so both could seemingly be replaced by half-wavelength resonant circuits having open- 
circuit terminations. But that solution itself raises two difficulties. First is the prob- 
lem of how to alter the physical length of the straight lines of wire forming the resonant 
circuits every time the wavelength of the required signal varies. At the bottom end of 
Band 4, at 470 MHz, the wavelength is about 635 mm. At the top end of Band 5, at 
854 MHz, it is a little less than (356 mm). 

The second difficulty is that it would be exceedingly inconvenient to have to fit into 
the TV receiver a number of straight and parallel lengths of wire (if that be the solution 
tried for the first difficulty), of which the longest would have to be half the length of the 
longest wavelength in the two Bands. You have just seen that this is some 635 mm; 
so the corresponding resonant line would have to be about 318 mm long. This 
would make the physical size of the screening box in which the tuner section has to be 
encased unacceptably large; so some other solution must be sought. 

Fortunately the resonant line has another property which can be used to overcome 
the problem. 



§11] 



2.73 



The UHF Tuner — Resonant Lines (continued) 

You learnt on page 4.54 of Basic Electronics that between the open-circuited end of a 
transmission line and a point one-quarter of a wavelength back from the open circuit, 
the impedance of the line is capacitive. In other words, any length of open-ended line 
whose length is less than one-quarter of the wavelength being transmitted behaves as 
if it were a capacitor. 

So why not simply substitute a capacitor for an appropriate length of the line, and 
see if it won't do just as well ? 

The longest wavelength to be handled in Bands 4 and 5 is about 635 mm. The 
corresponding half- wave resonant line is about 318 mm, and is (as you know) incon- 
veniently long. But a quarter- wavelength is all of 150 mm — and if you snip off a bit 
rather less than 150 mm long from each end of the half-wave resonant line and connect 
a capacitor of the correct value in the place of each bit snipped off, you have a line be- 
having just as it did before — but now only some 30 to 40 millimetres in length. 

Moreover, capacitors have the great advantage over bits of parallel transmission 
lines that they can easily be made variable. So if you fit variable capacitors in place of 
the snipped-off bits of line, you can vary the effective (electrical) length of the now- 
very-short resonant line so that it can be tuned to operate over a wide band of fre- 
quencies. 



u. 

1 




2 




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1 ■" 










1 1 
1 1 
1 , 

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1 
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How the 
Physical length 
ofaHatfWwe 
Resonant line 
is Shortened 



The usual method of tuning a UHF tuner thus employs a four-gang variable capa- 
citor operated by the viewer from the front panel of the receiver through a chain of 
high-reduction gearing linked to the Channel Selection dial. Each gang of the capaci- 
tor is connected across the termination of a resonant line in one of the five screened 
compartments into which (as you will see in a moment) the UHF tuner has to be 
divided up. The value of capacitance of all the gangs can be made-such that the lines 
which they terminate are resonant at any frequency between 470 MHz (the lowest 
frequency in Band 4) when the gang is set to maximum capacitance in Channel 21, and 
854 MHz (the highest frequency in Band 5) when the gang is set to minimum capa- 
citance in Channel 68. 

It is normal to find small-value trimming capacitors also connected at either end of 
the lines for use by the manufacturer in his initial alignment of the tuner. 



2.74 [§ll 

The Need for Screening in the UHF Tuner 

You have already seen that serious problems of component behaviour arise at ultra- 
high frequencies, such as you have not met in any of your earlier work on electronics. 
This is especially true when you are handling frequencies in Band 5, which can be as 
much as four times higher than the highest VHF frequencies encountered in Band 3. 

At UHF even a straight piece of wire develops a significant inductance of its own, 
which makes it impossible to wind coils in the usual way. The values of capacitance 
to which a UHF circuit responds are so small that the mere proximity of another UHF 
circuit is often sufficient to produce a capacitive coupling between the two. This 
coupling can be deliberately put to good use in the design of a UHF circuit, but it can 
also be a nuisance which has to be guarded against. 

Yet again, signals of very high frequency can also cause feedback of oscillator radia- 
tion from the tuner to the aerial, from which they could be re-broadcast to the detri- 
ment of other viewers. This back-radiation from the UHF tuner is the subject of 
stringent international control, and it is an important secondary function of the r.f. 
amplifier to isolate the frequency-changer stage in the UHF receiver from the aerial. 

For all these reasons, one of the paramount requirements in the UHF receiver is that 
screening arrangements must be effective— and the stage in which they are most needed 
is the tuner, where the frequencies handled have yet to be dropped to the i.f. level. 

As you know, the tuner stage itself is commonly screened from the other circuits in 
the receiver by being enclosed in a metal box of its own, placed well away from the 
other circuits. But many of the circuits inside the tuner stage itself also need screening 
from one another. This is generally done by dividing up the metal box of the UHF 
tuner into a number of separate compartments, each having special feed-through 
arrangements into its neighbours, and each containing only those components or 
stages which will not have undesirable reactions on one another when they are at work. 
A typical arrangement is the Mullard Ltd. design illustrated opposite. The tuner 
circuitry is there housed in five small compartments fully screened from one another by 
metal partitions. Holes are drilled in these partitions to take the necessary connecting 
wires, and some of these holes are fitted with what is known as a feedthrough capacitor. 
One electrode of a feedthrough capacitor is a cylindrical piece of metal soldered or 
bolted to the metal wall of the compartment through which it is desired to make con- 
nection. The other is the connecting wire itself as it passes through the first electrode. 
The purpose of a feedthrough capacitor is to pass high-frequency a.c. signals to 
earth, but to allow d.c. and low-frequency a.c. signals to pass along the wire. It acts, 
in short, as a high-frequency decoupler. 

The resonant lines themselves are all plated in silver. UHF waves find so much 
opposition to their passage through even good conductors that they tend to travel 
along the circumference of a wire rather than through its middle. (This phenomenon 
is known as skin effect.) So in order to keep the resistance of the current path as low 
as possible, the outside of the resonant lines are coated with a highly conductive metal. 
In the UHF tuner illustrated, the first two compartments contain respectively the 
cathode and the anode tuned circuits of the r.f. amplifier; the third and fourth com- 
partments contain the cathode and anode tuned circuits of the mixer; while the fifth 
compartment houses the tuned circuit of the i.f. output. The shaft of the four-gang 
variable capacitor runs through the first three internal compartment walls, and 
operates the variable capacitors in the first four compartments. 



§11] 



2.75 



A Typical UHF Tuner 







-MM 



E 

l< § 
2^.75 
ID cc > 



CO 

O 



^N* 






4 



2.76 



[§H 



Radio-Frequency Amplification in the UHF Tuner 

There are several special problems connected with the r.f. amplification of UHF 
signals received in the tuner. 

In the first place, UHF signals tend to be much smaller in amplitude when they 
reach the aerial than do their VHF counterparts. The reason is that their higher fre- 
quency causes them to suffer much greater attenuation as they pass through the atmos- 
phere — especially when they have to pass through fog, cloud or rain. 

In the second place, although (as you know) a UHF aerial system is designed to give 
much higher gain than is its VHF counterpart, its necessarily smaller dimensions give 
it a lower signal-collection efficiency. 

The result is that the UHF signal is generally much smaller than is a VHF signal 
when it leaves the aerial on its way into the receiver. To make matters worse, it is then 
apt to suffer greater attenuation during its passage through the aerial feeder cable, 
however good this cable may be. 

What reaches the r.f. amplifier is thus a signal of very high frequency but very low 
amplitude. This is an awkward combination, because the high frequency makes am- 
plification difficult, while the low amplitude makes it all the more necessary. 

A further difficulty stems from the small size of the UHF signal, in that the signal 
itself is not much greater than the noise level which is inherent in any type of triode 
with which you are familiar. This means that little improvement in the signal-to-noise 
ratio can be achieved with normal triode valves, even of the frame-grid type. 

The cascode amplifier, which works so well with the comparatively large VHF sig- 
nal, cannot therefore be used at UHF. A pentode cannot be used either, because it 
cannot handle UHF frequencies; so the only solution is to use a new kind of triode 
specially designed to meet the triple requirements of low inter-electrode capacitance, 
low inherent noise, and an acceptable factor of gain. 

The UHF Triode in the RF Amplifier 

To meet these requirements, a new type of UHF triode has been developed, in which 
the values of interelectrode capacitance are kept small by reducing to a minimum those 
areas of the anode and grid, and of the anode and cathode, which actually face one 
another. Such a triode is illustrated below. 



Glass 
Envelope 



Wiring of 
Frame Grid 




Anode 






Frame-Grid 
Supports 



Cathode 



PLAN VIEW 



Wi 



Heater 




Grid 



Anode 



Grid 

Line of Screening 
Panel in Tuner Circuit 



UNDERSIDE VIEW 



§1 1] 2.77 

The UHF Triode in the RF Amplifier (continued) 

You will note, in the illustration of the UHF triode opposite, that the grid of the 
valve is of frame-grid construction. Interelectrode capacitances are then further 
reduced by employing a wedge-shaped anode positioned on one side only of the grid, 
instead of completely surrounding it. Only a small area of the anode is in this way 
placed in close proximity to the grid and to the cathode, and the capacitances between 
them are so kept much smaller. 

Since only one side of the cathode is "seen" by the anode, only this side needs to be 
coated with emissive material. The electron-collection efficiency of this arrangement 
is naturally not as good as that of the double-sided cathode inside a cylindrical anode; 
but the whole conception of the valve is based on the need for compromise. Thus, a 
satisfactory anode current would call for a large emissive area on the cathode, and the 
high mutual conductance needed between grid and cathode would demand close 
spacing between them. Yet such an arrangement would give rise to a C gk which 
would be unacceptably large. 

The solution must be to decide on the maximum tolerable C gk , and to make do with 
the anode current and value of grid-cathode conductance which an emissive surface so 
regulated in size and position will provide. 

The grid of a UHF triode is always of frame-grid construction in order to obtain a 
high G m without an excessive C ek . 

One other construction detail of the UHF triode is of interest. When a triode is 
used for amplifying UHF signals, another potential source of instability is the induc- 
tance values of the connecting leads within the valve itself (i.e., the leads connecting the 
electrodes to the base pins). These inductances can be quite high; and unless their 
effect is reduced, they can cause instability by resonating with the interelectrode capa- 
citances to form tuned circuits. 

In the UHF triode, therefore, the control grid is connected at five separate points to 
five separate pins on the valve base. The UHF triode is always connected into the r.f. 
amplifier circuit in the grounded-grid configuration, in order to prevent the "Miller 
build-up" of C ag which would make it unstable. When the valve is so connected, the 
five pins aforesaid are connected both together and to the chassis of the set (which is 
effectively earth). Since the five connections are now in parallel, their effective 
inductance is only a fifth of that of a single lead. 

The inductance values of individual leads in the UHF triode are further reduced by 
constructing them from thick wires plated with silver. 

The degree of gain produced by a UHF triode of frame-grid construction connected 
in the grounded-grid configuration is a good deal lower than that produced by a cas- 
code amplifier handling a VHF signal— typically about eight times lower. Most 
multi-band Dual-Standard receivers make up for this deficiency in gain by passing the 
i.f. output signal through the unused mixer stage of the VHF tuner, using the latter as a 
kind of pre-amplifier for the intermediate-frequency UHF signal before it reaches the 
main i.f. amplifier section of the receiver. 

The extra amplification obtained in this way does much to ensure that the VHF and 
the UHF intermediate-frequency signals both reach the i.f. amplifier section at 
approximately equal strength. 



2.78 [§ll 

A Complete UHF Tuner 

The r.f. amplifier stage in the UHF tuner thus consists of a special UHF triode with 
its grid grounded by direct connection to the compartment wall. It is labelled Y x in 
the circuit diagram on page 2.75. Its cathode and anode tuned circuits are made up 
of specially-shortened, capacitively-tuned resonant lines, silver-plated in order to assist 
the passage of the UHF waves travelling along the outer surfaces of the lines. 

The signal from the aerial is taken to the coupling loop L 1( which is itself inductively 
coupled to a tuned circuit consisting of L 2 and the variable capacitors VC 2 , VC 3 and 
VC 4 connected across its ends. This tuned circuit of L 2 is in turn coupled to the 
cathode of V x through another coupling loop L 3 . This method of coupling is chosen 
in order to reduce the damping effect on the input signal of the low input impedance 
of the valve. 

The circuit of L 2 is initially tuned by the makers with the aid of the trimming capaci- 
tors VC 2 and VC 3 and of one section (VC 4 ) of the four-gang channel-tuning capacitor. 

The amplified signal at the anode of V x is coupled through Q to the tuned circuit 
formed by L 4 , VC 5 , VC e and VC 7 . This tuned circuit forms the primary of a band- 
pass transformer, the secondary of which (L 5 ) is an identical tuned circuit in the next 
compartment. Capacitive coupling between the two tuned circuits of this transformer 
is effected through two small holes in the compartment screen; and the circuits are 
further connected at their centre points through the small inductor RFC 2 . 

The degree of coupling between the tuned circuits of L 4 and L 5 is arranged to pro- 
duce a double-humped "over-coupled" response curve. You will see at the end of this 
Section that a response curve of this shape is needed to pass both the sound and vision 
signals to the mixer stage. 

The half-dozen feedthrough capacitors which you will see in the illustration on page 
2.75 pass d.c. but take high-frequency a.c. to earth through the metal walls of the tuner 
and the chassis of the receiver. An example is the feedthrough capacitor which 
supplies HT to the anode of Vj. (The purpose of the choke RFC! in this circuit is to 
prevent undesirable coupling between stages in the tuner from taking place through the 
HT supply.) 

You cannot use in the UHF tuner the kind of frequency-changer circuit, consisting of 
a triode oscillator and a pentode mixer, which does duty in the VHF tuner. The 
reason is that the high degree of noise generated in such a circuit (mainly in the 
pentode) would swamp the signal. 

Used instead is a single triode valve connected to form what is called a self-oscillating 
mixer circuit. Such a circuit not only provides the required local-oscillator signal, but 
also mixes it with the wanted signals fed in from the r.f. amplifier. The output is, of 
course, two i.f. signals of much lower frequency, one carrying the amplitude and the 
other the frequency modulations of the wanted vision and sound signals, respectively. 

Like its VHF counterpart, the local -oscillator in the UHF tuner is made to oscillate 
at a frequency above that of the channel signal. This means that it must operate at the 
very high frequency of 900 MHz. At this level, frequency stability is not easily main- 
tained. Some UHF tuners, notably those offering the push-button type of channel 
selection, solve the problem by incorporating an automatic frequency-control circuit 
which corrects the oscillator for frequency drift by deriving a control voltage from a 
frequency discriminator circuit driven from the vision i.f. amplifier, and by using this 



§11] 



2.79 



A Complete UHF Tuner (continued) 

voltage to vary the potential across a semi-conductor diode of variable capacitance 
connected across the tuned circuit of the oscillator. 

The self-oscillating mixer circuit mentioned on the last page (and designated V 2 in 
the large diagram on page 2.75) is most easily understood if it be reduced to its 
"equivalent" form. This is done in the illustration below. 



Ca-k 



Cg-k Ca-g 



L7 



Equivalent Circuit 

of the 
UHF OSCILLATOR 



The oscillator valve V 2 is connected as a grounded-grid Colpitts type (Basic Elec- 
tronics, page 3.63), and relies on its own inter-electrode capacitances to provide the 
feedback necessary for oscillation. Thus, in the diagram above, the feedback between 
the anode and cathode circuits is provided by the anode-to-cathode and the grid-to- 
cathode capacitances — C ak and C gk respectively — while the anode-to-grid capacitance 
— C ag — forms part of the anode tuned circuit. 

L 7 is a capacitively-tuned resonant line whose resonant frequency determines the 
operating frequency of the oscillator. The tuning of L 7 is carried out by variation of 
the VC 13 section of the four-gang channel-tuning capacitor (and also, of course, by the 
two trimming capacitors VCn and VC 12 , preset by the manufacturers). 

Mixing of the channel and local-oscillator signals takes place in the third, or middle, 
screened compartment of the tuner. This compartment contains essentially the 
cathode circuit of V 2 and the second part (L 6 ) of the tuned-transformer circuit which 
you saw was capacitively coupled to L 4 in the anode circuit of the r.f. amplifier. The 
signal is injected into the oscillator circuit by mutual inductance between L 5 and the 
small coupling loop L 6 in the cathode circuit of V 2 . 

In the output circuit of the UHF tuner, the i.f. signal produced in the mixer stage 
from the beating together of the channel signal and the signal from the lo is taken 
through a choke RFC 3 (whose purpose is to offer a high impedance to the oscillator 
signal and so keep it out of the HT supply and intermediate-frequency circuits) to a 
tuned coil, L 8 , in the anode circuit of V a . This tuned coil is situated in the fifth and 
last of the five screened compartments into which the UHF tuner is divided. 

It forms one-half of a tuned transformer circuit of which you will be meeting the 
second half in the IF Amplifier. The two halves of this transformer circuit, which are 
physically some distance apart, are coupled by a method known as bottom capacitance 
coupling, in which the capacitance of the coupling cable itself is made to provide a 
significant fraction of the total coupling capacitance. 



2.80 



[§H 



A Complete UHF Tuner {continued) 

The i.f. signal is taken to the output terminal of the tuner through another choke, 
RFC 4 . In combination with the two small capacitors C 3 and C 4 , this choke forms a 
filter which passes the low-frequency i.f. signal but filters off to earth any of the much- 
higher-frequency signals from the local oscillator which may find their way into the 
output circuit. 

Frequency Response in the TV Tuner 

Ideally, the overall frequency response of any TV tuner, VHF or UHF, would be 
completely flat over the full range of frequencies occupied by the sound and vision 
carriers of any given channel signal, and zero at all frequencies outside those limits. 
In practice, an adequate approximation to this ideal is attained by so arranging the 
coupling of the various tuned circuits in the tuner that a response of the shape illus- 
trated below is obtained. 



Sound 
Signal 




695-25 701-25 

Carrier Frequency (MHz) 

You will see that the pattern has a broad top with two humps on it — one correspond- 
ing to the frequency (or the i.f.) of the vision carrier, and the other corresponding to 
the frequency (or i.f.) of the sound carrier. 

The illustration shows the frequency response of a typical UHF tuner, but that of a 
VHF counterpart would not be very different. 

The overall frequency width of such a response curve as that shown is sufficient to 
embrace most of the frequencies contained in any wanted signal. 



§11] 

REVIEW of the TV Tuner 



2.81 



Tin BASK 



The tuner is the first stage in the TV receiver. Its function is to select the required 
channel signal from those fed to it by the aerial, to amplify this signal to a usable level, 
and to convert its sound and vision frequencies into the sound and vision intermediate 
frequencies of the receiver. 

Different types of tuner are needed for handling signals in the VHF Bands 1, 2 and 3 
(Channels 1-13), and for signals in the UHF Bands 4 and 5 (Channels 21-68). Dual- 
Standard receivers used in Britain to operate on both 405 and 625 lines are fitted with 
both types of tuner. Complex switching arrangements are involved when the viewer 
wishes to change over from one system to the other. 

The three main sections of a TV tuner cover the r.f. amplification of the wanted sound 
and vision signals, their conversion to a usable intermediate frequency, and selection of 
the desired signal channel. At VHF, all the operations required can be performed by 
making use of the properties of the LC tuned circuit, but at ultra-high frequencies this 
type of circuit will not work. The UHF tuner has therefore to make use instead of the 
properties of the quarter-wave resonant line. 

Radio-frequency amplification — In the 
VHF tuner, the r.f. amplifier consists of a 
double-triode valve connected in a cascode 
circuit. This type of circuit combines the 
stability of the grounded-grid configuration 
with the good gain of the grounded-cathode 
configuration, and minimizes their respective 
disadvantages. 



Frame-grid construction gives a valve the 
advantages of close spacing between grid and 
cathode while minimizing C gk . The grid is 
constructed of very fine wire wound in a tight 
spiral round a rigid frame. The small air 
gap between adjacent wires greatly reduces 
C Kk even when the cathode is positioned very 
close to them. 



R.F. amplification at UHF has to be 
achieved differently, for the much smaller 
UHF signal would be drowned by the in- E 
herent noise level of the cascode. A single 
UHF triode is used instead. The narrower 
end of a wedge-shaped anode is positioned , 
on one side only of a frame-type grid, so as 
to keep interelectrode capacitances as low as 
possible. 

The low gain of such a valve is often compensated by passing its output signal through 
the unused mixer stage of the VHF tuner. 






UNKRSK VEW 



2.82 



[§H 




Equivalent Circuit 

of the 
UHF OSCILLATOR 



REVIEW of the TV Tuner (continued ) 

Frequency changing in the VHF tuner is usually done by means of separate local- 
oscillator and mixer circuits combined in a single triode-pentode envelope. The triode 
functions as the local oscillator, the pentode as the mixer. The channel signal from the 
cascode amplifier and the signal produced by the local oscillator are both fed to the con- 
trol grid of the pentode, where mixing of the additive type takes place. 

The very small UHF signal would be 
drowned by the noise inherent in such a mul- 
tiple-valve arrangement, and its frequency 
has to be lowered by other means. A single 
triode circuit is used instead, and is made to 
perform the dual function of local oscillator 
and mixer by being connected as a self- 

"i i i i I i oscillating mixer. Such a circuit has poor 

gain; so the still-very-small UHF intermedi- 
ate-frequency signal is often passed for pre- 
amplification through the idle mixer stage of the VHF tuner. 

Channel selection is achieved in the VHF 
tuner when the viewer operates a control which 
connects physically-different inductor coils 
into the r.f. amplifier and frequency-changer 
circuits. On the turret tuner the 13 different 
sets of coils needed to tune into the 13 chan- 
nels in Bands 1, 2 and 3 are mounted on in- 
sulated biscuits to form a hollow drum. 
Wiper contacts connect the selected set of 
coils into the appropriate circuits in the tuner. 

An alternative is to use the incremental 
tuner, in which a rotary switch is moved to 
short out parts of a larger total inductance. 
A number of coils, wired together in series, 
are mounted round the periphery of flat disks 
called wafers or banks. They are then pro- 
gressively shorted out when the operating 
shaft of the switch (to which they are 
mechanically linked) is rotated by the viewer. 





One Bank of a 

VHF INCREMENTAL TUNING SWITCH 



When Set is switched to Channel 10 

— Coils 1-9 are shorted-oui 

— Coils 10-13 are in-circuit 



In the UHF tuner of the multi-channel type, channel selection is accomplished by 
means of a four-gang variable capacitor mounted on a shaft which runs through three of 
the compartment walls inside the metal box housing the tuner. When the viewer rotates 
this shaft by operating the channel-selection control knob, tuned circuits in four of the 
five compartments of the tuner are re-tuned to a different frequency. Continuous tuning 
over the full range of 44 channels used for TV in Bands 4 and 5 is thus made possible. 

UHF tuners of the push-button type are tuned to four channels only, in Bands 4 and 5. 
The receivers to which they are fitted would need adjustment before they could pick up 
signals from a station in another part of the country transmitting in different channels. 



§12: THE IF AMPLIFIER 



2.83 



When the sound and vision signals, still unseparated, leave the tuner, their ampli- 
tudes are very low— only of the order of a millivolt or so. This is not sufficient to 
operate the next main stages in the receiver, which require signal amplitudes ranging 
from 1 to 5 volts. 

It is the principal job of the intermediate frequency amplifier to apply to the signals 
amplification of that order of magnitude— say, an overall gain in the region of 12,000 
times. 

There is no particular problem involved in achieving this degree of gain. The sig- 
nals leaving the tuner have all, as you know, been reduced to a comparatively low 
intermediate frequency. (In the VHF system working on 405 lines, the i.f.'s are 34-65 
MHz for the vision signal and 38-15 MHz for the (AM) sound signal. In the UHF 
system working on 625 lines, they are 39-5 MHz for the vision signal and 33-5 MHz for 
the (FM) sound signal.) All these frequencies are low enough to be handled, without 
much danger of distortion or of excessive noise, by the kind of amplifying devices you 
learnt about in Parts 2 and 6 of Basic Electronics. 

Special problems arise, however, in the rejection of interference signals, and in the 
shaping of the frequency response curves to accommodate the single-sideband trans- 
mission of both the VHF and the UHF signals. There are also difficulties because of 
the different ways in which the VHF and the UHF sound signals are handled in the 
British Dual Standard receiver. To recapitulate : 

In the 405-line system, the sound and vision signals from the desired channel are 
separated as soon as they enter the i.f. amplifier stage, and are then passed to two 
different i.f. amplifier circuits. The sound signal then goes off to the sound detector 
and the vision signal to the video detector. (This is called the split-sound method of 
processing the sound signal.) 

In the 625-line system, the sound and vision signals are amplified together in a single 
i.f. amplifier circuit and are not separated until they reach the video detector. (This is 
called the intercarrier method of processing the two signals.) 

The i.f. amplifier stage of the Dual-Standard receiver thus requires two separate i.f. 
amplifier circuits for handling the VHF signal. For reasons of economy, the circuit 
which handles the VHF vision signal is also used to amplify the UHF signal, in which 
the sound and vision signals are still both present. 

Because of the presence of the UHF sound signal, this latter circuit cannot accu- 
rately be called the Vision IF Amplifier, though that is the principal job it has to do. 
So let us christen it IF Amplifier A— though you should remember that the name is an 
unofficial one. The functions of this IF Amplifier A are : 

The i.f. amplification of the VHF vision signal; 

The i.f. amplification of the UHF sound and vision signals. 

The second amplifier— let us call it (equally unofficially) IF Amplifier B— is a sound 
amplifier only. Its functions are : 

The i.f. amplification of the amplitude-modulated VHF sound signal; 

The i.f. amplification of the frequency-modulated UHF sound signal passed back to it 
from the later stage {the Video Detector) in which the UHF sound and vision signals are 
finally separated. 



[§12 



2.84 

The IF Amplifier Stage 

The rather complicated arrangement outlined on the last page will be easier to 
understand if you study the block diagram below. The dotted rectangle in it outlines 
the IF Amplifier block considered in this Section. 

Sound-and-Vision 
~" ~UIHF~Sig7iais 



UHF Vision 
Signal 



U.H.F. 
Tuner 



]>' 



UHF / | 

Sound Signal I 



Signal 



Sound-and-Vision i 
VHF Signals H 



V.H.F. 
Tuner 




/405 



I.F. 
Amplifier A 



VHF Vision 
Signal 



405 Ci 



UHF Sound 
- Signal 



6 625 



I.F. 

Amplifier B 



Video 
Detector 



VHF and UHF 
Video Signals 



Sound 
Detector 



.VHF and UHF 
Sound Signals 



r of the UiAMPUHCAWN SlAGi 
/*Mf British DuatStendertt Receiver 

Separating the VHF Sound and Vision Signals 

The first thing is to see how the VHF sound signal (on 38-15 MHz) is separated from 
the VHF' vision signal (3-5 MHz away on 34-65 MHz) before the latter reaches IF 
Amplifier A. Bear in mind that, as you will see later on, the frequency bandwidth of 
the sound signal is the comparatively narrow one of about 600 kHz, but that of the 
vision signal is very much wider. 

Separation of the two signals is accomplished by using in different ways members of 
a family of circuits which are generally called rejector circuits in Britain, and traps in 
the United States. Since you will be meeting quite a few of these circuits in the Dual- 
Standard receiver, it is worth taking a page or two at this point to see how they work. 

Traps are no more than LCR resonant tuned circuits, either series- or parallel- 
connected, consisting of inductors and capacitors plus a resistive element which can 
either be a physical resistor connected into the circuit or else the inherent resistance of 
the circuit components themselves. The four basic types are illustrated and explained 
on pages 2.86 and 2.87. 

Turn to those pages now to get the general principles into your mind. You will see 
that the important difference between the circuits lies in what they do to a signal whose 
frequency lies close to the resonant frequency of the trap. Circuits of Types A and B 
tend to pass signals of the resonant frequency while blocking or shunting signals of all 
other frequencies ; while circuits of Types C and D do the opposite, blocking or shunting 
signals of the resonant frequency while passing signals of other frequencies. 



§12] 



2.85 



Separating the VHF Sound and Vision Signals {continued) 

Bear the following points carefully in mind when you are thinking about the rejector 
circuits illustrated on the next two pages: 

(a) That when X L = X c in a series circuit, impedance to current flow is minimum. 
When X L = X c in a parallel circuit, impedance is maximum {Basic Electricity, Part 4). 

{b) That effective shunting only takes place when the parallel connection offers to 
the signal an impedance path which is low relative to the input impedance of the load 
across the output. This load is, of course, the next stage in the receiver. 

(c) That the terms "blocking" and "shunting" used in connection with rejector 
circuits are only relative. Although in all the illustrations on pages 2.86 and 2.87 an 
open switch has (for greater emphasis) been drawn to represent a very high reactance, 
and a closed switch to represent a very low one, it would have been more correct in 
both cases to draw in a resistance marked as being of high or low value respectively. 
Some current will always leak through even a very high Z; and neither inductive nor 
capacitive reactances are ever infinitely high. The "open circuits" shown should 
therefore be regarded as very high impedances severely attenuating the flow of current; 
while the "closed switches" represent paths of low relative impedance which most of a 
given current will take in preference to a path offering a higher impedance. 

The drawback to the basic types of trap shown on pages 2.86 and 2.87 is that they are 
not very selective. Their impedances alter only slowly as signal frequencies move 
away, in either direction, from the resonant frequency of the trap; and their frequency 
response curves tend to be rather low and flat. Such simple circuits are therefore not 
suitable for discriminating between signals whose frequencies lie comparatively close 
together; for the wanted signal may be severely attenuated (or, in parallel-resonant 
circuits, largely shunted) by meeting an impedance of inappropriate value. 

It is possible to design traps having much better selectivity and much sharper attenu- 
ation characteristics (that is to say, with a high and "peaky" response curve). Two 
such traps are shown below. The one on the left (A) is designed to block a particular 
unwanted signal whose known frequency is higher than that of the wanted signal; the 
one on the right (B) to block a signal whose known frequency lies below that of the 
wanted signal. 

L2 C2 

In » t TJfiT)?? 1 t-^*-Out In— »- 



C1 L1 



Out 



L1 




Trap A works as follows. The value of L 2 has been so chosen that, to signals of 
frequencies lower than the f T of the trap, its reactance offers little opposition. The 
signals therefore pass through the trap without attenuation. To the wanted signal 
itself, Ci and L x offer a predominantly capacitive alternative path, which still presents 
a high reactance. At a given frequency lying above that of the wanted signal, how- 
ever, it is arranged that the net capacitive reactance of the Ci-Li combination shall 
balance the inductive reactance of L 2 . The whole trap becomes a parallel-resonant 
circuit whose impedance at resonance is very high, and the unwanted signal is blocked. 

The working of Trap B is explained on page 2.88. 



2.86 



[§12 



RiJEGTOR ITS 



Input 



Output 




Series Connected: Series Resonant 



The Basic Circuit 

When Signal Frequency is Below Reson- 
ance, X L is low, but X c is high enough to 
Block the Signal. 

When Signal Frequency is Above Reson- 
ance, X c is low, but X L high enough to 
Block the Signal. 

When Signal Frequency is At Resonance, 
Xc = %l, and through the resulting low Z 
Maximum Signal Passes. 



B 



Input 



Input 



±< 



^St 



sr 






Output 



Output 




Parallel Connected: Parallel Resonant 



The Basic Circuit 

When Signal Frequency is Below Reson- 
ance, X L is low and provides a path of 
relatively low Z, so Shunting the Signal. 

When Signal Frequency is Above Reson- 
ance, X c is low and provides a path of 
relatively low Z, again Shunting the Signal. 

When Signal Frequency is At Resonance, 
X c = X L . The Parallel Combination offers 
a high Z. Little of the Signal is shunted, 
and Maximum Signal passes. 



§12] 



2.87 



OB TRAPS 



Input 
O 



Input 



kftMJU 

L 



Output 
O 



Output i 

_^_ • 

O 



Input 



--'— o~ 



Output + 



Input 
O 



Output 
O 



Series Connected: Parallel Resonant 



The Basic Circuit 

When Signal Frequency is Below Reson- 
ance, X L is low enough to pass Maximum 
Signal to Output. 

When Signal Frequency is Above Reson- 
ance, X c is low enough to pass Maximum 
Signal to Output. 

When Signal Frequency is At Resonance, 
X c = X L . The Parallel Combination offers 
a high Z, which Blocks the Signal. 



-o Parallel Connected: Series Resonant 



Input 



Output 



Input 



-V- 



Output f 



Input 



I 



-O 



\ 
Output I 
I 
/ 

^^6 



Input 



Output 
O 



The Basic Circuit 

When Signal Frequency is Below Reson- 
ance, X c is high and Maximum Signal 
Flows to Output. 

When Signal Frequency is Above Reson- 
ance, X L is high and Maximum Signal 
Flows to Output. 

When Signal Frequency is At Resonance, 
X c = X L , and the resulting low Z Shunts the 
Signal. 



2.1 



[§"2 



Separating the VHF Sound and Vision Signals {continued) 

Trap B in the illustration near the foot of page 2.85 works as follows. To frequen- 
cies above that of the wanted signal C 2 offers very low impedance, while the C x -L^_ 
combination offers a very high (and predominantly inductive) reactance. C u Z^ and 
C 2 are so chosen, however, that at a given frequency below that of the wanted signal, 
the net inductive reactance of d-Z-! becomes exactly equal and opposite to the capaci- 
tive reactance of C 2 . The trap becomes a parallel-resonant circuit at the frequency of 
the unwanted signal, and so gives it maximum attenuation. 

Series-resonant equivalents of the two traps described also exist, but you now know 
enough of the general way in which rejector circuits work to find little difficulty in 
understanding how one of them works when you meet it. 

All these traps are simple and cheap to manufacture, and they serve many purposes 
in the receiver adequately well. But it is often necessary to achieve much larger 
attenuation of an unwanted signal— and of an unwanted signal, moreover, whose fre- 
quency lies inconveniently close to that of a signal which you want to pass. More 
efficient traps having these capabilities are known as bridged-T rejector circuits. 

Bridged-T rejector circuits come in a good many variations, but the three shown 
below are among the most widely used. All, you will see, consist of a simple arrange- 
ment of inductors, capacitors and resistors, with one component always forming a 
"bridge" across the main circuit in the manner you briefly studied on pages 2.85-2.87 
of Basic Electricity. 



BRIDGCDT RMCCTOR CIRCUITS 




Out 




vW- 



ClT . TC2 



Out 



Each of the above traps offers maximum attenuation of the signal at a single fre- 
quency—which is the resonant frequency of the LC combination. The value of the 
resistor in each circuit is carefully chosen to ensure that the trap becomes "balanced" 
at this frequency, for in such a state of balance a bridge circuit has almost zero output 

The resistors used in bridge circuits in a TV receiver are usually of fixed value- but 
in other applications where low cost is less important a variable resistor is often 'used 
instead, so that precise balance can be achieved over a given range of frequencies. 

Bridged rejector circuits provide very sharply-tuned frequency response curves and 
possess an unusually high effective impedance at resonance. They are therefore useful 
for removing (or very severely attenuating) an unwanted signal whose frequency lies 
close to that of the wanted signal; and they can even be used for "cutting out" a 
narrow unwanted slice from a frequency response curve altogether. 



§12] 2.89 

Separating the VHF Sound and Vision Signals (continued) 

Now that you know in principle how rejector circuits work, you will soon grasp how 
the VHF sound and vision signals are separated from one another at the input of IF 
Amplifier A in the Dual-Standard receiver. 

Remember that no electronic circuit can differentiate between wanted and unwanted 
signals fed to it, unless it has been designed to do so. It will react according to its 
nature, irrespective of whether its input is hopefully labelled "sound signal", "vision 
signal" or anything else. Thus if a portion of the sound signal reaches the picture 
tube of a TV set, the latter will react just as if this signal were part of the wanted vision 
signal. A condition known as sound-on-vision will arise ; and the picture appearing on 
the screen will be distorted. 

In practice, sound-on-vision is so serious a danger that a high-grade rejector circuit 
is used to combat it. A bridged-T rejector resonant to the frequency of the sound signal 
is therefore connected in series with the circuit leading to the main part of IF Amplifier 
A (it is labelled F3 in the full circuit diagram on page 2.99), and the sound signal is 
effectively blocked by it from reaching that section. 

The converse effect of vision-on-sound is accepted as having a less damaging practical 
effect; and a simple parallel-resonant trap connected across the line taking the VHF 
signal to IF Amplifier B is generally considered sufficient. This trap also (Fl in the 
diagram on page 2.99) is made to resonate at the frequency of the sound signal. With 
its moderately good Q, it passes the 600 kHz bandwidth of the sound signal itself quite 
satisfactorily, but shunts all signals whose frequencies lie outside this band (including 
most of the VHF vision signal) to earth. 

The Requirements of IF Amplifier A 

You are now left with the principal section of IF Amplifier A having two inputs— the 
VHF vision signal and the UHF sound and vision signals. What requirements must 
it fulfil in amplifying these signals acceptably? 

It must do two things. 
A It must deal now, at a moment when they are still quite weak, with certain inter- 
ference signals which would be much harder to eliminate at a later stage when they had 
been amplified to greater strength. 

A It must apply a reasonably even degree of effective amplification over the opera- 
tional bandwidth of a signal which has had one of its sidebands partially suppressed. 

Let's deal with the interference problems first. You saw in Part 5 of Basic Elec- 
tronics (pages 5.48 and 5.56) how image-frequency (or second-channel) interference can 
be caused by the intrusion of unwanted signals on a frequency twice the value of the i.f. 
above or below the frequency of the wanted signal. (This is true of any type of 
receiver working on the superhet principle, whether it be used for sound radio, for TV 
or for anything else.) You also saw that it was possible to reduce the possibility of 
image-frequency interference by arranging for a high intermediate frequency to be 
produced in the mixer stage; but that this could only be done at the cost of reducing 
the selectivity of the receiver. 

In the British Dual-Standard receiver, the i.f.'s used are high enough to cut out most 
second-channel interference; but they are so high that the opposite danger of adjacent- 
channel interference becomes a real problem calling for special measures at the i.f. 
amplifier stage. 



2.90 



[§»2 



IF Amplifier A — The Requirements (continued) 

Adjacent-channel interference is introduced into the signal by unwanted signals 
whose frequencies lie close to its own. An example would be a TV receiver operating 
in Channel 3 but suffering interference from Channels 2 and 4. The very strong sound 
carrier signals radiated by the BBC in Channel 1 are also apt to intrude on signals 
several MHz away from them in frequency. 

These interference signals mix with the output of the local oscillator in the tuner, 
and produce additional, unwanted, sound and vision i.f.'s which pass into the IF 
Amplifier stage above and below the i.f.'s of the signals in the desired channel. 

The illustration below shows the frequency response of a typical TV tuner. You 
will see that the curve is broad enough, and tapers away slowly enough, for unwanted 
i.f.'s to be present in appreciable strength at the input to the IF Amplifier. The 
sharpening of the frequency response effected by the i.f. tuned circuit in the output of 
the mixer stage has not been enough; and it is an essential part of the job of the IF 
Amplifier to improve the shape of the curve (and so to sharpen the Q of the receiver) 
to its final form. 



FRCWCHCy 
SPiCTMM 

of the Input Signal 

to the 

I.F. Amplifier Stage 

(625-line System) 



Think of the frequency spectrum above as the "raw material" with which the IF 
Amplifier stage is presented when the receiver is switched to the 625-line system. (A 
similar curve, though with some differences, is presented to the IF Amplifier when the 
set is switched to 405 lines.) What you want to extract from these two curves are the 
two frequency spectra shown in the double diagram on the opposite page: they are, 
respectively, the "true" spectra of the sound and vision i.f. signals in the VHF system 
and the "true" spectra of the same signals in the UHF system. 

Note how both curves show maximum signal intensity throughout those sidebands 
which are transmitted in full, but maximum signal intensity only on those parts of the 
other sideband which are not suppressed. Note, too, how sharply the curves fall away 
beyond these limits. 

Note also (shown in dotted line) the positions in the frequency spectrum which the 
nearest of the unwanted i.f.'s— the upper and lower adjacent channels— would in each 
case occupy if they were present; and also the position in each spectrum of the respec- 
tive sound carriers. (The bandwidths of both sound carriers have, of course, been 
greatly exaggerated in relation to the widths of their corresponding vision signals, in 
order to show them at all.) 




§12] 



2.91 



IF Amplifier A — The Requirements (continued) 

In both the diagrams below, frequencies below and above the vision carrier have 

been marked with (-) and (+) values respectively. Remember that these scales do 

not aim to show the actual magnitude of either signal, but only its value relative to the 

frequency of its vision carrier. 

UHF 
Sound Carrier 



Lower 

Adjacent 

Channel 




Upper 
L Adjacent 
'^Channel 



VHF 
Sound Carrier 



VHF 
Vision Carrier 
i I 



Lower 

Adjacent 

Channel 



Upper 

Adjacent 

Channel 




HO-5 L*_ Lower Sideband — »«J 
MH *~ (3 MHz) ^ 



Upper Sideband 
y (0 75 MHz) 



UHF 



VHF 



Imagine that you are the designer of the IFA stage of a receiver which has to handle 
two signals having the frequency spectra shown in the double illustration above. 
What frequency response are you going to build into the IFA to ensure that it will 
provide the correct distribution of signal power, after i.f. amplification, over the entire 
spectrum of both signals ? 

The problem centres, of course, round those partially suppressed sidebands. You 
will recall from Part 1 that, in the British 625-line system, the upper sideband of the 
vision carrier is transmitted in full up to 5-5 MHz, whereas the lower (vestigial) side- 
band is transmitted in full only up to 1-25 MHz and is suppressed thereafter. In the 
405-line system, the lower sideband is transmitted in full up to 3 MHz, while the upper 
(vestigial) sideband is suppressed after 0-75 MHz. 

What is the real difficulty which these partially suppressed sidebands present? 



2.92 



[§12 



IF Amplifier A — Signal Power Distribution 

You learnt in Basic Electronics that it is the sidebands of any AM wave which carry 
the signal information, and which contain the signal power. You also know, from 
Part 1 of this Series, that it is the higher frequencies, those furthest away from the vision 
carrier, which are needed to reproduce on the screen the sharp variations between 
black and white which constitute the fine detail ("highlights") of the picture. The 
lower frequencies, those nearer the carrier, reproduce only the varying shades of grey 
which go to make up picture background. (Recall the much wider bandwidth 
required to reproduce the finish of the 100 metres at the Olympic Games than that 
needed to represent the angling competition on the pier at Southend.) 

You also know that, with the aim of limiting overall bandwidth, most of one of the 
sidebands of a TV transmission is suppressed. Part of the process of doing so is to 
give zero amplification to the frequencies carrying the unwanted bits of the sideband. 

But with a partially suppressed sideband a difficulty arises at the receiver over pic- 
ture signal distribution. The remaining part ("vestige") of a partially suppressed 
sideband is the "grey-reproducing" section of it which lies nearest to the carrier. So if 
you apply 100% amplification to the full range of frequencies received, you will have 
two frequency ranges (one above and one below the carrier) transmitting "grey" back- 
ground, to only one range of frequencies transmitting picture highlights. The result- 
ing over-emphasis on background at the expense of action will look very unnatural on 
the screen. 

The solution adopted is to arrange for IF Amplifier A to give varying degrees of 
amplification to the band of frequencies lying closest to the vision i.f. carrier on its 
either side. Minimum amplification is applied to the received signal at the limit of the 
partially suppressed sideband, and the degree of amplification applied then rises 
linearly until it reaches 100% at the equivalent point on the other sideband (which is, 
of course, being transmitted in full). The vision i.f. carrier itself lies (theoretically) 
half-way up this slope, at the point at which 50% of the maximum amplification is 
applied to the signal. 

This solution will be easier to understand if you look at the diagram below. It 
shows the theoretically-desirable frequency response curve of the 405-line vision i.f. 
amplifier when allowance is made for the partial suppression of the upper sideband of 
the signal. (It is one of the difficulties of explaining this business that the polarities 
of both sidebands have at this stage been reversed by the local oscillator in the tuner, 
so that the sideband whose values are marked with (+) values in the diagram is in fact 
the lower sideband, and vice versa.) 



Vision I.F. Carrier 
34-65 MHz 



c too f 



E 
< 



Sound I.F. Carrier 
38 15 MHz 




The 

Frequency Response Curve 

theoretically desirable 

for the 

40S*UNe VISION 

U StCN/U 



I Upper I 
1 Sideband T 



♦ 2 

Frequency (MHz) 
Lower Sideband 



H 



§12] 



2.93 



IF Amplifier A — Signal Power Distribution (continued) 

You will see from the diagram on the last page that no problem arises with signal 
frequencies more than 0-75 MHz away from the carrier. Full amplification is applied 
to all frequencies between ( + ) 0-75 MHz and ( + ) 3 MHz, and zero amplification to all 
frequencies below ( — ) 0-75 MHz. 

Within the range (-) 0-75 MHz to (+) 0-75 MHz, however, the degree of amplifica- 
tion applied varies. As soon as frequencies in the partially-suppressed upper sideband 
start receiving a degree of amplification, so it becomes necessary to reduce from 100% 
the degree of amplification applied to the corresponding frequencies on the other side 
of the i.f. carrier. Thus a signal whose frequency is, say, 0-50 MHz above that of the 
i.f. carrier will receive about 80% of full amplification, while the corresponding signal 
whose frequency is 0-50 MHz below the carrier will receive only about 20%. 

At all other points on the slope, signals whose frequencies lie similarly equidistant, 
(+) and (-), from the i.f. carriers likewise receive degrees of amplification which add 
up to 100% of maximum. In other words, the gross amplification applied over the 
(-) 0-75 to ( + ) 0-75 MHz range of the two sidebands is always 100%, and the initial 
power distribution of the transmitted signal is accurately preserved. 

Exactly the same considerations apply to the basic frequency response curve desired 
of IF Amplifier A when the receiver is operating on the 625-line system. In the 
diagram below, the linear slope in the response curve starts dropping at (-) 1-25 
MHz, and the percentage of full amplification applied to the signal falls to zero at ( + ) 
1 -25 MHz. At all points on the slope, the gross amplification applied to "correspond- 
ing" frequencies above and below the carrier adds up to 100%. 

The Frequency Response Curve theoretically desirable for the 



Sound I.F. 
Carrier (33-5 MHz) 



625-LINE VISION I.F. SIGNAL 



-125 MHz 




Vision I.F. 
Carrier (39-5 MHz) 
.1 



♦ 1-25 MHz 



--100 



—i 1 ' T 

-3 -2 

Frequency (MHz) 



Upper Sideband 




i— Lower - 
Sideband 



Note that, once again, the polarities of all frequencies have been reversed by the LO 
in the tuner, so that values in what looks like the lower sideband are shown as ( + ) 
quantities, and vice versa. It is better to put up with this awkwardness than to risk 
forgetting which of the sidebands in each system is partially suppressed, and which is 
transmitted in full. 

Observe the deviation in the desired linear fall of the frequency response curve at the 
frequency (33-5 MHz) of the sound i.f. carrier in the 625-line system. It is essential to 
keep the value of the sound signal low at this stage, lest it modulate the vision signal in 
the common amplifying stages through which both signals are soon to be put. 



2.94 



»I2 



IF Amplifier A — The Frequency Response Curve in Practice 

You have seen that two very different shapes are required for the frequency response 
curves of IF Amplifier A for the 405- and 625-line signals. Since the Dual-Standard 
Receiver must be able to accept signals on either line standard, it must either possess 
completely separate i.f. amplifiers for each, or else have common amplifying circuits 
whose frequency curves can be adjusted to suit the particular signal being received. 
For obvious reasons of simplicity, compactness and economy, the latter alternative is 
preferable if it can be achieved. 

IF Amplifier A, as you know, has to handle signals which are very similar in fre- 
quency (405-line vision at 34-65 MHz; and 625-line vision at 39-5 MHz and sound at 
33-5 MHz). It therefore cannot rely on wide frequency differences to select auto- 
matically the necessary tuned circuits when the viewer switches from one standard to 
another by rotating the Standard Selection control. What is needed instead is a 
response curve of the right shape to suit the wide bandwidth of the two UHF signals, 
together with means of narrowing and re-shaping the curve to the narrower bandwidth 
and different dimensions required by the VHF vision signal. 

You will learn about the technical means used to achieve this in later pages. Here, 
meanwhile, is the variable response curve produced by a typical Dual-Standard 
Receiver of the present day (though you should note that a good many perfectly work- 
able variations on the pattern also exist). 



625 Sound 405 Vision 
I.F. Carrier I.F. Carrier 




Frequency (MHz) 



The F*£<Wettcy *£$POHS£ *f <r typical 

DuolStttnttntd U Amplifier 



§12] 2.95 

IF Amplifier A — The Frequency Response Curve in Practice (continued) 

The Dual-Standard i.f. amplifier whose actual frequency response is pictured on the 
page opposite is a wideband amplifier having an unrestricted bandwidth of 4-85 MHz 
extending from 34-65 MHz (the frequency of the VHF vision i.f. carrier) at one end to 
39-5 MHz (the frequency of the UHF vision i.f. carrier) at the other. The response 
curve produced by the amplifier when the Receiver is switched to the 625-line standard 
is drawn in solid unbroken line in the diagram; while the curve produced when the 
receiver is switched to the 405-line standard appears as a dotted line. 

Note how the curves differ from the theoretically desirable shapes shown in the 
illustrations on pages 2.92 and 2.93. It is quite possible to achieve more nearly the 
sharply angular response curves there shown. Indeed, much sharper curves are essen- 
tial to the proper performance of, e.g., a high-performance radar receiver. But the 
circuitry required to achieve them is too complex and costly to be installed in a mass- 
produced TV receiver whose price must be kept down to a figure which the public will 
pay. All commercial TV receivers involve many such compromises between the tech- 
nically ideal and the economically possible; and response curves of the shape pictured 
opposite have been found to give results good enough to "get by" with the average 
viewer. 

Note another important feature of these two response curves. In both of them the 
vision i.f. carriers occur at a point rather higher up their respective curves than the 
"half-way down from maximum amplification" which you would theoretically expect. 
In the curves shown, both carriers occur at about the point of 67% relative response, 
or "4 dB down from maximum". 

The object of thus altering the theoretically ideal positions of the vision i.f. carriers 
on their respective response curves is to reduce an effect called quadrature distortion 
which is introduced in the vision detector in the process of handling a vestigial side- 
band transmission. The causes and cure of quadrature distortion are subjects frankly 
beyond the scope of this Series; but its effect is to cause streaking in the signals rep- 
resenting white (in systems using negative modulation) and in the signals representing 
black (in systems using positive modulation). In both cases the result is a noticeable 
loss of resolution. 

It is possible to correct quadrature distortion by careful "counter-distortion" of the 
transmitted signal; but its harmful effects are also in practice reduced if the vision i.f. 
carrier is made to occur at a point rather higher up the response curve than it theoreti- 
cally should. 

The penalty to be paid is, as you will have guessed, some over-emphasis of picture 
detail compared with the "grey" background on the screen; but this has been found to 
give very little impression of unreality in practice. 

When the viewer switches from the 625-line to the 405-line standard by operating the 
"Standard Selection" control on the front of his receiver, the 4-85 MHz width of the 
response curve of the amplifier is narrowed by compressing its high-frequency end, and 
by re-shaping its low-frequency end so as to get rid of the "10% bump" which you 
know is required by the 625-line sound i.f. carrier. The resulting bandwidth of about 
2-85 MH (measured at the "4 dB down" points) is found adequate for the frequency 
content of the VHF signal. 

The high-frequency end of the curve is brought sharply down to zero in the region of 
the 405-line sound i.f. carrier frequency in order to keep that signal out of IF Amplifier 
A. 



2.96 [§I2 

Shaping a Response Curve 

The next step is to see how a response curve of unsatisfactory shape can be altered 
into something more desirable. 

You already know that the frequency response of a tuned circuit can be varied by 
placing varying amounts of resistance across the circuit. Adding resistance to a cir- 
cuit for this purpose is called damping, and the more heavily a tuned circuit is damped, 
the more is the curve of its output frequency flattened. 

The illustration below shows the effect of damping of varying degrees of severity on 
an ordinary tuned circuit. You will see that, as damping is increased, so both the 
sharpness of tuning (i.e., the selectivity, or Q, of the circuit) and the amplitude of its 
peak response are reduced. 

On the other hand, the width of the response curve is increased, even though its 
extremities become too ill-defined to be (normally) of much use. 




Damping 
Resistor 



Frequency 



P4M* v,'t 



In other words, though damping techniques can be used for broadening a given 
response curve, the price to be paid is loss of peak amplitude and loss of Q. 

Another method of altering the shape of a response curve to something more nearly 
what is desired is described on the page opposite. The illustration shows the effects of 
varying the mutual coupling of a pair of tuned circuits. 

Physically, the mutual coupling of a pair of coils can be varied by moving the coils 
either closer together or farther apart. Alternatively (and more usually) it can be 
done by altering the positions of the two metal cores within a former on which both 
coils are mounted. In both cases the patterns of flux produced by the interacting 
magnetic fields of the two coils are altered by the movement. 



§12] 



2.97 



Shaping a Response Curve (continued) 

The frequency response shown in Curve 1 in the illustration below results from a 
very weak (or "loose") coupling, produced either by the two tuned circuits being 
moved far apart or by their cores being moved away from their respective coils. The 
circuits are said to be under-coupled, and the amplitude of peak response is low. 

Mutual 

Coupling 




Frequency 

The Effect of vtry/tif the MUTUAL COUPUHG 

of Two Tuned Circuits 



Curve 2 is produced by critical coupling. Again the response curve has a single 
peak, but it is steeper and of much greater amplitude than before. 

If coupling is made closer still, the single peak of Curve 2 begins to broaden out into 
the double-humped Curve 3. This is now an over-coupled curve ; and as over-coupling 
is increased, so the two humps become emphasized into two distinct peaks. 

Curve 4 is produced by introducing resistive damping into the tuned circuits pro- 
ducing these two peaks. The overall amplitude of the curve is reduced, but its top is 
much flattened. A flat-topped, relatively steep-sided curve of this type is of excellent 
shape for use as a band-pass filter. It can be made to pass a fairly broad band of 
frequencies giving a near-equal degree of amplification to each, but its steep sides 
impose sharply defined limits to the band of frequencies passed. 

If a near-rectangular shape of curve having a higher peak amplitude than Curve 4 
can give is desired, it is possible to "fill in the valley" between the two humps of the 
over-coupled Curve 3 by means of a following stage of amplification having a single- 
peaked response curve whose amplitude is maximum at a frequency which coincides 
with that producing the valley in the preceding stage. 

It is possible to produce response curves of almost any desired shape by means of the 
techniques described above, and by using (if necessary) a number of cascaded amplifying 
stages in the shaping circuit, each having a response curve carefully selected to play its 
part in producing a final curve of the exact shape required. 



2.98 [§12 

IF Amplifier A — Operation 

You are now ready to study in detail the circuit diagrams of the two i.f. amplifiers 
used in the British Dual-Standard Receiver, and to follow out how each works. 

Take, first, the one we have called IF Amplifier A. Its function is to amplify the 
VHF vision signal and also the sound and vision signals which are delivered to it from 
the UHF tuner on the 625-line standard. 

The Amplifier consists essentially of two high-gain amplifying valves, plus a number 
of coupling transformers, plus also a number of rejector (trap) circuits which are 
switched into or out of circuit according to which line standard is being received at any 
given time. 

The Standard Selection switch operates (as you know) at many points in the cir- 
cuitry of the Dual-Standard receiver when the viewer turns or presses the appropriate 
control on his set. No fewer than five sections of it operate within the circuits com- 
posing IF Amplifier A. They are labelled S1-S5 in the circuit diagram on the page 
opposite. 

Both the valves (VI and V2) are of the frame-grid type and have very large values of 
mutual conductance (G m ) — typically greater than 13 mA/V. This high ability to 
amplify at the anode an input to the control grid enables the two valves to do a job 
which used to require three of the older type of pentode, whose values of G m were no 
more than about 7-5 mA/V each. 



Operation on the 405-line standard 

All sections of the Standard Selection switch are set to their "405" positions, and S4 
passes HT to the VHF tuner. The input signal to the Amplifier reaches SI from the 
tuner (usually, you will recall, through a short length of coaxial cable forming part of 
the coupling between two bandpass coupling elements), and is applied to the junction 
between the capacitor CI and the trap F3. 

F3 is a bridged-T rejector circuit tuned to resonate at the frequency of the sound signal 
(38- 1 5 MHz). It thus offers a very high impedance to this signal, and prevents it from 
reaching VI . Presented with this high Z in the direction of F3, the sound signal takes 
the path of much lower impedance through CI and passes on its way to IF Amplifier B. 

F3, however, offers very little impedance to a signal of the frequency of the VHF 
vision signal, which passes through it virtually unimpaired. 

At the farther end of F3, the vision signal meets two more rejector circuits connected 
in shunt (F4 and F5), of which F5 is of the highly selective rapid-action type. F4 is 
tuned to resonate at 39-65 MHz, which is the frequency of the vision signal in the upper 
adjacent channel. (Channels on the 405-line standard are 5 MHz wide; 34-65 + 5 = 
39-65 MHz.) A series-resonant tuned circuit such as F4 offers a very low impedance 
to a signal of the frequency to which it is tuned ; so that the unwanted adjacent-channel 
interference is shunted away through F4 to earth, whereas the wanted vision signal 
passes unimpaired on its way. 

F5 is tuned to resonate at 33-15 MHz, the frequency of the sound signal in the lower 
adjacent channel (38-15 - 5 = 33-15 MHz). Acting in the same way as did F4, it 
blocks or greatly attenuates this unwanted signal, while allowing the wanted vision 
signal to continue on its way through S3. 



§12] 



2.99 



The Dual-Standard I.F. Amplifier "A 



J 


I 2k 




0) 
— O- 

CD 


. 




-C o 




+- <D 








£> 


1 w 










UoJ 


i . 










2.100 [§I2 

IF Amplifier A — Operation (continued) 

After S3, the vision signal encounters yet another rejector circuit, F6, this time a 
series-connected, parallel-tuned trap of the quick-acting type. F6 is tuned to 41-5 
MHz, which is the frequency of the very powerful sound signal in Channel 1 of Band 1. 
This signal is often present at quite large strength despite the nearly 7 MHz difference 
between its frequency and that of the VHF vision i.f. signal; and it must be kept off the 
picture-tube of the receiver lest it spoil the picture. 

Traps of the F6 type offer a very high impedance indeed to signals of the frequency 
to which they are tuned, but pass all other frequencies practically unimpaired. 

The wanted vision signal next encounters the coil L7 which, with its own stray 
capacitances, forms a parallel-tuned circuit tuned to a frequency of 34-3 MHz. 
Though it does not look like it on the circuit diagram, L7 physically forms the second 
half of a bandpass coupling circuit whose first half is a similar tuned circuit in the out- 
put stage of the tuner. 

This tuned circuit in the output stage of the tuner is tuned to resonate at a frequency 
of 37-15 MHz, with the result that the coupling circuit as a whole passes a bandwidth 
of some 2-85 MHz. As you learnt on page 2.95 (and saw on the illustration on page 
2.94) this is the bandwidth which is required for good reception on the 405-line stan- 
dard. 

The reactance of the capacitor C12 contributes to the coupling between the two 
tuned circuits mentioned above. 

The VHF vision signal is considerably amplified by VI, and is then developed across 
the primary (L8) of the bandpass transformer which forms its anode load. The tuned 
circuit formed by L8 and its stray capacitances resonates at 36-4 MHz; the secondary 
tuned circuit (L9 plus stray capacitances) resonates at 34-75 MHz. 

From the secondary of this bandpass transformer the vision signal passes for further 
amplification in V2, from which it emerges at an amplitude of some 5 volts. V2 anode 
load is another bandpass transformer, formed by L10 and LI 1 . L10, plus stray capa- 
citances, resonates at 37-2 MHz: Lll (plus stray capacitances) at 35 MHz. 

The two bandpass transformers (L8-L9 and L10-L1 1) together thus pass an overall 
bandwidth extending from 34-75 to 37-2 MHz. If this 2-45 MHz bandwidth had been 
achieved by use of a single bandpass transformer, the signal would have suffered an 
unacceptable degree of attenuation. The use of two such transformers provides an 
example of the type of cascading arrangement which you read about on page 2.97. 

The two bandpass transformers in question are said to be "stagger-tuned" (see Basic 
Electronics, page 6.29). Together, they pass a bandwidth of the required width with- 
out undue loss of amplification. 

The amplified signal appearing across L10 and Lll is inductively coupled to a ter- 
tiary winding (LI 2) on the same transformer, from which it is transferred to the Video 
Detector. 

The primary winding L10, however, is also connected through S5 to another sound- 
rejector circuit, F7. Formed by C24-L13, this circuit is tuned to resonate at 38-15 
MHz, the frequency of the sound i.f. on the 405-line standard. Any signal whose fre- 
quency lies at or near 38-15 MHz will therefore find an easy path to earth through F7. 
The presence in the circuit of F7 therefore serves to sharpen-up the high-frequency end 
of the response curve of IF Amplifier A (see page 2.94), and to prevent it trailing 
slowly away to zero in the way it is required to do on the 625-line standard. 



§12] 2.101 

IF Amplifier A — Operation (continued) 

AGC voltage is applied only to VI in this Amplifier A circuit. You will see later 
on, when you study how this voltage is generated, that it varies in value according to 
the strength of the received signal. Therefore, when the AGC voltage is fed to the 
control grid of VI (through the low-pass filter formed by R7, C15 and the grid resistor 
R4), it affects the input capacitance of the valve as its own value varies. Some degree 
of negative feedback to compensate for this variation in input capacitance is provided 
by means of the undecoupled 22-ohm resistor R5 connected in series with the cathode 
bias components (R6-C17) of the valve. 

AGC voltage is not applied at all to V2, for it is essential that this valve shall main- 
tain a steady mean anode current in order to be able to handle the now-large vision 
signal without distorting the peak amplitude variations of the waveform. Any such 
distortion at this stage would be passed on as genuine by the video detector, and would 
impair the picture presented to the viewer. 

Observe (at the foot of the circuit diagram on page 2.99) that the AGC voltage fed 
to VI is the same voltage as that which you saw being fed to the tuner in the last 
Section. 

Operation on the 625-line standard 

When all sections of the Standard Selection switch are set to "625", HT is fed to the 
UHF Tuner through S4, and the input from the tuner is re-routed by S 1 . The capaci- 
tor CI is shorted to earth by the action of S2 so as to prevent the 6 MHz intercarrier 
signal which is now present in the circuits of IF Amplifier B from intruding into the 
circuits of IF Amplifier A. 

Remember that the UHF sound and vision signals remain unseparated all the way 
through IF Amplifier A. They first pass from SI to the series-tuned rejector Fl, 
which resonates at a frequency of 31-5 MHz. This is the frequency of the adjacent- 
channel vision signal, which at UHF occurs 8 MHz below the 39-5 MHz frequency of 
the vision signal itself. Fl offers very low impedance to a signal of its own resonant 
frequency, so the interference is virtually shorted to earth. 

The next rejector, F2, performs an important special function on this line standard. 
This is to reduce the level of the sound component of the combined UHF signal by as 
much as 20 dB, thus lowering it to only about 10% of its former value. This needs to 
be done before any amplification of the two signals takes place; for a too-large element 
of sound can easily impose unwanted modulations on the vision signal as the two sig- 
nals pass through their common amplifying stages. This would cause a kind of dis- 
tortion known as cross-modulation, which cannot be corrected at a later stage. 

To do its job, F2 (a series-connected trap of the rapid-action type) is tuned to reso- 
nate at the 33-5 MHz frequency of the wanted sound signal. If the efficiency of the 
trap were perfect, the sound signal would be suppressed altogether; but sufficient cir- 
cuit losses are deliberately built into the design of the filter to provide an overall signal 
attenuation of only about 90%. 

After passing Fl and F2, the two UHF signals reach the rapid-action rejector F6, 
which (you will recall) used its resonant frequency of 41-5 MHz to block the intrusive 
sound signal in Channel 1 when the Receiver was set for 405-line reception. It now 
uses this frequency to offer a similarly high impedance to the sound signal in the upper 
adjacent channel, which at UHF lies 8 MHz above the 33-5 MHz frequency of the 
sound signal itself. 



2.102 [§I2 

IF Amplifier A — Operation on the 625-line Standard {continued) 

You saw at the foot of the last page that the job of F6 is to block the 41-5 MHz 
sound signal in the upper adjacent channel. It is far more important to suppress this 
intruding sound signal when the receiver is set to 625-line operation than it was on 405 
lines, because the unwanted frequency now lies much closer to the all-important vision 
i.f. of 39-5 MHz. Hence the much greater importance of F6 at UHF. 

After passing F6, the signal is handled in the same way as was the 405-line vision 
signal until it has received its double measure of amplification in the two valves and 
has been developed across the transformer L10. 

On "625", however, the sound rejector F7 is disconnected by S5 from the "earthy" 
end of L10, and its place is taken by the 39 pF capacitor C23. The purpose of this is 
to allow the bandwidth of the transformer to open out to its full width, and its high- 
frequency end to decay appreciably more slowly than it did on "405" (see illustration 
on page 2.94). 

The output signals (the UHF vision and sound i.f.'s) are again taken from the ter- 
tiary winding (LI 2) to the Video Detector. 

The purpose of the little group of components R9-C18-C19 in the grid circuit of V2 
is to prevent a phenomenon known as AGC blocking, which you will learn more about 
when you study the vision AGC circuit. Briefly, it can occur when a sudden large 
signal is applied to the Video Detector, resulting in a loss of AGC to IF Amplifier A 
and a corresponding excessive flow of current through V2 (and through certain other 
valves in the receiver as well). 

The danger is met by the combination R9-C19, the working of which will be ex- 
plained in the Section explaining Automatic Gain Control. 

The very-small-value capacitor CI 8 serves to counteract the inductive properties 
which the large- value capacitor C19 begins to acquire at frequencies as high as those 
handled in this Amplifier. At these frequencies, large-value capacitors of the electro- 
lytic type tend to behave as much like coils as like capacitors ; and it becomes necessary 
to nullify this effect by connecting in parallel with the electrolytic capacitor another, 
much smaller, capacitor whose value of inductance, even at these frequencies, is almost 
zero. 

IF Amplifier B — Operation 

The full circuit diagram of IF Amplifier B is shown on the next page, and the working 
of the circuit is explained on the pages following. 

IF Amplifier B is a sound amplifier only. Its inputs are the 38-15 MHz sound signal 
of the 405-line system (which you saw being separated from its vision signal by the 
rejector circuit F3 in the circuit diagram on page 2.99), and the intercarrier signal of 
6 MHz frequency which you have been told is fed back from the Video Detector to IF 
Amplifier B in the 625-line system. 

This intercarrier signal, as you will learn later on, is a "beat frequency" derived from 
the heterodyning of the amplitude-modulated vision i.f. signal with the frequency- 
modulated sound i.f. signal. The intercarrier therefore carries modulating com- 
ponents from both signals, being amplitude-modulated by the vision i.f. carrier and 
frequency-modulated by the sound i.f. carrier. It is the sound signal that interests you 
in IF Amplifier B, and the amplitude modulations imposed on the intercarrier signal 
by the vision i.f. carrier are a nuisance which must be suppressed. 



§12] 

The Dual-Standard I.F. Amplifier "B" 



2.103 



.1, 3 ° 

cm trt ° 

I— u-O 



O 

— >/w 



UJU 






lis* 1 

i2<ci 



3m 



r^r 



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o 



J^L 







CO 



CB 






3 



2.104 



[§12 



IF Amplifier B — Operation (continued) 

The reason why the amplitude modulations imposed on the 6 MHz intercarrier 
signal are a nuisance which need to be suppressed is this. You will recall from Basic 
Electronics, pages 6.32 and 6.33, that a commonly-used way of deriving an audio signal 
from a frequency-modulated carrier is to use a frequency discriminator circuit, and that 
such a discriminator is very sensitive to variations in the amplitude of the signal. If 
these are not adequately suppressed, the discriminator will reproduce them as a form 
of noise on the audio signal which (in this method of signal processing) is known as 
intercarrier buzz. 

IF Amplifier B must therefore provide circuitry capable of sharply limiting the amp- 
litude of i.f. carrier variations when the receiver is switched to "625". 

IF Amplifier B needs little help from the Standard Selection switch to perform its 
function as a dual-frequency tuned amplifier. This is because of the big difference 
between the two frequencies (38-15 MHz and 6 MHz) which it has to handle. 

To frequencies as widely separated as these, the same tuned circuit reacts in very 
different ways. It is therefore possible to connect in series two parallel-tuned circuits, 
each resonant to one of the desired frequencies, and to pass signals on either line- 
standard through the combination. Each tuned circuit blocks the frequency to which 
it is tuned, but behaves like a very low capacitive or inductive reactance to the other 
frequency passing through the combination, so having almost no effect on it. 

If, therefore, you have two signals of widely different frequency, it is only a matter of 
feeding the two signals into the appropriate part of the double trap, and you are bound 
to be able to pass one signal and to block the other. You will see the technique in 
action shortly. 

The frequency responses required from IF Amplifier B are very similar in shape 
whether the receiver is switched to "405" or to "625", though the frequencies them- 
selves (and the bandwidths) are of course different. The illustration below shows 
side-by-side the response curves produced by a typical sound IF Amplifier on each of 
its two settings. 



00- 


38-15MHZ 
37-85 MHz l 


38-45 MHz 




80- 


I / ! 
1 / 1 
' / 1 


\ ' 6 


db 


60- 


1/ ' 


\| 






— 4— f- 
/ i 
/ i 


J_ ■ 


' 


40- 




20- 


/ i 
/ i 




O- 


/ i 





Frequency (MHz) 
ON 405 LINES 




Frequency (MHz) 
ON 625 LINES 



FREQUENCY RESPONSE of a typical SOUND I.F. AMPLIFIER 

Note that the bandwidth of the 405-line curve is a full 600 kHz at the two points 
'6 dB down from maximum amplification". 



§12] 2.105 

IF Amplifier B — Operation (continued) 

So wide a bandwidth as 600 kHz for the 405-line curve seems at first sight surpris- 
ingly large. Few people can hear audio frequencies higher than 15 kHz, so that the 
sideband frequencies created by the audio signal need seldom exceed ±15 kHz above 
and below the frequency of the carrier which it is modulating. A bandwidth of little 
more than 30 kHz would therefore seem quite adequate. 

At "625", it is true, a bandwidth of at least 100 kHz is essential to accept the ± 50 
kHz frequency deviations of the i.f. carrier caused by the AF modulations superim- 
posed upon it; but a bandwidth of 600 kHz on "405" seems far wider than is needed. 

Two significant advantages, however, plus a third potential advantage, arise from 
using so wide a bandwidth. First, the effects of frequency drift in the local oscillator 
in the tuner are almost eliminated (if LO frequency drifts, so will the intermediate fre- 
quency which it produces when it beats with the incoming signal). This is particularly 
important when the receiver is operating in Band 3, where the frequencies involved are 
comparatively high. A small percentage shift in a Band 3 frequency can easily 
amount to 200 kHz, so it is highly desirable that the i.f. carrier should have room in 
which to drift a bit within the available bandwidth. 

The second advantage of a wider bandwidth is that the shape of sundry noise pulses 
is thereby kept sharply defined, which makes their rejection in a later stage much easier. 

The third (potential) advantage is that the quality of the sound reproduction could 
be made greatly superior to that of the ordinary AM sound radio receiver, whose trans- 
mission bandwidth is restricted by international agreement to about 10 kHz only. 
Few TV receivers exploit this potential advantage, however, for reasons of cost; and 
quite low-quality output transformers and loudspeakers are in fact standard fittings. 

Now look at the circuit diagram of IF Amplifier B on page 2. 1 03 . Of the two valves 
shown, VI is a high-gain pentode of frame grid construction, V2 a pentode of the 
normal type. 

The amplifier has two input terminals, for its input signals come (as you know) from 
two different sources. When the Standard Selection switch is set to 405-line operation, 
the "625" intercarrier signal is not generated at all. Nor is the 405-line sound i.f. 
generated when the receiver is set to "625". 

For operation on the 405-line standard, the 38-15 MHz sound i.f. signal is applied to 
the grid of VI via the capacitor CI 1. Two parallel-tuned circuits, Ll-Cl and L2-C2, 
form two filters (Fl and F2) which are connected in series with one another and in 
shunt with the incoming signal. Together, they form the "trap combination" you 
read about on page 2.104. 

Fl is tuned to resonate at 38-15 MHz, F2 at 6 MHz. To a 38-15 MHz signal Fl 
therefore offers very high impedance, while to the same signal F2 acts as no more than 
a very low capacitive reactance. Thus the "405" signal is developed across Fl, while 
F2 connects the lower end of Fl to earth through the even lower reactance of the 
AGC decoupling capacitor C13. 

Two similar series-connected, parallel-tuned circuits form the anode load of VI. 
The one nearest the anode, L4-C4 (with C4 actually the stray capacitance of the L4 
circuit) is tuned to accept the "405" signal. It also, in combination with L6-C6, forms 
a coupled circuit passing a bandwidth of some 600 kHz distributed about a centre fre- 
quency of 38-15 MHz (L4-C4 having a resonant frequency of about 37-85 MHz and 
L6-C6 a resonant frequency of about 38-45 MHz). 



2.106 [§I2 

IF Amplifier B — Operation (continued) 

L3-C3 and L5-C5 (whose resonant frequencies are 5-9 MHz and 6-1 MHz respec- 
tively) form a similar bandpass coupled circuit for the "625" signal, having a 200 kHz 
bandwidth centred on 6 MHz. At 38-15 MHz, however, they behave as very low 
capacitive reactances, and so offer low impedance to the "405" signal. 

The effective anode load of VI is therefore the high impedance of L4-C4. The 
amplified "405" signal is developed across this impedance, and is fed to the grid of V2 
by the transformer action of L6-C6 and through the low series-connected reactance of 
L5-C5. 

The switch SI is operated by the Standard Selection control. In the "405" position, 
it open-circuits resistor R9 so that the screen grid of V2 is connected to the HT line 
through R8. Connected in this way, V2 behaves as a straightforward amplifier (just 
as does VI, whose screen grid is connected to HT through R2). 

C14 and CI 8 are decoupling capacitors for the screen grids of VI and V2 respec- 
tively. R7 and C17 come into play on "625" only. 

The anode load of V2 consists of tuned transformers working exactly as did the 
anode loads of VI, save that the secondaries of the two transformers are differently 
connected. L8-C8 and L10-C10 form the "405" tuned transformer, with its secon- 
dary taken to the AM sound detector. Conversely, L7-C7 and L9-C9 resonate to the 
6-MHz frequency of "625" operation, with their output being taken to the FM sound 
detector with the aid of a tertiary winding, Lll. 

R3-C15 and R10-C19 are the normal decoupling components for the HT supply. 

AGC voltage is derived from the 405-line sound detector only, not being needed for 
the frequency-modulated 625-line signal. It is applied to VI grid via R14, which is 
decoupled by CI 3. 

The resistors R5 and R12 in the cathode circuits of VI and V2 respectively operate 
as undecoupled negative-feedback resistors serving to offset changes in the input capa- 
citances of the two valves. Such changes are often caused by variations in their 
respective gains, either from AGC or by reason of valve ageing. 

The other components in the cathode circuits of the two valves (R6-C16 and R13- 
C20) are there to provide normal cathode bias. 

When the receiver is set for 625-line operation, all those tuned circuits in the grid and 
anode circuits of both valves which are tuned to resonate at 38-15 MHz behave as low- 
value inductive reactances when signals of 6 MHz frequency are applied to them. The 
6 MHz tuned circuits, on the other hand, now offer high Z to the signal and function as 
the grid and anode loads respectively. 

The intercarrier input is applied to the amplifier circuit through the capacitor CI 2, 
and is developed across the resonant circuit F2. The resistor Rl connected across this 
circuit provides the damping needed to give the required bandwidth. Similar damp- 
ing resistors (R4 and Rl 1) are connected across the "625" anode loads of VI and V2. 

The main point of interest in the circuit is that V2 is now connected as an "over- 
driven amplifier" so that it acts effectively as an amplitude limiter to the "625" signal. 
What happens is that, when SI is moved to the "625" position, R9 is brought into cir- 
cuit so as to form, with R8, a potential divider across the HT supply. The positive 
voltage applied to the screen of V2 is thus reduced, and this in turn reduces the grid 
base of the valve. 



§12] 



2.107 



IF Amplifier B — Operation (continued) 

You will remember from Basic Electronics that the "grid base" of a valve is the 
range of grid voltage which extends from that at which anode current is cut off to that 
at which grid current starts to flow. A reduced grid base for V2 therefore means that 
even a small variation in the amplitude of the signal applied to its grid will cause the 
valve to overload and cut off. 

The amplified output signal will thus be limited to a nearly constant amplitude, a 
condition which you know to be desirable for the proper operation of the detector cir- 
cuit in the next stage. (This is true even though the type of detector actually used in 
that stage is one that is relatively insensitive to amplitude variations, as you will find is 
the case.) 

The limiting action of V2 is reinforced by the "leaky-grid" bias afforded by the two 
components C17 and R7. When the application of a positive signal to the control 
grid of V2 causes grid current to flow in the valve, electrons flow on to the upper plate 
of C 1 7 through L6 and L5, and the capacitor charges negatively. When the valve cuts 
off, the charge on CI 7 leaks away across R7; but this resistor is of such a value that 
leak-away occurs only slowly. 

Thus a nearly steady negative bias is maintained on the control grid of V2, which 
also helps to limit the amplitude of the amplified "625" signal. 

REVIEW of the Intermediate Frequency Amplifier 

The principal function of the IF Amplifier in the British Dual-Standard receiver is to 
raise the amplitudes of the sound and vision intermediate-frequency signals leaving the 
two tuners (VHF and UHF) until they become large enough to activate the Sound and 
Video Detector stages. 

The ways in which the sound and vision signals are processed in the D7 Amplifier stage 
are completely different in the VHF and UHF systems. 




i 0fl*» tf.*MHJHC*nO* STACC 
J» t*t Brltltk »uri.Stm4u4 ttthtr 



2.108 [§12 

REVIEW of the IF Amplifier (continued) 

On 405-line operation, the sound and vision signals are separated as soon as they leave 
the VHF tuner. The vision signal then receives its i.f. amplification in what has been 
(unofficially) christened in this Section IF Amplifier A, while the sound signal receives its 
own degree of i.f. amplification in the completely separate IF Amplifier B. 

This is called the split-sound method of signal processing. 



On 625-line operation, the sound and vision signals coming from the UHF tuner are 
amplified together in IF Amplifier A, and are not separated until they reach the Video 
Detector. They are there caused to beat together to produce a 6 MHz intercarrier sig- 
nal of constant amplitude, which carries the signal content of the sound signal in the form 
of small frequency deviations about the 6 MHz beat frequency. 

The UHF vision signal continues on its way through the Video Detector, but the inter- 
carrier bearing the UHF sound signal is taken back to the input of IF Amplifier B. It 
then receives its due measure of i.f . amplification in DJ Amplifier B before travelling on to 
the Sound Detector stage. 

This is called the intercarrier method of signal processing. 



Frequency Response — IF Amplifier A. 
The frequency response curve of IF Amplifier 
A needs to be re-shaped, and its width in- 
creased, every time the Dual-Standard re- 
ceiver is switched from VHF to UHF. The 
reasons are: (a) the different transmission 
band-widths of the two systems, (p) the fact 
that the two carriers are modulated with op- 
posite polarity, and (c) because in the 405- 
line system it is the upper sideband which is 
partially suppressed, whereas in the 625-line 
system it is the lower. 











N 






/;' 


405- Line 
Response 


\ 


I 

4db 

1 1 

\ 625- Line 
\ Response 



Frequency (MHi) 

m mwiHcv Ktsroitse * • &/** 



The other important functions of the IF Amplifier are to improve the overall selectivity 
of the receiver, and to prevent the wanted sound and vision i.f. signals from being im- 
paired by either second-channel or adjacent-channel interference. 



§12] 



2.109 



REVIEW of the IF Amplifier (continued) 

Rejector circuits, or traps, are used (a) to change the shape of the frequency response 
curve of IF Amplifier A, (A) either to shunt unwanted signals away from the path of the 
desired signals throughout the vision i.f. amplifier stage or to block them off it, and (c) to 
separate the 405-line sound and vision signals from one another at the input to IF Ampli- 
fier B. 

Many kinds of trap exist but all are made up of varying combinations of L, C and R. 
The types most commonly used in the TV Receiver are these: 



WmjuJ 



rr- & 



„ • Output i 



-CD^ 



A. parallel resonant rejector connected in 
series with the signal path BLOCKS an 
unwanted signal when the resonant frequency 
of the rejector is made equal to the fre- 
quency of the intruding signal. 



A series resonant rejector connected in 
parallel with the signal SHUNTS an un- 
wanted signal away from the signal path 
when the resonant frequency of the rejector 
is made equal to the frequency of the un- 
wanted signal. 

This latter technique is also known as 
signal absorption. 







t 




Output 




Input 




1 




Output ^ 


n 


Input 




1 




Output t 



2.110 



§13: THE SOUND SIGNALS 



It will be convenient now to take the two sound signals (VHF and UHF) from their 
common IF Amplifier right through to the loudspeaker, even though you have not yet 
seen how the 6 MHz intercarrier carrying the UHF sound signal is generated in the 
Video Detector. All the other circuitry involved will be familiar to you from your 
work in Basic Electronics; and it will simplify future explanations if there is no need to 
worry much more about what happens to the sound signals in the British Dual- 
Standard Receiver. 

Remember what were the two output signals from IF Amplifier B : 

405-line (VHF): An i.f. carrier of 38-15 MHz, amplitude-modulated by the 405-line 

sound signal. 
625-line (UHF) : An i.f. carrier of 6 MHz, frequency-modulated by the 625-line sound 

signal. 

These two frequencies are so widely separated that there is no great difficulty in keeping 
each signal out of the detector circuits designed for the other. 

The circuit diagram on the page opposite is that of the sound detector stage of a 
typical receiver capable of operating on both the 405-line and the 625-line systems. 
Note that all that part of the circuit lying to the left of the heavy vertical dotted line has 
already appeared as the right-hand portion of the circuit diagram on page 2. 1 03 . This 
will explain the apparent absence of inputs in the circuitry shown. 

Lighter dotted lines surround the circuits handling the UHF and the VHF signals 
respectively. Physically, the greater part of each detector circuit is located within the 
screening can containing the tuned transformer to which it is related in the anode cir- 
cuit of the second i.f. amplifier valve in IF Amplifier B. (These tuned transformers are 
labelled L7-L9 for the UHF signal, and L8-L10 for the VHF signal, in the illustration 
on page 2.103.) 

Detecting the VHF Sound Signal 

Take first the detection of the VHF sound signal. You will recall from page 5.29 of 
Basic Electronics that the essential stages in the process of detecting the audio- 
frequency content of an AM signal are: (a) the rectification of the i.f. signal into pul- 
sating d.c. ; (b) the filtering out of the i.f. component of the rectified signal, leaving only 
its audio-frequency component ; (c) the amplification of this a.f. component to a value 
suitable for (d) its reproduction as audible sound in the loudspeaker. 

In the circuit diagram opposite, the 405-line detector (D3) is a normal half-wave 
diode detector of the semiconductor type, so connected as to pass only the negative- 
going half-cycles of the signal. These cycles, in the form of negative-going pulsating 
d.c, then encounter the low-pass filter formed by CI, C2 and Rl whose purpose is to 
remove the residual i.f. component from the signal. The filter passes to earth signals 
of all frequencies higher than those detectable by the human ear (the audio-frequency 
range for most people is of the order of 1 5 kHz). Thus it is only the negative-going 
half-cycles of a wave having the comparatively long wavelength characteristic of the 
audio-frequency range which can get past the filter to be developed across R7. 

The resistor R2 is connected into the circuit so as to form, with R7, a potential 
divider across the output of the detector. This reduces the loading placed on the 
detector by subsequent circuitry — especially by the interference-limiter circuit formed 
by R8, D4, R9 and C8. 



§"3] 

The Sound Detector 



2.111 




CO 



C9 



CO 



I 



s 



0) 3 " 
J= O ■- 



CO 

CO 



2.112 [§13 

Detecting the VHF Sound Signal (continued) 

Across R7 there are two paths for the signal to take. One is through R14, a rela- 
tively large (1 M) resistor, to be developed across the 0-1 mfd capacitor shown as CI 3 
on the illustrations on pages 2. 1 03 and 2.111. This is an AGC voltage whose purpose 
is to ensure that the gain of VI (seepage 2.103) remains reasonably constant whenever 
the amplitude of the received signal varies. What happens is that this AGC is caused 
to follow, by means which will be explained in a later Section, not the variations of the 
individual half-cycles of the audio signal, but the mean value of these variations through- 
out the period of transmission. As the strength of the received signal (e.g.) increases, so 
does the mean value of the (negative) AGC voltage. More negative bias is applied to 
the grid of VI, and the gain of the valve is reduced. 

Similarly, if the received signal should start to fade, the mean value of the negative 
AGC voltage would be reduced, less bias would be applied to the grid of VI, and the 
gain of the valve would increase to offset the falling value of the signal. 

The other direction which the audio signal across R7 can take is through C6 to a 
second semiconductor diode, D4. 

The purpose of D4 is to act in conjunction with R8, R9 and C8 as an interference- 
limiter circuit, smoothing out (and if possible eliminating) the effect on the signal of 
unwanted noise. Noise pulses derive from such sources as motor-car ignition sys- 
tems, electric drills and the like. They are typically of short duration and very steep- 
sided — in other words, they contain very high-frequency components in their make-up. 
You will remember that great care has been taken in previous circuits (especially by 
giving IF Amplifier B a much wider bandwidth than it theoretically needs) to preserve 
the shape of these intruding noise pulses without distortion, so that they can be satis- 
factorily eliminated at a later stage. That is what is about to happen to them now ! 

When no signal from C6 is present, D4 is forward-biased by the positive voltage 
applied to it from HT (+) through R8. The internal resistance of a forward-biased 
diode is no more than a few hundred ohms, so that the diode in this condition rep- 
resents only a low resistance between the two resistors R8 and R9. These two thus 
form a potential divider, and the capacitor C8 charges to a voltage determined by their 
relative values. The actual value of this charging voltage is not of great importance, 
but is approximately equal to the value of the expression R9/(R8 + R9) x V HT - 

Now assume that an audio signal, varying in amplitude according to the modulation 
applied to it at the transmitter but carrying no attendant noise, reaches D4 through 
C6. The frequency of the variations in this signal will be quite low— well within the 
audio-frequency range of 1 5 kHz. Within this audio range, the voltage developed by 
C8 is able to follow accurately the voltage fluctuations of the incoming signal, but can- 
not do so for any signal whose frequency varies any faster. This is achieved by careful 
selection of the values of R2, R7, R8, R9 and C8, so that the time constant and the 
charging resistance of C8 can be precisely adjusted. 

With C8 able to follow the fluctuations of the signal when it has no appreciable con- 
tent of noise, D4 remains forward-biased. You know that, when a diode is conduct- 
ing normally, the polarities of both its terminals move "up and down" in unison, so 
the diode presents itself to the signal as a closed switch. The audio signal is therefore 
satisfactorily developed across C8, and passes via the Standard Selection switch S2 and 
the coupling capacitor C9 to the Audio Amplifier. 



§13] 



2.113 



How the YHF/ Interference Limiter Works 

Now suppose that the audio signal reaching D4 has impressed upon it sudden noise 
pulses of large amplitude. Pulses of this shape and duration are over and done with 
much too fast for the charge on C8 to be able to follow them. So while the left-hand 
terminal of D4 will be driven rapidly negative by the large negative noise signals 
arriving from the detector diode D3, its right-hand terminal will be unable to follow 
and will therefore be less negative than its fellow for the duration of the pulse. With 
the right-hand terminal effectively positive with respect to the left-hand one, the diode 
becomes reverse-biased. In this condition it will not conduct, and the noise pulse is 
blocked. 

In practice, of course, blocking is never perfect; and the illustration below gives an 
idea of the approximate extent to which the effect of noise pulses is reduced by the 
action of a typical interference limiter. 



Negative 
Output 
Voltage 
from D3 




One Cycle of Audio Signal 
Output from D3 



D.C. Output of D3 



NOISE PULSES 



Positive 
Output 
Voltage 
from D4 




Improved Shape of Output 
from Limiter Diode D4 



Voltage Drop across R9-C8 



now the INTERFERENCE LIMITER reduces Noise 



Take care, by the way, to distinguish the interference limiter circuit represented by 
D4-R8-R9-C8 from the simple type of limiter circuit you learnt about in Basic Elec- 
tronics, whose function was to limit the amplitude variations of an FM signal so as to 
allow its frequency variations to be satisfactorily converted into an audio signal. 



2.114 [§13 

Detecting the UHF Signal 

When the receiver is set to operate on 625 lines, the output from the VHF detector, 
including its AGC line, is short-circuited to earth by the switching of SI to its "625" 
setting; while the similar switching of S2 disconnects the interference limiter and en- 
sures that the audio output signal is taken only from the UHF sound detector circuitry. 

The detector used is of the ratio detector type which you learnt about on pages 6.41 
and 6.42 of Basic Electronics. The principal advantage of this type of detector is that 
it is relatively insensitive to variations in the amplitude of the incoming signal. There 
is therefore no need for an additional amplitude limiter in the detector circuitry (see 
foot of last page) to remove the amplitude variations which normally need to be 
eliminated from an FM signal before it can be satisfactorily detected. In any case, as 
you already know, V2 is connected as an overdriven amplifier when the receiver is set 
for 625-line operation, and so in itself acts as a limiter to reduce the sharp variations of 
amplitude which are present on the 6 MHz intercarrier. Without this limiting action 
of V2, the picture reproduced on the screen would almost certainly be impaired by 
intercarrier buzz, for not even the ratio type of detector can give perfect rejection of 
amplitude variations. 

Glance back now to the circuit diagram on page 2.111 and you will see that the 
detector consists of the two oppositely-connected semiconductor diodes Dl and D2, 
with their associated circuitry C3-C4-R3-R4-R5-R6-C5. Compare this layout with 
that shown on Basic Electronics page 6.41 (which had valve diodes instead of semi- 
conductor ones), and you will have no difficulty in seeing how the detector works. C5, 
of course, is the capacitor across which it is essential to maintain a constant voltage. 

The audio-frequency output from the detector is taken from the junction of C3 and 
C4 through the combination R10-C7. The values of RIO and C7 are so chosen that 
they form a de-emphasis filter having a time-constant of 50 microseconds. Their 
function is to counteract the 50 jjls pre-emphasis given (as you know) to the signal at 
the transmitter with the object of increasing the gain of the transmitter's audio ampli- 
fier at the higher audio frequencies. (For pre-emphasis see Basic Electronics page 
6.15; for de-emphasis, page 6.45.) 



The Audio Amplifier and Loudspeaker 

When they emerge from the sound detector circuitry, the VHF and UHF signals are 
both of audio frequency. The same circuits can therefore be used to handle them 
from now on. These circuits are shown in the illustration opposite. 

The input to the audio amplifier is derived from the capacitor C9, which is the same 
component as the one similarly labelled in the illustration on page 2.111. 

The heart of the audio amplifier is a single valve of the triode-pentode type, with the 
triode section functioning as a simple voltage amplifier feeding the pentode power 
amplifier. The signal is developed across the anode load (R2) of the triode, and is 
taken to the grid of the pentode through C2 and R5. 

The lower end of the Volume Control VR1 (which the viewer can adjust to his con- 
venience) is taken to earth through the secondary winding of the transformer which 
forms the anode load of the power amplifier. This arrangement provides some degree 
of negative feedback to the grid of the triode, so helping to improve the overall fre- 
quency response of the audio amplifier circuitry. 



§13] 



2.115 



The Audio Amplifier and Loudspeaker (continued) 

A similar function is performed by the undecoupled resistor R3 in the cathode cir- 
cuit of the triode, which also provides negative feedback. 

The loudspeaker itself is generally of the moving-coil type described on pages 2.89 to 
2.96 of Basic Electronics, Part 2, and thus needs no further description here. 



IK AUDIO AMPLIFIER 

km QUIPUT swat 



H.T. (+) 



Loudspeaker 




2.116 



§14: THE VIDEO DETECTOR 



Having been duly amplified by IF Amplifier A, the vision signal is now of sufficient 
size to operate the video detector. This stage generally requires an input signal of a 
few volts for efficient operation. 

The basic purpose of the video detector is to remove the video modulation from the 
vision i.f. carrier so that it may be used (a) to modulate the raster on the picture tube, 
and (b) to synchronise the operation of the line and field scanning generators. You 
will recall that the video signal contains all the detail of the picture signal produced by 
the camera, plus the synchronising and blanking pulses required to trigger the scanning 
circuits of the receiver. 

In many ways, the function of the video detector is similar to that of the sound 
detector whose job, you will remember, was to remove the sound signal modulation 
from the sound i.f. carrier so that it could be used to operate the loudspeaker. Indeed, 
since both these detectors have in common a demodulating role, they are often 
referred to as sound and vision demodulators. 

When the Dual-Standard receiver is switched to "405" and is therefore receiving on 
VHF, its sound and vision circuits are, as you know, separated before they reach the 
detection stage, and demodulation is all the video detector has to do. On "625", 
however, where the intercarrier method is used, the video detector is also required to 
accept both the sound and vision i.f. carriers, and to produce from them a 6 MHz 
difference, or beat-frequency, signal for input to the separate sound detector. The 
two essential tasks of the video detector in an intercarrier type of receiver are therefore: 

O To remove the video signal modulation from the vision i.f. carrier. 

© To produce the 6 MHz beat signal from the sound and vision i.f. carrier. 



You will now see how these requirements are met. 




Video Modulation removed 
from Vision Carrier 



39-5 MHz I.F. Vision Carrier 
Amplitude Modulated 





33-5 MHz I.F. Sound Carrier 
Frequency Modulated 



6 MHz Intercarrier Signal 

(Produced by Heterodyning the I.F 

Sound and Vision Carriers) 



THE FUNCTION OF THE VIDSO MTiCTOR 



§14] 



2.117 



The Video Detector— Basic Circuit 

You know that both the i.f. vision carriers (VHF and UHF) are amplitude-modula- 
ted by the video signal, which means that the video detector circuit must respond to 
changes in signal amplitude. This is achieved by using a simple half-wave diode 
rectifier circuit which differs little from the detector circuit in an ordinary AM radio 
receiver— the main difference between the two lying only in the frequency of the 
modulating signals. 

In the 405-line TV system, the polarity of the video modulation is positive, so that 
the amplitude of the vision carrier becomes greater as the whiteness of the camera 
scene is increased. The converse is true of the 625-line system, in which earner 
amplitude is decreased as the scene is made whiter. A video detector designed to 
operate on a positively modulated signal needs to have its diode connections reversed 
if it is to provide a rectified output signal of the correct polarity after being supplied 
with a negatively modulated signal. Receivers designed for dual standard operation 
achieve this by having the diode connections electrically reversed when the Standard 
Selection control is switched from "405" to "625". 

The diagrams below show, in a simplified way, how the video modulation is removed 
from carriers having opposite modulation polarities. The arrows indicate the direc- 
tion of current flow in the circuit during the half-cycles when the diode D is conducting. 
The rectified output signal (i.e., the video modulation) is the mean voltage developed 
across the load resistor R from the pulses of current charging the load capacitor C 
during these conducting half-cycles. 



(+) 

- - White 




Video Modulation 



I.F. Carrier 



POSITIVE Demodulation (405-Line System) 





Output 



1 1 'TrTc \ / 

E ,11. -u 



White 




.F. Carrier 



Video Modulation 

HBGAWt Demodulation (625-Line System) 



2.118 [§|4 

The Video Detector — Frequency Response 

There are several reasons why the simple detector circuits you have just looked at are 
unsuitable in their present form — one of the principal ones being their inadequate fre- 
quency response. 

You will recall from what you learnt in Part I that the frequency content of the video 
signal ranges from zero ( = d.c.) up to about 5-5 MHz (the width of the unsuppressed 
sideband of the UHF vision carrier). This frequency spectrum contains not only the 
variations of the picture signal itself, but also the unvarying repetition frequencies of 
the line and field blanking and synchronising pulses. It is of the first importance that 
the shape of these pulses — especially the sharp leading edges of the sync pulses — should 
be preserved, for otherwise the synchronisation of the line and field scanning pulses 
will become erratic and the picture reproduced on the screen (particularly of the larger 
sizes of picture tube) will become unacceptably ragged. 

These sharp edges of the sync pulses introduce into the video signal a comparatively 
high-frequency component which, though not as important numerically as the high- 
frequency component of the video signal itself, is a good deal more important opera- 
tionally and must at all costs be preserved. Since the rise-and-fall times of the sync 
pulses are of the order of 0-25 [is (corresponding to a maximum harmonic frequency of 
1 MHz), the video detector must have a frequency response of at least that value if it is 
to be able to reproduce the sync pulses without distortion. 

The simple detector circuits on the last page in fact have a poor frequency response. 

This is because the load impedance (Z) formed by the combination of R and C is 

sharply reduced as signal frequency rises. You learnt in Basic Electricity, Part 4, that 

R x X 
the impedance of an RC combination is given by the equation Z = 



VR 2 + X c 2 
where X c is the capacitive reactance, which can be quantified as X c = l/2w/C. 

Work out the elementary algebra of the equation, and you will see that at very low 
signal frequencies the value of X is high and the load is mostly resistive. A good 
signal can therefore be developed across the large R. But as signal frequency rises, 
the value of X c drops to a low value, the load becomes mainly capacitive, and signal 
development is poor. 

At HIGH Signal Frequencies, Frequency Response is POOR 



OUTPUT SIGNAL 





Signal Frequency (f) 

One solution to the problem which is theoretically possible would be to make the 
value of R so small that any variations in the value of X c would become insignificant. 
But a very low value of R would result in an unacceptably small output signal, and 
some other remedy must be sought. 



§14] 



2.119 



The Video Detector— Frequency Response (continued) 

A more hopeful solution is to add something to the circuit which automatically 
increases the value of the load as the value of X c falls. The obvious answer is some 
sort of inductance, since the reactance of an inductor (as you know) increases as the 
frequency rises (* L = 2w/L). So a parallel tuned circuit is formed from a coil and its 
stray capacitances, and is connected in series with the load. 

Now, as signal frequency rises, so the impedance of the parallel circuit increases and 
compensates for the reduction in X c . In other words, the "droop" in the frequency 
response curve shown in the illustration on the last page is "jacked up" by the parallel 
circuit, and the all-important leading edges of the sync pulses are kept sharply defined. 

At HIGH Signal Frequencies, Frequency Response is now IMPROVED 



Original ^, / , 
Response — /~- 

/ 



\ 



Signal Frequency (f) 



Response of 
Parallel -Tuned 
Circuit 




OUTPUT SIGNAL 



The Video Detector Circuit on a Practical Dual-Standard Receiver 

The illustration below shows a video detector circuit representative of the many 
types used in present-day dual-standard TV receivers. 



8 



L12 



D2 



o 
o 

§ Cf 



Ll 



L2 




-o 



4p 



C2 



L.3 




405 \ 



^N 



Video 
Amplifier 



Si 



^ 



V1 



405 
■O 



'Rl 



_62 £o ^*-l 

S2 -±- 



T~C3 



6MHz 



X 



THE DUAL-STANDARD VIPtO P(T£CT0K 




2.120 [§ I4 

The Video Detector Circuit on a Dual-Standard Receiver (continued) 

The detector is the semiconductor diode Dl, which receives its input signal from 
the tertiary winding L12 on the second i.f. transformer in the anode circuit of the 
second i.f. amplifier valve— V2 in the illustration on page 2.99. It is common practice 
to house this diode, its filter capacitor CI and the frequency compensating components 
L1-L2 within the screening can of the i.f. transformer itself— as indicated by the light 
dotted lines surrounding these components in the illustration on the last page. The 
function of the filter capacitor and of the inductors LI and L2 is to ensure that the 
frequency response of the detector does not extend much above 6 MHz. To the i.f. 
frequencies much higher than 6 MHz which are encountered on both line standards the 
circuit acts as a very low shunt impedance — thereby preventing these frequencies from 
reaching the Video Amplifier. 

The two output terminals of the detector circuit, labelled A and B in the illustration, 
are taken to two sections (SI and S2) of the Standard Selection switch. The circuit 
arrangement is such that the signal fed to the Video Amplifier is always positive-going, 
irrespective of the polarity of the signal modulation. 

When the switches are set to "405", terminal B, and with it the negative electrode 
of the diode, is taken to earth through S2. Terminal A, and with it the positive 
electrode of the diode, is taken to the resistor R2 which forms the detector load. The 
video modulation from the 405-line i.f. carrier is developed across this load, and taken 
via SI as a positive-going signal to the grid of the Video Amplifier valve VI . You will 
see that this form of connection resembles the theoretical circuit shown in the top half 
of the illustration on page 2.117. 

When the switches are set to "625", the video modulation is taken from the negative 
electrode of the diode, so achieving the same result as if the connections to the diode 
were physically reversed when the Standard Selection switch is turned. Terminal A 
and the positive electrode of the diode is now taken to earth through S2. Terminal B 
and the negative electrode is taken to the resistor Rl, which is connected in series with 
a 6 MHz parallel-tuned circuit formed by L3 and C3. This combination R1-L3-C3 
forms the detector load across which the video modulation of the 625-line i.f. carrier is 
developed and taken (again as a positive-going signal) through SI to the grid of the 
Video Amplifier valve. This form of connection can be compared with the theoretical 
circuit shown in the lower half of the illustration on page 2.117. 

The 6 MHz tuned circuit L3-C3 offers a very high impedance to signals of that fre- 
quency. There will therefore be strongly developed across it the 6 MHz intercarrier 
signal produced by the beating together (heterodyning) of the sound and vision 
carriers (39-5 and 33-5 MHz respectively). The intercarrier signal is taken through the 
capacitor C2 to IF Amplifier B, where you have already seen how its audio modulation 
is removed. 

To signals whose frequencies are significantly different from 6 MHz, the impedance 
of the tuned circuit L3-C3 is very low; and this low impedance, forming a potential 
divider circuit with Rl, ensures that negligible signal voltages other than the wanted 
intercarrier signal are developed across it and fed to IF Amplifier B. 

Note especially in the illustration the ways in which the 405-line and the 625-line 
detector loads are shorted to earth, both through S2, when the Standard Selection 
switch is set to "625" and "405" respectively. This arrangement ensures that only 
wanted signals are applied to the Video Amplifier, and (on the 625-line standard) only 
the wanted 6 MHz intercarrier signal is sent on its way to IF Amplifier B. 



§14] 



2.121 



The Video Detector Circuit on a Dual Standard Receiver (continued) 

One further point. Since the 625-line sound and vision i.f. carriers are made to beat 
together to produce a 6 MHz beat-frequency intercarrier signal, why (you may ask) 
is not a 3-5 MHz beat-frequency signal produced from the heterodyning of the 405-line 
sound and vision i.f. carriers? The answer is, of course, that the 405-line sound i.f. 
carrier does not pass through IF Amplifier A at all, and that numerous precautions in 
the way of tuned rejector circuits are taken in IF Amplifier A to keep it out. 

Despite these precautions, however, it is still possible for some sound i.f. signal to 
reach the video detector, particularly in strong signal areas; and where this happens, a 
3-5 MHz beat-frequency signal is in fact produced. If this were allowed to reach the 
Video Amplifier, it would appear on the picture tube as a pattern of interference in the 
form of dots. 

To eliminate this danger, some video detector circuits incorporate a 3-5 MHz 
rejector circuit connected in series with the output; but a more usual method is to con- 
nect a parallel-tuned rejector circuit of 3-5 MHz frequency in series with the cathode • 
lead of the Video Amplifier valve. This you will see in the next Section. 



REVIEW of the Video Detector Circuit 

The function of the Video Detector is to remove the video-signal modulation from the 
i.f. vision carrier, and to apply it to the video amplifier. In both 405-line and 625-line 
systems, the vision carrier is amplitude-modulated by the video signal. 

The British Dual-Standard receiver, when it is switched to 625 lines, also requires the 
video detector to produce a 6 MHz intercarrier signal from the heterodyning of the sound 
and vision carriers, whose frequencies differ by this amount. This intercarrier signal has 
impressed on it the frequency modulation of the sound signal, which it takes to the sound 
detector in another part of the receiver. 

The Video Detector usually consists of a single semiconductor diode whose connections 
can be electrically reversed so that it can be used on both the VHF (405-line) and UHF 
(625-line) standards. 




Video Modulation 



MSITtVe Demodulation (405- Line System) 




Video Modulation 



HeWTIVf Demodutatkwi (625-Line System) 



2.122 



§15: THE VIDEO AMPLIFIER 



The video signal has now been separated from the vision carrier, but its amplitude is 
still far below that required to modulate the picture tube and so build up the picture on 
the screen. The actual amplitude of the signal at the video detector will depend partly 
on the strength of the received signal and partly on the degree of overall amplification 
given by the IF Amplifier; but it is seldom more than a volt or two, whereas some 
picture tubes require a modulating signal of as much as 70 V to produce a peak-white 
picture. 

Clearly, then, considerable amplification is required, which it is the job of the video 
amplifier to provide. 

One of the difficulties in the design of a video amplifierfor a Dual-Standard Receiver 
arises from the need to maintain unimpaired the shape of the video signal over the 
extremely wide frequency band-width of the signal (3-5 MHz in the 405-line system and 
5-5 MHz in the 625-line system). Unless this shape is preserved right up to the point 
at which it is applied to the picture tube, the quality of the reproduced picture will 
suffer and its synchronisation will be impaired. 

Another requirement the video detector must fulfil is that the d.c. level produced by 
the video detector (i.e., the average value of the video signal) should also be passed by 
the amplifier and amplified along with the video signal. This d.c. level, as you know, 
represents the average brightness level of the scene (its background illumination). It 
must therefore be applied to the picture tube along with picture-signal content of the 
video signal if the true tonal composition of the scene is to be preserved. 

Since a d.c. level represents a frequency of zero Hz, the frequency response deman- 
ded of the video amplifier should ideally extend from zero up to 5-5 MHz (625-line 
system). 



Ide 

OF * OMt-STANOARO RECEIVER 



Amplifier Gain 
{Arbitrary Scale) 



•> 


Upper-Frequency Limit of 
^ Average Hi-Fi Amplifier 


^ 




c 




12 3 4 

Frequency (MHz) 


1 "I 

5 


1 
6 



As any hi-fi enthusiast will tell you, this is a formidable frequency response to ask 
for— much greater than that demanded of the most expensive amplifier used for the 
reproduction of music, which seldom exceeds 100 kHz and is usually much less. 



§15] 



2.123 



The Video Amplifier (continued) 

Most modern TV receivers employ a single stage for the video amplification. This 
automatically introduces one inversion (180°) of the video signal— whereas, of course, 
a two-stage amplifier would introduce two inversions of the signal (360°) and so pro- 
duce an output signal of the same polarity as the input. 

It is usual to feed the amplified, and inverted, video signal to the cathode of the 
picture tube — which means that maximum brilliance of the reproduced image will 
occur when the amplitude of the video signal is at its minimum value. (A reduction in 
voltage on the cathode of the picture tube produces the same effect as does an increase 
in the voltage on the control grid, the intensity of the electron beam being thereby 
increased.) 

It would, of course, be perfectly possible to reverse the connections to the video 
detector diode (and therefore the polarity of the separated video signal) and to feed 
the inverted output signal from the video amplifier to the grid of the picture tube, 
instead of to its cathode. This method, known as grid modulation, is seldom used, 
however, since under certain operating conditions, particularly when switching ON 
and OFF, it is possible to cause damage to the picture tube by excessive beam current, 
and extra circuit components are necessary to prevent this happening. 

Since the 405- and 625-line systems employ opposite modulation polarities, the 
video detector diodes in the two circuits are, as you know, connected in opposite 
directions. The video signals fed to the video amplifiers are therefore of the same 
relative polarities in both systems. Thus peak-white level in the video signal pro- 
duced by the 625-line system detector is represented by the maximum positive-going 
excursion of the waveform above the sync level (even though the whole of the wave- 
form lies below zero volts); while in the 405-line system detector, peak- white is rep- 
resented by the maximum positive excursion of the waveform above the zero- volt level. 

After inversion by the video amplifier, however, peak-white is represented in both 
systems by the minimum excursion of the video signal, which thus produces maximum 
scanning beam intensity and maximum brilliance of the picture. 



405-LiNE SYSTEM 




625- LINE SYSTEM 



Picture Tube 



mOWUMHG the PICTURE TUBE 



2.124 [§ | 5 

Frequency Compensation 

The video amplifier itself usually consists of a single pentode valve having a resistive 
anode load (R t ). Within limits, the larger the value of this load, the greater will be the 
overall gain (A) of the stage. To put it mathematically, A = G m x R L , where G m is 
the mutual conductance of the valve expressed in mA/V. 

In practice, however (as was also true of the video detector), the actual value of the 
anode load is not just R L , but rather the combination of R L and the reactance of sundry 
stray capacitances in parallel with it. These stray capacitances are the output capa- 
citance of the valve itself, plus the capacitances existing between individual wires and 
components, and plus the input capacitance of the picture tube and of the sync 
separator stage. 

If, correctly, the load is now expressed as an impedance Z L , the gain of the pentode 
becomes A = G m x Z L , with 

„ _ Rr.x X,.. 

^L — 



Vr l 2 + x c 2 

where X c (the reactance of the stray capacity in parallel with R L ) = \\2-nfC. 

You will see that the magnitude of Z L depends on the value of X c , which in turn is 
inversely proportional to the frequency of the signal. It follows that Z L , and therefore 
A, will decrease as the frequency rises. The extent of the variation in Z L depends, of 
course, on the value of the stray capacity, which is typically of the order of 25 pf. 

Take a simple example. Assume a value of 10 k for R L , and say the frequency of the 
applied signal is as low as 50 Hz. The value of X c at this frequency is 1/(2 x 3-14 x 
50 x 25 x 10" 12 ), or about 130 megohms. The effect of 130 M in parallel with 10 k 
is, of course, negligible and the presence of X c may therefore be ignored. At this 
frequency, then, the value of Z L is effectively R L , and A = G m x 10*. 

But consider what happens when the frequency of the applied signal is increased to 
5 MHz— as it very well may be in the 625-line system. The value of X c is now only 
1/(2 x 3-14 x 5 x 10 8 x 25 x 10~ 12 ), or about 1,300 ohms— some 10,000 times less 
than it was at 50 Hz. With the value of X c now comparable with that of R L , it be- 
comes important in determining the value of Z L . The equation is now: 

7 10,000 x 1,300 . n . , 

7 — - — 130 ohms, 



V10,000 2 + 1,300 2 
so that A = G m x 130. 

With the value of the anode load impedance varying from 10,000 ohms at 50 Hz to 
130 ohms at 5 MHz (a gain change of nearly 80 : 1), some form of frequency compen- 
sation is clearly required to stabilize the value of the anode load as the frequency alters. 

Alternatively, the value of R L can be made so small compared with the lowest value 
of X c that the effect of changes in X c become insignificant. In practice, both methods 
are employed. R L is commonly reduced to a value equal to about twice the value of 
X c at the highest frequency in the bandwidth, and inductive compensation is achieved 
by placing a small inductor in series with R L . 



§15] 



2.125 



Frequency Compensation (continued) 

Various arrangements have been tried for stabilising the frequency response of the 
video amplifier by means of an inductor coil connected as part of the effective load 
(Z L ) of the valve. Some receivers employ two coils connected in a series/shunt 
arrangement, while others favour a single coil with a cathode-follower feed to the 
picture tube. The simplest arrangement, however, consists of a single coil connected 
in series with the anode load (R L ), the presence of this coil being enough to cause the 
effective value of R L to increase with frequency and so compensate foE the reduction in 
Z L caused as X c falls. (You will recall that a similar arrangement was used to boost 
the frequency response of the cascode amplifier in the tuner stage. The coil was there 
referred to as a "peaking coil", and the same name is given to the coil used in the video 
amplifier.) The value of the inductance of the peaking coil is carefully chosen so that 
it will form a parallel resonant circuit with the stray capacitances of the circuit at a 
frequency slightly higher than the maximum frequency it is desired to handle. 

A further advantage of the peaking coil is that the presence of R L in series with it 
introduces considerable damping of the tuned circuit, and so makes its frequency 
response broader. 

The illustration below shows three frequency response curves — G m R L , G m (2R L ) and 
G m (3R L }— in which the value of the anode resistance is progressively doubled and then 
trebled, but in only one of which, GJ2R L ), is a compensating peaking coil used. 
(Note that it is customary, when assessing the effective bandwidth of an amplifier, to 
cite the frequency at which the gain of the amplifier has fallen by a specified amount. 
This amount is usually (-)3 dB, at which the gain is 70% of maximum.) 

How FREQUENCY COMPENSATION tmpfWeS GAIN 



Gm (3Rl) 




Over-compensated 



Frequency -3db Gm Rl 

Other things being equal, the gain of an amplifier, as you know, becomes greater the 
larger the value of its R L . G m R L , therefore, though its frequency response shows com- 
paratively little drop at the (-)3 dB point, gives only poor gain. G m (3R L ), equally 
uncompensated, shows excellent gain — but one which drops off so sharply as frequency 
rises that the crucial (-)3 dB point is reached with a considerably smaller frequency 
excursion. 

Note, by contrast, the effect which a peaking coil of the proper value has on the 
G m (2R L ) curve, permitting the use of a load resistor of twice the uncompensated value 
yet with a frequency response at least as good. 

The two dotted lines in the illustration show the results of connecting a peaking coil 
of the wrong value. Typically, such coils are not physically larger than is a quarter- 
watt resistor. They have inductance values of between 100 and 200 microhenries. 



2.126 



[§15 



Ringing and Overshoot 

If the current flowing in a peaking coil (or in any other tuned circuit for that matter) 
is suddenly cut off, the energy stored in the capacitor and coil at the time of cut-off will 
cause the circuit to oscillate for a few cycles before it settles down to the zero-current 
condition. This phenomenon is known as "ringing". The same thing happens when 
current is suddenly applied to a tuned circuit, the circuit going through a short oscil- 
latory phase before settling down. 

In some applications (radar and oscilloscope calibration circuits are examples) the 
ringing of a tuned circuit may be of value. In the TV video amplifier, it is most un- 
desirable. This is because it causes what is known as "overshoot" on the video signal 
waveform fed to the picture tube, and this causes a peculiar shimmering effect on the 
edges of the component parts of the picture appearing on the screen. What happens is 
that normally white areas of the picture are very rapidly succeeded by identically- 
shaped black areas, and then by more white areas. A similar alternation of white and 
black areas follow each original black area which appears. 

Overshoot can also affect synchronising efficiency, particularly of the line timebase 
circuit, causing the timebase to be prematurely triggered and so giving parts of the 
picture ragged edges. 

Ringing can be controlled to an acceptable degree by connecting a resistance of care- 
fully chosen value in parallel with the peaking coil. Controlling the ringing of a tuned 
circuit in this way is known as "damping". The choice of a damping resistor of the 
correct value is crucial. The illustration shows what will happen if a mistake is made. 



lH 



Ar — 



Ideal 


Undamped 


Correct 


Excessive 


Waveform 


Waveform 


Damping 


Damping 




Bad Overshoot and 


Negligible 


No Overshoot but 




Too-high Frequency 


Overshoot 


Impaired Frequency 




Response 




Response 



THE EFFECTS OF Damping ON A STEEP-SIDED WAVEFORM 

Thus under-damping leads to overshoot, and excessive damping reduces the fre- 
quency response of the peaking coil — which defeats the whole object of putting the coil 
there in the first place. 

It is common practice to wind a peaking coil over the top of a resis- 
tor whose value is just right to stop the coil ringing. The ends of the 
coil are connected to the leads of the resistor, the body of which 
serves as a support, or "former", for the coil. The outcome is a 
compact and easily-handled peaking circuit which is already pre- 
cisely damped before use. 



Peaking 
Coil 




§15] 



2.127 



Biasing Levels in the Video Amplifier 

If the video amplifier is to produce a faithful amplification of the video signal 
applied to its grid, the grid bias voltage must be so adjusted that the full amplitude of 
the signal coming from the video detector can be accommodated within the grid base 
of the valve. In other words, the extremities of the grid waveform must not extend 
beyond the cut-off and grid-current limits of the grid base. (This sort of requirement, 
of course, applies to any amplifying system in which fidelity of reproduction is im- 
portant and is in no way peculiar to the VA.) 

If the signal applied to the video amplifier grid were symmetrical in shape, having 
equal positive and negative areas like a sine wave, it would be sufficient to set the grid 
bias level in the centre of the linear region of the grid base. The positive and negative 
excursions of the input signal would then extend by equal amounts above and below 
the grid bias value, and the whole of any signal of normal amplitude would be con- 
tained within the grid base. 

But the shape of the video signal waveform is not symmetrical; and its mean value, 
or d.c. level, being representative of the mean brightness level of the transmitted scene, 
can never be predicted. In some cases, the picture signal may not even be present at 
all in the video signal, even though the sync pulses are there— as sometimes happens 
just before the start of a programme when a blank screen appears on the picture tube. 
All video waveforms are basically unidirectional in nature, i.e., their polarity 
referred to zero is either wholly positive or wholly negative. Bias voltage levels for the 
video amplifier must therefore be referred to this zero voltage reference, and not to 
some arbitrary level which may not always be present. 

You know that, on both the 405 and the 625-line systems, the signal coming from 
the video detector is of positive-going polarity, but that the relative potentials of the 
two waveforms are different. In the video waveform of the 405-line system the sync 
level is only slightly above chassis potential, the complete waveform being positive with 
respect to chassis. In the 625-line system, the near-zero condition of the video wave- 
form is that corresponding to peak white, the complete waveform being negative with 
respect to chassis. It is because of these differences that a video amplifier designed to 
operate from a 405-line video waveform needs a different bias level from one required 
to operate from a 625-line signal. 

In the British Dual Standard Receiver, the grid bias level of the video amplifier is 
one of the many things which are changed when the Standard Selection switch is 
moved from one of its positions to the other. 



la 



The entire 
Video Waveform 
roust h$ 




405- Line Video 
Waveform (above Zero) 




Both Video 
Waveforms are 
Positive-Going 

but their relative 



-ov 




(Chassis) 



625-Line Video 
Waveform (below Zero) 



2.128 



BIS 



Biasing for the 405-line System 

The lowest value of the 405-line video waveform is that which corresponds to sync 
level, which is slightly above chassis potential. The remainder of the waveform- 
black level and picture-signal content — all represent increases above this level. To 
handle this type of signal, the grid bias voltage of a video amplifier should be set to a 
level sufficiently negative to ensure adequate grid base accommodation for the full 
positive excursions. 

Consider, for instance, a video signal waveform having a maximum overall ampli- 
tude of 3 volts, and a sync level corresponding to zero potential. To accommodate 
such a signal at its grid, the grid bias of the VA must be set to a level of at least 
( — )3 V. The video signal will then "sit" on this value of bias, and all its positive 
excursions will just be accommodated within the grid base of the valve, the most posi- 
tive excursions of the video signal just reaching the zero voltage region of the I J V g 
curve. 

But if, as is the case in the illustration below, the grid base of the valve is consider- 
ably more than 3 V, then a larger value of negative bias is possible. A grid bias of 
( — )4 V, for example, would not only ensure that the positive peaks of the video signal 
were well clear of the zero- volt region (the point at which grid current starts to flow in 
the valve and introduces distortion of the signal), but would also reduce the standing 
current in the valve. (This standing current is the value of anode current which flows 
in the absence of a video signal. In the case shown, it is 10 mA.) 



la 
(mA) 




Standing Current 
for (-)4V Bias 



3V Video 



GRID BIAS for the 



40S-IME 



It is important that the biasing point be kept well clear of the curved portion of the 
IjV g curve that lies in the region approaching 7 a cut-off. If the tips of the sync 
pulses extend into this region, amplification of the pulses will be very nonlinear. 
Distortion of the sync pulses and consequent impairment of synchronising efficiency 
will result. 

So the bias level should be set to a value which (a) ensures that the full amplitude of 
the video signal is embraced by the linear portion of the I J V e curve, and (b) maintains 
the lowest possible standing anode current. 



§15] 



2.129 



Biasing for the 625-line System 

In the 625-line system, the polarity of the video waveform coming from the video 
detector circuit is entirely negative with respect to chassis, and maximum negative 
amplitude occurs at the tips of the sync pulses (i.e., at sync level). The minimum level 
reached by the signal is that representing peak white, which is slightly below zero. 

For the VA to be able to handle the 625-line video signal, its bias level must be fairly 
close to the zero V g point of the valve so that the full negative excursions of the signal 
can be accommodated within the grid base. Consider, for example, a video signal of 
3 V amplitude reaching the grid of a pentode having a grid base of about 6 V. A bias 
level of about (-)IV would be required to accommodate the 3 V excursions of the 
signal, which would extend from (— )1 V to (-)4 V. 

In the absence of a video signal, the standing current corresponding to a ( - )1 V bias 
level would be about 27-5 mA. This is considerably higher than was the standing 
current in the 405-line system, and is a disadvantage of this type of circuit. 




1-volt Bias ' 
Setting | 



6 
-Vg 



3V Video 
Signal 







"iV«i 



■ 






** 30 ^27-5mA 
*r Standing Current for 
j a (-11-volt Bias 

(mA) 

-20 



-10 






ov 



+Vg 



GRID BIAS ***** 



2.130 



[§I5 



The Dual-Standard Video Amplifier 

It follows from what you have just read that an essential part of an adequate video 
amplifier for a Dual-Standard receiver is some mechanism for changing the bias con- 
ditions on the cathode of the amplifying pentode when the Standard Selection switch is 
rotated from "405" to "625", or vice versa. The illustration below shows one way in 
which this is done. 



+ 200V 



Video Signal 
Output 




-+200V 




-0V 



0V (Chassis) 



The Basic Circuit of the DUAL-STANDARD VMeO Ampftffef 



The pentode is powered from a 200 V HT supply and has a 2-7 k anode load 
frequency-compensated by a 150-microhenry peaking coil. Screen grid voltage is 
derived from the 200 V supply through a 5-6 k resistor, producing a screen grid voltage 
of about 150 V. The 5-6 k resistor is decoupled by a 1-mfd capacitor. 

The cathode circuit of the valve contains two resistors connected in series. One of 
these is quite small in value (only 18 ohms); the other, comparatively large (150 ohms), 
has a 250 mfd decoupling capacitor connected in parallel with it. These two com- 
ponents form a conventional cathode bias circuit. One section of the Standard 
Selection switch (SI) is connected across this cathode bias circuit in such a way that in 
one of its two positions it "shorts" the whole of the circuit to earth. 

When SI (and with it, of course, S2) is set to accept the positive-going 405-line video 
signal, SI is open-circuited and the cathode bias components are able to develop the 
large negative bias required. The value of the 1 50-ohm bias resistor is carefully chosen 
so that the bias voltage developed across it sets the operating point of the valve well 
down in the negative region of the I J V e curve— but also well clear of the point at 
which the curve really begins to curve near 7 a cut-off. With components of the values 
shown, the bias voltage developed is about 3-5 V, which means that the grid circuit of 
the valve can accommodate a maximum video signal amplitude of about that value. 



§15] 2.131 

The Dual-Standard Video Amplifier (continued) 

When SI is set to accept the negative-going 625-line video signal, the two cathode 
bias components are short-circuited to chassis, leaving only the 18-ohm resistor in 
series with the valve. With such a small value of resistance in the cathode circuit, only 
a small bias voltage (about 0-6 V) will be developed across it; and the operating point 
of the valve on the IJV e will be close to the point where V g is zero. 

This is, as you know, the required operating point for the "625" signal. The illus- 
tration opposite also shows that a bias setting of about 0-6 V will allow the grid circuit 
of the valve to accommodate a maximum video signal amplitude of about 3-5 V, as it 
did for the "405-line signal. 

There are three reasons for the presence in the circuit of the 18-ohm resistor. First, 
it serves to protect the valve from the excessively high standing current which would 
result if grid voltage were reduced completely to zero. It does this by virtue of the 
small bias voltage which it develops across itself. 

Second, this small bias voltage ensures that the signal peaks corresponding to the 
highlights of the picture do not extend into the positive grid-voltage region of the 
IJV e curve. If this were allowed to happen, grid current would flow at these 
moments; and a heavy load would be placed on the video detector which would cause 
distortion of the video signal and of the picture appearing on the picture tube. 

Lastly, by careful choice of the value of this resistor and by omitting to put into cir- 
cuit with it any decoupling capacitor, negative feedback is introduced and the gain of 
the valve for both 405-line and 625-line signals can be made the same. This reduces 
the need for adjustment of the Contrast Control every time a different standard is 
selected. 



Some TV receivers avoid the need for changing the video amplifier bias whenever the 
line standard is changed by employing a.c. coupling between the video detector and the 
grid circuit of the VA. In this arrangement, a steady Class A bias voltage is developed 
by the cathode bias components, and the video signals from either standard swing 
about their average values within the linear region of the IJV e characteristic. 

The penalty to be paid is, of course, that the d.c. component of both signals is lost, 
and must be put back in again at a later stage by the technique of d.c. restoration. 

Another disadvantage of a.c. coupling is the sudden, momentary changes which 
occur in the operating point of the valve whenever the mean brightness level of the 
studio scene changes abruptly. This is caused by the flow of current in the coupling 
capacitor as it charges to the level required of it on the new standard. The changes in 
operating point introduced in this way are often large enough to cause distortion of the 
video signal, either because grid current momentarily starts to flow or because part of 
the signal momentarily extends into the curved portion of the characteristic. 

Distortion of this nature can be avoided, at the expense of reduced amplification, by 
restricting the amplitude of the video signal applied to the VA to about 70% of what 
could otherwise be accommodated. 



2.132 



[§15 



Reducing Beat-Frequency Interference in the Dual-Standard Video Amplifier 

A somewhat more elaborate arrangement is adopted in the cathode circuit of some 
Dual-Standard VA's, with the object of further reducing the ever-present danger of 
signal distortion arising from the unwanted presence in the amplified video signal of 
the beat-frequency signals of the respective i.f. carriers on both standards. In this 
arrangement, a tuned rejector circuit is connected in series with the cathode lead com- 
ponents in the circuit shown in the last illustration. 

In the circuit below, the two resistors now marked Rl and R2, and the capacitor 
now marked C2, function exactly as they did before, and SI acts on them in the same 
way. But SI is now arranged to act also on a section of the rejector circuit L1-L2- 
Cl so that this circuit can perform two specific tasks. 

When the switch is set to the "405" position, the rejector circuit functions as a 
parallel-tuned circuit resonant at 3-5 MHz, the total inductance being the sum of the 
series-connected inductors LI and L2. In other words, (LI + L2)C1 = 3-5 MHz. 
When connected in this way, the rejector offers a very high impedance to the difference 
frequency of the 405-line sound and vision i.f. carriers (3-5 MHz) and causes the 
cathode bias of the valve to be extremely high to signals of this frequency. Thus the 
valve offers very little amplification to such signals, and they are prevented from 
reaching the picture tube with any magnitude. 

When the switch is set to the "625" position, the inductor L2 is shorted to earth, and 
the rejector tuned circuit is formed by LI and CI only. With the inductance reduced 
in this way, the circuit has a higher resonant frequency — and this frequency is arranged 
to be 6 MHz, the frequency of the 625-line intercarrier frequency. 

A rejector circuit of this type thus causes the valve to give almost no amplification to 
either the 405- or the 625-line i.f. carrier beat-frequency signals, and thereby reduces 
the likelihood of these signals distorting the picture. 



^ 



REDUCING 

BeATFReauency 
ntnuFmeace 

in the 

DUAL-STANDARD 
VIDEO AMPLIFIER 




You could put the above explanation more shortly by saying that, on 405 lines, 
L1-L2-C1 form a 3-5 MHz trap for the 405-line i.f. carrier; while on 625 lines, Ll-Cl 
form a 6 MHz trap for the 625-line intercarrier signal. 



§15] 2.133 

Contrast Control 

One other item of circuitry is normally located in the video amplifier stage of the 
Dual-Standard Receiver. It is the contrast control, which is usually situated physically 
somewhere between the circuitry of the VA and that of the picture tube. The contrast 
control itself needs always to be operated in conjunction with the brightness control, 
which you will be learning about in the next Section. 

Contrast is a term used in TV to describe the ratio of the illumination of the brightest 
to that of the darkest parts of the reproduced image on the picture tube. A picture 
having excessive contrast is composed of unnaturally black and excessively white 
areas, with few intermediate half-tones. Dark grey suits appear jet black, and ordi- 
nary pale faces snowy white. At the other extreme, all the constituent parts of a 
picture having poor contrast appear in almost the same greyish half-tone, with few 
contrasting dark and bright areas. A pale face now, whatever its owner's state of 
health or emotion, looks just about the same colour as does the dark grey suit he is 
wearing in the studio. . . . 

The contrast of a TV picture is determined by the difference of amplitude between the 
two extreme levels of the picture-signal content of the video waveform applied to the 
picture tube— the levels representing black and white. The further these are apart, the 
greater will be the contrast; the closer they are together, the poorer will be the contrast. 
Regulating the contrast of the reproduced picture is therefore only a matter of devising 
a control which allows the amplitude of the video signal fed to the picture tube to be 
adjusted by the viewer to what he considers gives the best results. 

The brightness control, which can also be operated by the viewer, allows him to 
adjust the overall background brightness of the picture. It should normally be set to 
reproduce as closely as possible the mean illumination level of the studio scene. Since 
the contrasting dark and bright areas of the picture must all be referred to (or com- 
pared with) this steady background brightness level, it follows that the contrast and 
brightness controls are mutually related and need to be adjusted in conjunction with 
one another to produce the best results. 

There are several ways in which the amplitude of the video signal applied to the 
picture tube can be controlled. Three possible ones are shown overleaf. 

In (A), the amplitude of the video signal from the VA is controlled by varying the 
amplitude of the un-amplified signal applied to the grid of the pentode. This is called 
low-level contrast control, and is performed by varying the setting of the potentiometer 
RV1. 

In (B), the amplitude of the video signal produced by the VA is controlled by varying 
the gain of the amplifier itself. This is done by varying the screen-grid potential with 
the aid of the potentiometer RV1. 

In (C), the amplitude of the signal applied to the VA and the gain of the VA itself 
are both unaffected by the operation of the contrast control knob. Instead, the full 
amplitude of the amplified video signal is developed not only across the anode load of 
the VA, but also across a large-value potentiometer connected in parallel with it. The 
value of this potentiometer (again RV1 in the illustration overleaf) is nearly ten times 
the value of the anode load resistor R L . The object is to keep its power dissipation as 
low as possible. The variable contact of the potentiometer is connected directly to the 
picture tube, and enables the amplitude of the video signal applied to the tube to be 
varied simply by adjustment of the control. This method is known as high-level 
contrast control and is the type used in most Dual-Standard TV receivers. 



2.134 



[§IS 



Contrast Control (continued) 

The principal disadvantage of Circuits A and B below is that in both of them the 
amplitude of the signal applied to the sync separator circuit is affected. Variations in 
the amplitude of the sync pulses contained in this signal can easily upset the syn- 
chronisation of the receiver. In Circuit C (given adequate signal strength at the 
aerial) the amplitude of the signal applied to the sync separator circuit is not affected 
by adjustments to the contrast control, and the amplitude of the sync pulses can be 
maintained at a constant level by the AGC circuit which controls the gain of the pre- 
ceding i.f. stages. 

One danger in Circuit C needs to be guarded against. The connection of a com- 
paratively bulky component like a variable resistor in parallel with the anode load of 
the VA can increase the stray capacitances existing at that point. In practice, these 
effects of added capacitance are kept to a minimum by physically positioning the 
potentiometer RV1 as close as possible to the anode circuit of the valve (using a long 
operating shaft extending to the front or side of the receiver). 

In addition, further frequency compensation is provided by the small-value capaci- 
tor CI connected across part of the potentiometer control. Without this capacitor, 
some of the higher-frequency components of the video signal would be lost by 
reason of the extra stray capacitances; and unequal attenuation of the video signal 
would result when the control was adjusted. 




To Sync Separator 



To Picture Tube 




HT+ 



To Sync 
Separator 

--AAA 



From Video 
Detector 





HT+ 

To Sync 
Separator 

AAA-^ 



To Picture 
Tube 



RV1 



To Picture 
Tube 



Three Methods of 

CONTRAST CONTROL 



§15] 2.135 

REVIEW of the Video Amplifier 

The job of the video amplifier is to impart to the video signals coming from the video 
detector on both line standards an amplification of the order of 35 to 50 times, while at 
the same time maintaining unimpaired the shape of the video signals over the whole of 
their wide frequency bandwidth (3-5 MHz on the 405-line standard and 5-5 MHz on the 
625-line standard). 

Most TV receivers employ a single stage for video amplification, so automatically 
introducing a 180° inversion of the signal. On both standards, the video signal is 
positive-going, though in the 405-line system the whole of the signal lies above chassis 
potential, while in the 625-line system it lies below it. 

Peak white is.represented, in the 405-line system, by the maximum positive excursion 
of the waveform above chassis potential; while in the 625-line system it is represented by 
the maximum positive excursion of the waveform above sync level. 

Frequency compensation, needed to stabilise the value of the anode load of the ampli- 
fying pentode when the picture-signal frequency increases, is achieved partly by limiting 
the value of the anode load (and thereby the gain of the valve), and partly by connecting a 
small peaking coil in series with R L . The result aimed at is a frequency response curve 
of the general shape of the curve of G m (2R L ) in the illustration below. 

Haw FREQUENCY COMPENSATHW Imff*** QMN 

-MbGm(3RL) 



„ Ovsr-comp*lMtiW 



Grid biasing is required in the pentode of the video amplifier in order to keep the volt- 
age on its grid sufficiently negative to provide adequate grid-base accommodation for the 
full positive-going excursions of the video signal on both standards. 

The Contrast Control enables the viewer to adjust to his liking the ratio between the 
brightest and the darkest parts of the reproduced image on the picture tube. The control 
is operated in conjunction with the Brightness Control, whose circuit is situated in the 
picture tube stage. 



2136 §16: COUPLING THE VIDEO SIGNAL TO 

THE PICTURE TUBE 

The signal leaving the video amplifier is of sufficient amplitude and of the correct 
polarity to modulate the electron beam of the picture tube, and so to build up on the 
screen the picture transmitted from the studio. The basic connecting circuits are 
shown in the illustration below (from which the contrast control circuitry, which you 
already know about, has been omitted for the sake of clarity). 

Note that the connection shown, again for simplicity, is a d.c. one. In practice, you 
will see that there are objections to this, and that something rather more complex is 
normally used instead. 



Picture Tube 



HT+ 




Brilliance 
Control 



How the VMeo Signal is taken to the Picture tube 



The picture tube itself is nothing more than a specialized form of display-type 
cathode-ray tube (CRT), about which you learnt all you need to know at this stage in 
the last dozen pages of Basic Electronics, Part 5. The actual screen is a coating of 
fluorescent material on the inside face of the tube. 

The electron stream, constantly modulated in intensity by the variations of the video 
signal, is made to travel very fast indeed over the face of the screen in a series of inter- 
laced lines one below the other. The deflections of the beam from left to right, and 
from one line to another below it, are caused by varying the currents flowing in the line 
and field deflection coils, respectively, of the CRT — the whole process being performed 
so fast that the fluorescent glow excited on the screen by the passage of the electron 
beam is renewed before the viewer's eye has had time to note any sign of fading. 



§"6] 



2.137 



Modulating The Picture Tube 

In a modern TV receiver, the video signal is usually applied to the cathode of the 
picture tube, and its polarity is negative with respect to its grid. In other words, the 
highlights of the scene are represented by the most negative excursions of the wave- 
form. The grid is maintained at a constant potential selected by the brilliance control 
(of which more later), and is decoupled by a large-value capacitor so that it is effec- 
tively at earth potential with regard to the variations of the video signal. 

With the cathode made negative with respect to the constant potential on the grid, 
the beam current is affected just as it would be if the grid were made positive with 
respect to a constant-potential cathode. In either case, the brilliance of the picture 
appearing on the screen will be greatest when the cathode is most negative. 

The illustration opposite shows in principle how the video signal is connected to 
the cathode of the picture tube from the anode of the video amplifier. With such a 
connection, the cathode is normally maintained at a positive voltage equal to the 
steady anode potential of the VA, but will faithfully follow the variations in potential 
which occur across it. When there is no picture-signal content in the video signal 
(sync pulses and blanking pulses only), the VA anode will be at a reladvely high 
positive potential because, with the average value of the video signal low, little anode 
current flows and there is only a small voltage drop across the anode load. 

Under these conditions, the bias on the picture tube (the difference between its grid 
and cathode voltages) is highly negative. It is at this point, as you will see in a 
moment, that the viewer should adjust the brilliance control so that the raster on the 
screen becomes just invisible. 

Now when a picture signal appears at the anode of the VA, the cathode potential of 
the picture tube rises and falls in sympathy with the variations of the picture signal and 
the density of the electron beam is modulated accordingly. The greater the amplitude 
of the picture signal, the greater the voltage drop across the VA anode load and the 
lower the anode voltage. The resulting much-less-positive potential on the cathode 
of the picture tube reduces the negative grid bias of the tube and increases the brilliance 
of the picture on the screen. 



Beam 
Current 




Black Level 

t -I 



Picture Tube 
Characteristic 



Black Level 




Beam 
■ Current 



Small 
Video 
Signal 



Average 
Beam Current 



Blacker-than- 
Black Region 
(Beam cut off) 



Average Value 
(d.c. level) 



■ ■ A 

(-) (+) \ 
id Bias x 




Average 
Beam Current 



Large 
Video 
Signal 



Average Value 
(d.c. level) 



When grid bias is Reduced picture brilliance increases 



2.138 



[§16 



Brilliance Control 

You know that the average value (d.c. level) of the video signal represents the average 
illumination level of the studio scene, and that the average brightness of the picture 
produced on the screen is determined by the average value of the grid bias of the 
picture tube. This bias can be controlled by the viewer by manipulation of the 
Brilliance Control knob on the outside of his receiver set. 

The brilliance control circuitry is quite simple, consisting essentially of a potentio- 
meter connected between HT (+) and earth, with its movable slider connected to the 
grid of the picture tube. With the grid made highly negative (V c > V g — see illustra- 
tion below), the beam is cut off. With the grid less negative, a faint raster appears on 
the screen. With both grid and cathode at the same potential (K c = V e and grid bias 
is zero), beam current is much increased and the raster on the screen shows up 
brilliantly. 

How the BnWance Control 

Affects the Raster 



HT + 



HT + 



HT + 




-±- Grid Highly Negative 

(Vc»Vg) 




-±- Grid Less Negative 

(VoVg) 




Grid and Cathode at 
Same Potential 

(Vc = Vg) 



ri^ 




Screen Blank 



Faint Raster 



Brilliant Raster 



Normally, brilliance control should be adjusted until the unmodulated raster is made 
just invisible. This setting corresponds to the black level of the video signal. As 
soon as picture-signal content is added to the latter, VA anode voltage drops, the 
cathode of the picture tube becomes less positive, the intensity of the scanning beam is 
increased above that corresponding to black (invisible) level, and a picture appears on 
the screen with a tonal value according well with that of the studio scene. 



§16] 



2.139 



The D.C. Component of the Picture Signal 

You have just seen the importance of the d.c. component of the video signal in 
controlling the overall brightness of the scene traced out by the electron beam on the 
face of the picture tube, and you will have noticed the straightforward d.c. connection 
used to couple the video signal to the cathode of the picture tube in the illustration on 
page 2.136. Such a connection is essential for coupling a d.c. signal to a subsequent 
stage; for if a capacitor is used instead, the d.c. content of the video signal will be 
blocked, only the a.c. content of the signal will reach the tube, and the picture on the 
screen will appear most unnatural. 

Yet, for all the value of the job it does, there are good technical reasons why the size 
of the d.c. component of the signal should be at least reduced. One of these reasons, 
put forward by some TV receiver manufacturers, is the fact that large changes in the 
illumination of a studio scene will give rise to correspondingly large changes in the 
beam current in the picture tube. These changes impose large load variations on the 
EHT supply to the picture tube; and unless this supply is stabilised in some way, it 
will tend to vary with every change in beam current. This would be most undesirable, 
for variations in EHT affect the deflection sensitivity of the tube and hence the size of 
the picture presented. (A voltage supply which behaves in this way is said to have 
poor regulation.) 

But EHT stabilising circuits capable of producing good regulation are expensive, so 
some manufacturers prefer (despite its disadvantages) an a.c. type of coupling between 
the video amplifier and the picture tube. With this type of coupling, no d.c. com- 
ponent can reach the cathode of the picture tube, and no automatic variation can take 
place in the level of the bias on its grid. 

Another disadvantage of d.c. coupling is the accentuation it gives to the type of inter- 
ference known as "aeroplane flutter" which you met briefly earlier on. When an 
aeroplane is flying close to the receiver and to the line-of-sight path between receiver 
and transmitter, the body of the aeroplane will reflect some of the signal from the 
transmitter and the receiver will pick this signal up along with the signal it receives by 
the normal direct route. If the two signals happen to arrive in phase, they combine to 
produce an abnormally strong signal which causes a sudden increase in the amplitude 
of the video signal fed to the picture tube. The result, of course, is an increase in the 
contrast of the picture appearing on the screen. 

But since the aeroplane is moving, the distance between it and the receiver is con- 
stantly changing. So the phase of the reflected signal at the receiver relative to the 
normal signal will progressively change, and will at some time be in complete anti- 
phase to the normal signal. It will therefore subtract from it instead of adding to it. 
There will be a decrease in combined signal strength, a corresponding decrease in the 
amplitude of the video signal— and therefore in the contrast of the picture on the 

screen. k 

. Reflected Signal 



Receiving 
Aerial 




Aeroplane 



2.140 



[§16 



The D.C. Component of the Picture Signal (continued) 

As the aeroplane continues to move nearer and nearer to the receiver, the maximum- 
to-minimum variations in the received signal continue to occur quite smoothly, but the 
time difference between them becomes progressively shorter until the eye fails to detect 
the rapid changes in contrast. This usually occurs at about 20-30 variations per 
second, at a time when the noise of the aeroplane itself can be heard by the viewer. 

Two principal methods are used for coupling the video signal to the picture tube in 
such a way that the advantages of d.c. coupling are retained while its disadvantages are 
minimized. The first is a technique called partial d.c. coupling. The second is to use 
a.c. coupling, but to restore a d.c. component to the video signal after coupling has 
been achieved. 

The illustration below shows a typical circuit providing partial d.c. coupling 
between the VA and the picture tube. 



HT + 



HT + 



Anode 
Load 



RT 



f \ 100K< -T-, 



C 

o-vf 



D.C. 



t >^s Brilliance 



Video 
Amplifier 



i^y R2, 

100K< 



Picture 
Tube 



The video signal at the anode of the VA is developed across two series-connected 
100-kilohm resistors (Rl, R2), the upper one of which is shunted by a 0-1 mfd capa- 
citor. The cathode of the picture tube is connected to the junction between the two 
resistors. To the d.c. component of the video signal the capacitor C behaves like an 
open-circuit; so the amplitude of the d.c. level is reduced by one-half before it is 
applied to the picture tube. (Since R x and R 2 are of equal value, half the level will be 
developed across each.) 

To the higher-frequency a.c. components of the video signal, however, the capacitor 
offers a very low reactance (X c = ll2nfc), which effectively short-circuits R r . Signals 
of such frequencies are therefore developed across R 2 only, and are applied to the 
picture tube with no attenuation. But since the reactance of the capacitor varies 
inversely with frequency, its reactance to the lower-frequency components of the video 
signal is much greater than it is to the higher-frequency ones. As the reactance of the 
capacitor becomes greater, so its shunting effect on the parallel-connected resistor 
(R x ) becomes less, and the attenuating effect of the two resistors becomes greater. 



§16] 2.141 

The D.C. Component of the Picture Signal (continued) 

Take, first, a high-frequency a.c. component of 100 kHz. The reactance of the 
capacitor at this frequency is : 

1 1 100 , 

Xc ~ Wc ~ 6-28 x 10 5 x 0-1 x lO" 6 ~ 6-28 ^ lb ° nmS - 

A reactance of this value is so small compared with the 100 k resistor R^ with which 
it is connected in parallel that almost all the signal will be developed across R 2 and 
practically none across R x . The signal will therefore be applied to the picture tube 
without attenuation. 

But consider a very low-frequency a.c. component like that produced by aeroplane 
flutter— say, 10 Hz. The reactance of the capacitor at 10 Hz is 10,000 times greater 
than it was at 100 kHz, and is therefore about 1 60 k. A reactance of this value is com- 
parable to the value of Rj itself (100 k) and must therefore sharply raise its effective 
resistance. Indeed, the effective resistance of the parallel combination of C and i? x 
now becomes 

R t x X c 100 x 160 16,000 ,,,,-,,, 

— i -? = — — t-tzl = ' = 61-5 kilohms. 

Rj, + X c 100 + 160 260 

The signal amplitude actually applied to the picture tube (i.e., that which is de- 
veloped across R 2 at varying values of the frequency of the a.c. component of the video 
signal) can be worked out by means of a formula. Let V volts be the value of the a.c. 
component of the video signal, and R' be the value of the effective combination R x 
and the reactance of C. Then V = R 2 I(R' + R 2 ) x V. 

In the case of a low-frequency a.c. component like that produced by aeroplane 
flutter, and using the figures calculated above, V = 100/(61-5 + 100) x V = 0-62 
volts. 

So with V reduced by a factor of 0-62 before it is applied to the picture tube, the 
interference caused by aeroplane flutter will be reduced by a similar ratio. 

When purely a.c. coupling is used to couple the video signal to the picture tube, the 
lost d.c. component can be reinstated by the circuit technique of d.c. restoration. For 
economic reasons, the technique is seldom employed nowadays in TV receivers, but 
you may sometimes find it in receivers of earlier design in which grid modulation was 
employed. 

An alternative technique is called d.c. clamping. It is similar in operation to d.c. 
restoration, but requires a continuous source of operating pulses called keying or 
clamping pulses. 

Put shortly, the differences between the two techniques are as follows : 

The d.c. restorer circuit operates on the most positive (or, alternatively, on the most 
negative) excursion of the video signal waveform in any given period, and restores this 
level to a predetermined potential. 

The d.c. clamping circuit operates on any selected part of the video signal waveform 
and restores this level to a predetermined potential. The circuit is more versatile than 
the d.c. restorer, but more complex. For this reason, clamping circuits are generally 
confined to TV transmitting equipment, and are seldom used in receivers. 



2.142 



THE SERIES CONTINUES 

The story of the British Dual-Standard receiver is now continued in 
Part 3 of this Series on Basic Television, beginning with an account of 
how the synchronising pulses are separated from the picture-signal 
content of the video waveform, and then how the field sync pulses used 
to enable the field scan generator to produce an accurate vertical scan 
of the picture tube are separated from the line sync pulses used to 
enable the line scan generator to produce an accurate horizontal scan 
of the tube. 



INDEX TO PART 2 



2.143 



Acceptance Angle 
Acceptance Bandwidth 
Aerials 

Aeroplane Flutter 
AGC Blocking 
Audio Amplifier 



2.44 

2.25, 2.39 et seq. 

2 A, 2.17 et seq. 

2.139 

2.102 

2.114 



Automatic Gain Control (AGC) 



2.9 



Bandpass Filter 2.51, 2.97 

Bat-wing Aerial 2.40 

Beat-frequency Interference (in VA) 2.132 

Bias levels in VA 2.127 et seq. 

Bottom Capacitance Coupling 2.79 

Brilliance Control 2.138 

Cascode Amplifier 2.61 et seq. 

Channel Selection (in VHF Tuner) 2.66 et seq. 

Clamping Pulses 2.141 

Conical Dipole Aerial 2.40 

Contrast Control (in VA) 2.133 et seq. 

Knob 2.9 

Corner Reflectors 2.38 

Critical Coupling (of tuned circuits) 2.97 

Cross-Modulation 2.101 

Damping 2.96 

D.C. Clamping 2.141 

— component of Picture Signal 2.139 

— Restoration (in some types of VA) 

2.131, 2.141 

De-emphasis Filter 2.114 

Delta Matching (of feeder cable to aerial) 

2.35, 2.47 

Diplexer 2.49 et seq. 

Dipole Aerials 2.19 et seq. 

— Length 2.23 et seq. 
Directors 2.28 et seq. 
Double Yagi, with Skeleton Slot 2.35 

End-fed Dipole 2.48 

Feeder Cables, from Aerial 2.46 et seq. 

Field Scan Section of Receiver 2.11 

Flutter, Aerial 2.27 

— , Aeroplane 2.139 

Folded Dipole 2.30 

— Slot Aerial 2.34 
Forward Gain (of aerial array) 2.28 
Frame Grid Valves 2.64 et seq. 
Frequency Changers (VHF) 2.68 et seq. 

(UHF) 2.78 et seq. 

— Compensation (in VA) 2.124 

— Discriminator 2.104 

— Response, of Aerial 2.25 et seq. 

in IF Amplifier 2.92 et seq. 

in Tuner 2.80 et seq. 

in Video Detector 2.118 et seq. 



Front-to-Back Ratio 
Full- Wave Dipole Aerial 
Ghosts (TV) 

Grid Modulation (in VA) 
Grounded-Cathode Triode 
Grounded-Grid Triode 



2.27 
2.20 
2.33, 2.42 et seq. 
2.123 
2.61 
2.61 



H-Aerial 2.26 

Half- Wave Dipole Aerials 

2.19 et seq., 2.22 et seq. 
High-level Contrast Control (in VA) 2.133 

Home-made Aerials 2.52 et seq. 

2.6, 2.83 et seq. 



IF Amplifier 
"IF Amplifier A" 

2.83, 2.89 et seq., 2.98 
"IF Amplifier B" 2.83, 2.102 

Incremental Tuning 
Intercarrier Buzz 
— Signal 2.8, 2.102 

Interference Limiter (at VHF) 

Keying Pulses 

Line Scan Section of Receiver 

Loudspeaker 

Low-level Contrast Control (in VA) 

Miller Effect 

Mixer 

Modulating the Picture Tube 

Multi-Element Aerial Arrays 

Mutual Conductance (G m ) 



2.29 



et seq. 

et seq. 

2.67 

2.104 
et seq. 

2.113 

2.141 

2.12 
2.115 
2.133 

2.60 

2.5 

2.137 

et seq. 

2.65 

2.117 
2.63 

2.97 
2.126 



Negative Demodulation 
Neutralising Capacitor 

Over-coupled Tuned Circuits 
Overshoot 

Partial D.C. Coupling 2.140 

Peaking Coils 2.63, 2.125 

Polarisation (of transmitted signals) 2.31 

Positive Demodulation 2.117 

Power Gain of Aerials 2.27 

Power-Supply Section of Receiver 2.13 

Radiation Acceptance Pattern 2.26 

R.F. Amplifier Stage of Tuner 2.5, 2.60 

Ratio Detector 2.114 

Reflectors 2.26, 2.38 

Rejector Circuits 2.84 et seq. 

Resonant Lines 2.72 et seq. 

Ringing 2.126 

Screening (in UHF Tuner) 2.74 
Shaping, of Frequency Response Curves 

2.96 et seq. 

Signal Power Distribution 2.92 et seq. 



2.144 



Signal Strength Meter 
Signal-to-Noise Ratio 
Skeleton Slot Aerials 
Skin Effect 

Slot Aerials 2.31 

Slotted Cylinder Transmitting Aerial 
Sound Section of Receiver 2.9 

— Signals, Development of 2.110 

Sound-on- Vision 
Stacked Aerial Arrays 
Sync Section of Receiver 2.10 

T-Matching (of feeder cable to aerial) 

Traps 2.84 

Triplexer 

Tuner Stage of Receiver 2.5, 2.59 



2.41 


Turret Tuning 




2.66 


2.44 








2.35 


UHF Aerials 




2.37 et seq. 


2.74 


— Triode (in RF Amplifier) 




2.76 et seq. 


et seq. 


— Tuner 




2.71 et seq. 


2.36 


Undercoupled Tuned Circuits 




2.97 


et seq. 


Unipole 




2.48 


et seq. 








2.89 


Video Amplifier (VA) 


2.8, 


2.122 et seq. 


2.45 


— Detector 


2.8, 


2.116 et seq. 


et seq. 


Vision-on-Sound 




2.89 




VHF Tuner 




2.60 et seq. 


2.47 








et seq. 


Wire Mesh Reflectors 




2.39 


2.51 








et seq. 


Yagi Aerials 




2.30 et seq. 



BASIC TELEVISION 

Part 3 




C OMMON -COMB 



A Basic Training Manual developed by 

H. A. COLE, C.Eng.. M.I.E.R.E., 

working in conjunction with 

the Editorial and Art Staff of the Publishers. 




OXFORD 

THE TECHNICAL PRESS LTD 



NEW YORK 
THE BROLET PRESS 



Copyright © 1972 by 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 

New York, U.S.A. 

All rights reserved 



First published 1972 
Reprinted 1976 



The words "COMMON-CORE", with device and without device, 
are trade-marks of the Copyright owners 



SBN 291 39412 4 



Made and printed by offset in Great Britain by 
William Clowes & Sons, Limited, London, Beccles and Colchester 



PREFACE 

lhe aim of this Series on BASIC TELEVISION is to explain in simple language the 
physical principles which make television possible and the way in which a typical 
television system works — from the generation of the signal in the TV camera to the 
final presentation of the picture on the screen by your own fireside. The Series is based 
on the two TV systems working in Great Britain today— the very-high frequency (VHF) 
one working on 405 lines per picture and the ultra-high frequency (UHF) one working 
on 625 lines per picture. The receiver considered in Parts 2 and 3 is the British Dual- 
Standard Receiver which is capable, on operation of the "Standard Selection" control, 
of receiving programmes on either of these two considerably different systems. 

Two decisions of particular importance had to be made in planning the Series. 
The first was to describe the working of the TV receiver almost wholly in terms of 
valves, even though in many of the latest single-standard and colour receivers the 
thermionic valve is being progressively replaced by semiconductor devices. This 
decision was made on two grounds. The first was that a large majority of the 
millions of receivers operational in Britain in the second half of 1971 are wholly or 
mainly valve-operated rather than transistorized and that, for technical and economic 
reasons which are more fully discussed in the final Section of Part 3 "trends in tv 
receiver design", the valve will in all probability continue to play an important part 
in TV receivers, especially in those built on the Dual-Standard principle, for a signi- 
ficant number of years to come. The second reason was that, since the COMMON- 
CORE Series as it exists at present is planned on the basis of explaining the working 
of electronic devices in terms of current flow through a valve, it was desirable to 
keep this account of the basic principles on which television works compatible with 
the foundation COMMON-CORE volumes in their present form. 

The other major decision in planning BASIC TELEVISION was to cover black-and- 
white ("monochrome") transmission and reception only, in the interest of keeping 
the descriptions of the various stages in the studio camera, the transmitter and the 
receiver relatively simple and relatively short. With the basic principles involved thus 
established (it is hoped) in the reader's mind, a further Series on Basic Colour TV, 
fully transistorized to reflect modern progress, is currently planned. 

Most of the measurements given in the Series have been expressed (or in Part 1, 
which was first published in 1967, re-expressed) in SI Metric units. In particular, 
"Hertz" and "MHz" have been used in place of "cycles per second" and "Mc/s" 
throughout. But certain measurements either familiar to the viewer {e.g., the sizes 
of picture tube) or else representative of orders of magnitude rather than of precise 
distances have been left in inches, miles, etc., as being more likely in that form to give 
the ordinary reader a clear picture of the point being made. 

The Series has been written and illustrated to take its place in the growing 
COMMON-CORE Series of Illustrated Training Manuals on subjects connected with 
electricity and electronics. Originated in the United States by the distinguished 
New York firm of technical education consultants and graphiological engineers, 

VAN VALKENBURGH, NOOGER & NEVILLE, INC. 



the twenty-one Manuals of which the COMMON-CORE Series now consists have 
already sold over 1,500,000 copies in their British and Commonwealth editions. Six 
of the Manuals have been wholly conceived, written and illustrated in the United 
Kingdom; while all the remainder have been extensively rewritten to conform with 
British terminology and notation. 

The BASIC TELEVISION Manuals presuppose in the reader a working know- 
ledge of the contents of the foundation volumes of the COMMON-CORE Series, 
principally the five Parts of BASIC ELECTRICITY and the six Parts of BASIC 
ELECTRONICS. Prior acquaintance with the two-part series BASIC ELEC- 
TRONIC CIRCUITS will also prove useful when the operation of the TV receiver 
is studied in Parts 2 and 3. 

The BASIC TELEVISION Series has been written, in conjunction with the editorial 
staff of the Publishers, by Mr. H. A. Cole, a Senior Scientific Officer in the Elec- 
tronics and Applied Physics Division of the Atomic Energy Research Establishment 
at Harwell. Mr. Cole is a Chartered Engineer, and a Member of the Institution 
of Electronic and Radio Engineers. All illustrations of a technical nature have been 
drawn by Mr. Cole himself, with the Art Department of the technical press 
responsible for their "decoration" and captioning. 



TABLE OF CONTENTS 

Section Page 

17 Separating the Sync Pulses 3.2 

18 The Scanning Generators 3.16 

19 Developing the Field Scan 3.40 

20 Developing the Line Scan 3.48 

21 The Picture Tube 3.66 

22 Vision Interference 3.87 

23 Automatic Gain Control 3.93 

24 Power Supplies 3.112 

25 Fault-finding in the TV Receiver 3.1 17 

26 Trends in TV Receiver Design 3.127 
Index to Part 3 3.137 
Cumulative Index to Parts 1-3 3.138 




The COMMON c i c W\ CORE Series 



of Basic Training Manuals 
embraces so far the following titles : 

BASIC ELECTRICITY 

BASIC ELECTRONICS 

BASIC SYNCHROS AND SERYOMECHANISMS 

BASIC ELECTRONIC CIRCUITS 

BASIC RADAR 

BASIC INDUSTRIAL ELECTRICITY 

BASIC TELEVISION 



Foreword on International 
TV Systems 



The television set round which this Series has been written is the so-called British 
Dual-Standard Set, which is capable of receiving signals on two distinct line-systems — 
the 405-line and the British 625-line systems. 

If you wonder at the emphasis placed on the word "British" in that phrase, "the 
British 625-line system", the reason for it is that it has regrettably not yet been possible 
to secure international agreement on all the technical details of any standard 625-line 
system. 

For some time past, it has been the aim of the C C I R {the Comite" Consultatif 
International des Radio, or International Radio Consultative Committee) to persuade 
all the countries of the world to adopt a common TV system, on the grounds that it 
would be of great benefit to everyone from the point of view of convenience, ease of 
programme exchange, and manufacturing economy. Although complete agreement 
is still a long way off, progress has certainly been made over the past few years. 

There are at present seven major TV systems in the world : the American 525-line, 
the French 625-line, the French 819-line, the West European 625-line, the East 
European 625-line, the British 405-line, and the British 625-line systems. The 
British 405-line system is due to be gradually discontinued over the next few years 
and will eventually be replaced by a 625-line system. 

Unfortunately, not all European countries — even the Western ones — agree on the 
technical details of a standard 625-line system. It is true that they agree on such 
important features as aspect ratio, scanning sequence, method of interlacing and a 
few others ; but differences still exist over (for example) the choice of vision bandwidth, 
channel spacing, sound-to-vision carrier spacing, and the degree of modulation which 
shall correspond to black level. These differences, though not very great, can some- 
times prevent satisfactory exchange of two 625-line programmes. For example, the 
625-line system employed by Belgium and France uses amplitude modulation for the 
sound carrier, whereas all other European countries use frequency modulation. 
Similar differences exist elsewhere in Europe over the relative spacing of the sound 
and vision carriers. 

The Western European and Eastern European systems differ mainly in the values 
chosen for channel width and vision bandwidth. The Western European system uses 
a 5 MHz vision bandwidth and 7 MHz channel spacing, whereas the Eastern European 
system uses a 6 MHz vision bandwidth and 8 MHz channel spacing. 

The British 625-line system differs from both European systems in that it uses 
a 5-5 MHz vision bandwidth and 8 MHz channel spacing. Other differences concern 
the width of the vestigial sideband and the setting of the black level. 



3.2 



§17: SEPARATING THE SYNC PULSES 



The signal leaving the Video Amplifier is of sufficient amplitude and of the correct 
polarity to modulate the electron beam of the picture tube. But you know from page 
1.68 of Basic Television, Part 1, that "if the image seen by the viewer is to be a faithful 
reproduction of that sent out by the studio, it is essential that the scanning spot shall 
move across the picture tube in the receiver at the same speed and at the same time as 
the scanning spot moving across the target of the camera tube, and that it shall at all 
times occupy the same relative position in its scanning field. If any of these con- 
ditions are not realised, it will be impossible to keep the picture steady at the receiver; 
and it may either drift across the screen, dissolve into multiple images, or even break 
up altogether." 

To ensure accurate synchronisation between transmitter and receiver, a series of so- 
called sync pulses were, as you know, mixed with the picture signal in the studio and 
transmitted with it so as to provide a means of keeping the field and line circuits in the 
receiver operating exactly in step with similar circuits in the studio camera. You must 
now see how these sync pulses are separated from the picture-signal content of the 
video waveform, and then how the field pulses used to enable the field scan generator 
to produce an accurate vertical scan are separated from the line pulses used to enable 
the line scan generator to produce an accurate horizontal scan. 

Later on, you will see how the separated pulses are developed so as to make them 
capable of synchronising their respective scanning generators; and how the latter are 
enabled to produce scanning waveforms accurately synchronised and of suitable shape 
to be applied to the scanning coils of the picture tube. 

In block diagram form, the process can be followed in the illustration below: 



■H.T.W 



Video 
Amplifier 




Scanning 
Coils 



Video Signal Cathode / \ 



Field Sync 
Pulses 



Video Signal 



Sync 
Separator 



Line and 
Field Sync 
Pulses 




Field Sync 
Separation 



Line Sync 
Separation 



How the 

Field and Line Sync Pulses are 
SEPARATED and DEVELOPED 




Line Scan 
Generator 





Picture Tube 

. Scanning 
Current 
Waveforms 



Line Sync 
Pulses 



§17] 3.3 

The Sync Separator 

Separation of the sync pulses from the video signal takes place in a stage which 
occurs immediately after the Video Amplifier, known as the sync separator. At this 
stage the video signal is of large amplitude (typically 50-100 V) and, for both 405 and 
625-line systems, of negative-going polarity — which means that the picture signal 
becomes progressively more negative as the brightness of the scene increases. The 
shapes of the two waveforms are as follows : 

- 100% Sync Level 

- 77% Black (& Blanking) Level 



100V 
Typical 




100 V 
Typical 



Jl 




625-LINE 

19% Peak White Level 
0% 



0% 



0-3% Sync Level 

- 30% Blanking Level 

- 38% Black Level 



405-LINE 



- - 100% Peak White Level 



The Video Signal 
Waveforms 

at the Anode of the 

Video Amplifier 



You can see that on both line standards the sync pulses extend into the blacker-than- 
black region of the waveforms. In the 625-line waveform, the sync pulses extend from 
about 77% to 100% of the modulated vision carrier, and the picture signal from about 
19% to 77%. Thus, the overall amplitude of the video waveform, expressed as a 
percentage of the carrier modulation, is (100-19=) 81%. Of this 81%, (ff x 100=) 
71-6% is occupied by the picture signal, and (ffx 100 = ) 28-4% is occupied by the 
sync pulses. So with a peak-to-peak video signal amplitude of 100 V, 28-4 V would 
be represented by the sync pulses and 71-6 V by the picture signal content. 

In the 405-line waveform, almost the whole of the vision carrier amplitude is occu- 
pied by the video signal modulation ; and of this amplitude, 30% is occupied by the 
sync pulses and (100-38 = ) 62% by the picture signal. The remaining 8% is occu- 
pied by the blanking and black level difference — claimed by the BBC to be nowadays 
zero but theoretically determined by the contrast range of the camera tube. So with 
a peak-to-peak video signal amplitude of 100 V, 30 V would be represented by the 
sync pulses and 70 V by the picture signal-plus-blanking level content. 

These proportions are almost the same as for the 625-line waveform, so that the 
composition of the video signal which the sync separator is required to process is 
about the same on both line standards. 

Since the polarity of the signal is also the same, no switching is required at the sync 
separator stage of the British dual-standard receiver. 



3.4 



[§17 



The Sync Separator {continued) 

There are several theoretical ways of separating the sync pulses from the video 
waveform, but for some years most receivers have employed a simple and efficient 
method based on a single pentode valve. The circuit of a typical sync separator of this 
type is shown below: 




Video Signal 
to Picture Tube 



Sync Separator 
Pentode 



Video 
Amplifier 




HT(+)200V 




To Field Sync Separator 
To Line Sync Separator 



R1 C1 

•-vVHr-rf 

22K 05 



"WUM^UUJ^ 



T" .j Line Pulses 



Field Pulses 

Line Pulses 




The sync separator pentode shown has its cathode connected directly to earth. The 
control grid is also connected to earth through its grid-leak resistor R 2 . The initial 
bias on the valve is therefore zero. 

The negative-going video signal from the anode of the VA is applied to the control 
grid of the separator through R x and C ± . R x is used only to isolate the input capaci- 
tance and the (low) input impedance of the separator from the anode circuit of the VA: 
without it, the capacitance would affect the frequency response of the VA and the 
impedance would damp its anode load. The presence of Ri also serves to reduce any 
noise which might be present with the video signal, and which could impair the 
accurate triggering of the field and line scanning generators. 

Q and R 2 together give a coupling time constant of long duration which, in coupling 
the video signal to the separator grid, removes its d.c. level — thereby dividing the 
signal into equal areas above and below a zero-voltage reference level. This places 
the sync pulses and part of the picture signal content in the positive region of the video 
waveform, and the major part of the picture signal in the negative region. 

During the positive regions of the waveform, the valve grid takes on a positive 
polarity; and grid current starts to flow which causes C x to charge to the most positive 
excursions of the waveform. These are, of course, the tips of the positive sync pulses. 

During the periods between successive sync pulses, the charge built up on Q begins 
to leak away slowly through the large-value (1 M) grid-leak resistor R 2 . Before it has 
lost much of its charge, however, the next sync pulse arrives, and the charge lost 
through R 2 is replenished by grid current. 

The net result is that the right-hand plate of C x acquires a steady negative polarity, 
and a steady negative voltage is developed across R 2 . (This, you will no doubt have 
recognised, is the familiar "leaky-grid" method of producing a negative bias for the 
valve.) 



§17] 



3.5 



The Sync Separator {continued) 

The steady negative bias produced at the grid of the sync separator (typically, it is 
of the order of 2 or 3 V) is enough to bias the valve beyond the point of anode-current 
cut-off on the I a V g curve — even though current still flows through the effective diode 
formed by the grid and the cathode. These biasing conditions are illustrated below. 




BUS 



Anode Saturation 
Voltage 



Time »- 



To facilitate the build-up of cut-off bias, both the anode and screen-grid voltages of 
the pentode are deliberately made much lower than usual. This is achieved by supply- 
ing HT(+) voltage to these electrodes from potential divider circuits across the HT 
line. The anode is supplied via R 3 and R 4 , and the screen grid via R 5 and R 6 . 
Typical voltages so supplied are 96 V for the anode and 72 V for the screen grid. 

Such operating conditions give the pentode a very short grid base — which means 
that a much smaller negative voltage is required on the grid in order to cut off the flow 
of anode current. (Note at this point that the negative voltage built up at the grid of 
the sync separator is used in many TV receivers to control the gain of the vision 
circuits. You will learn more about this when you come to study Automatic Gain 
Control.) 

You will see from the illustration that the average value of the applied video signal 
"sits" on the negative bias which it creates at the grid of the pentode. By careful 
choice of the components Q-Ra, R 3 -R 4 and R 5 -R 6 , this bias can be made of such a 
value that only the sync pulses extend into the conduction region of the IJV, curve, 
the remainder of the signal (i.e., the picture signal content) being left in the cut-off 
region. This, of course, means that the pentode will only conduct during the period 
of the sync pulses, and that these will appear at the anode amplified and inverted. 

Actually, a little more than this happens. You will notice that the sync pulses not 
only reach into the / a -conduction region of the curve, but that they also extend right 
up to the grid-current region. Thus the maximum possible value of anode current 
will be caused to flow by the sync pulses; and this, coupled with the unusually low 
anode voltage, will cause the valve to saturate, or "bottom". The valve in fact 
bottoms well before the sync pulses reach the grid-current region, and clean steep- 
sided inverted sync pulses appear at the anode shortly after the sync pulses at the grid 
have started to turn-on anode current. In this way, some unwanted curvature of the 
edges of the sync pulses at the grid (caused by the series resistor Ri and by the input 
capacitance of the pentode) is "straightened out" at the anode, thus preserving the 
sharp-sided pulses originally existing at the anode of the VA which are required to 
ensure positive separation of field from line pulses in following circuits and clean 
triggering of the field and line scanning generators. 



3.6 



[§17 



The Sync Separator (continued) 
Other points to be briefly noted about the sync separator are : 

(a) The circuit relies on the fact that the video signal applied to its input is of 
negative-going polarity (picture signal increasing in a negative direction and sync 
pulses increasing in a positive direction). A signal of such polarity is also required by 
the cathode of the picture tube. Had this tube required a positive-going signal (as it 
might have done if it had employed grid modulation), the VA would have had to pro- 
vide such a signal. This would not have suited the sync separator, and an extra 
valve would have been needed purely for the purpose of inverting the signal. 

(b) The series resistor R t and the saturated operating condition of the pentode both 
contribute valuably to the suppression of noise pulses which might exist on top of the 
sync pulses. 

(c) Since the sync separator "clips off" the top of the video signal to remove the 
sync pulses from the picture signal, the stage is sometimes referred to as the "sync 
clipper". 

Separating the Line Sync Pulses 

Now that the mixed sync pulses have been separated from the video signal, the next 
thing to do is to separate the field and the line sync pulses from one another. 

Consider, first, the line sync pulses of the 405-line standard. They are simpler be- 
cause they contain no equalising pulses to complicate matters. The illustration shows 
the sync pulse sequence which appears at the anode of the sync separator pentode at 
the end of an even-numbered scanning field. (For simplicity, the lines are numbered 
in the order in which they are produced, ignoring the demands of interlacing.) 

Sync Pulses at Anode 
of Sync Separator 

A. 




h* 


Field Sync Pulse 


—\ 








f 











Half-Line Pulses 



u Line 
Periods 



.» -, — Line 
Numbers 



Differentiated Sync Pulses 
developed across the R of 
the RC Circuit on page opposite 




AT THE 



You see the field sync pulse divided into eight half-line pulses, each of 40 p.s dura- 
tion, and preceded and followed by a number of line sync pulses. The line scan 
generator requires sharp-edged pulses of short duration and negative polarity, occur- 
ring at line frequency even during the period of the field sync pulse. As things stand, the 
mixed sync pulses at the sync separator anode are of the correct polarity and of the 
correct frequency— including the necessary half-line frequency; but they are of 
different durations, and they are not short enough. 



§17] 



3.7 



Separating the Line Sync Pulses {continued) 

The solution is to pass the pulses through a type of RC differentiating circuit 
such as you learnt about in Basic Electronic Circuits (page 1.14). You will recall 
that such circuits consist of a capacitor connected in series with a resistor, the out- 
put being developed across the resistor. For an RC circuit to function as a dif- 
ferentiating circuit, the values of capacitance and resistance must be chosen so that 
five times the product of their time constant (5CR) is less than the duration of any 
pulse applied to it. In other words, 5CR must be less than t where t is the pulse 
duration. 

In the 405-line system, the duration of the line sync pulses is about 9 (xs. Thus 
the time constant of the differentiating circuit must be not greater than t/5, or 1-8, 
(i.s. In practice, the time constant is generally made even shorter than this, a value 
of 0-5 [xs being typical. Representative values of C and R to produce a time con- 
stant of this value could be 50 pF and 10 k. The resulting 5CR product of 2-5 jxs 
is, of course, well within the required limit of 9 fj.s. 

In the circuit below, the resistor marked R 2 is used to isolate the line sync separa- 
tion circuit from the field sync separation circuit. The latter is, of course, also 
connected to the sync separator anode and has an isolation resistor, R y , of its own. 



I 



Ry 

r-V\A- 



A 



V C> 



To Field Sync Pulse Separator Circuit 



Rz 



inrL_n_n_r 



Sync 
Separator ' 
Pentode | 



Mixed Sync 
Pulses 



3P I J 

R > I 



tVtV 

Differentiated Sync Pulses 
to Line Scan Generator 



RC Differentiating Circuit 



SEPARATING the LIME SYNC PULSES 



When the sequence of mixed sync pulses shown in the illustration opposite is 
passed through the differentiating circuit, a corresponding sequence of positive and 
negative spikes is produced, the whole sequence having a mean value of zero because 
the capacitor has removed any d.c. content. Every negative-going pulse of the input 
waveform produces two corresponding spikes at the output — one positive and one 
negative — the negative spike when the input waveform suddenly drops to a more 
negative value, and the positive spike when it returns to the no-pulse level. The 
important point is that all the spikes, whether positive or negative, are of exactly the 
same shape and duration — even though some were produced from pulses of very 
different duration. 

The negative spikes, which occur at regular line and half-line intervals, are taken 
to the line scanning waveform generator, where they are used (either directly or 
indirectly) to synchronise its operating frequency. The positive spikes are either 
rejected (by means of a diode of some sort), or simply ignored because, as you will see, 
the scanning generator will generally be insensitive to pulses of positive polarity. 



3.8 



[§17 



Separating the Line Sync Pulses (continued) 
625-line Operation 

The 625-line method of treating the line sync pulses is exactly the same, except that 
the value of the RC time constant needs to be less than 4-7 \xs. In a dual-standard 
receiver, the time constant is therefore chosen to suit the 625-line sync pulses, for it 
will then automatically be short enough for the 405-line sync pulses. No switching is 
therefore necessary in this stage of the receiver. 

Separating the Field Sync Pulses 

You will recall that the field sync pulse is composed as follows: 

In the 405-line system, it consists of a cluster of eight pulses, each of 40 u.s duration, 
occurring at twice line frequency. The field pulse is preceded and followed by 
ordinary line sync pulses of about 9 (xs duration each. 

In the 625-line system, the field sync pulse is composed of a cluster of six 27 ;xs pulses 
occurring at twice line frequency, preceded and followed by five equalising pulses of 
2-3 fxs duration also occurring at twice line frequency. The equalising pulse clusters 
are themselves preceded and followed by normal line sync pulses of 4-7 \ls duration. 

Thus in both systems the field sync pulse is made up of a carefully calculated cluster 
of other pulses of much shorter duration than its own. It is the job of the field sync 
pulse separator circuit to separate out these pulse clusters from the mixed sync pulses 
coming from the anode of the sync separator. 
405-line Operation 

The most widely-used method of separating out the field sync pulses is to pass the 
mixed sync pulses through a simple RC integrating circuit like that shown in the 
illustration below. The type of circuit was explained in Basic Electronic Circuits, 
pages 1.15 and 1.16. It functions in precisely the opposite manner to an RC differen- 
tiating circuit (if you are mathematically minded, you will know that integration is the 
opposite of differentiation). 

HT(+) 



Field Sync Pulse 

to Field Scan Generator 




Sync Separator 
Pentode 



RC Integration 
Circuit 

Mixed Sync i •* 

p U ses i f jj U9 $m p QILSi sipMAT/OH 



In an RC integration circuit, the input waveform is applied to a series arrangement 
of a capacitor and resistor, as before, but the output waveform is developed across the 
capacitor. The mixed sync pulses are applied to the integration circuit and, by careful 
selection of the values of capacitor and resistor to give the correct time constant, the 
field sync pulse required is developed across the capacitor. 



§17] 



3.9 



Separating the Field Sync Pulses (continued) 

The time constant of the RC combination in the integration circuit is chosen to be 
about 25 (jls, which is comparable to the duration of the individual half-line pulses 
which make up the field sync pulse. Typical values of C and R to give such a time 
constant are 500 pF and 47 k respectively. 

Consider what happens when a train of mixed sync pulses passes through the integra- 
tion circuit, beginning at the moment when the last three line sync pulses of an even- 
line field are about to occur at the anode of the sync separator. The pulse sequence 
which follows is shown in the illustration. (To make the explanation easier to follow, 
the pulses have been numbered 1 to 14 just as they occur.) 



Line Sync 
Pulses 




fiiip sy*c miss 



Field Sync 

Output Pulse 

from Integrator 



When Sync Pulse No. 1 is applied to the integrating circuit, the capacitor begins to 
charge up through the 47 k resistor towards the peak voltage of the pulse, on a time 
constant of 25 |i.s. This rate of accumulation of charge, however, is slow compared 
with the short duration of the pulse (9 jxs), and before the voltage across C has had a 
chance to build up to any significant amount, the pulse has ended. The small 
quantity of charge accumulated leaks away through R, having ample time to do so 
because the time between the line sync pulses is almost 100 f*s. So by the time the 
next sync pulse arrives, C will have completely discharged. 

When Sync Pulse No. 2 arrives, C again begins to charge up towards the peak volt- 
age of the pulse, but is again prevented from charging very far— and the same thing 
happens when Sync Pulse No. 3 passes through. 

But when the first of the broad half-line pulses arrives (Pulse No. 4), it is 40 u.s long 
and C has a much longer time in which to accumulate charge. So a much greater 
charge is accumulated, and a much larger voltage across C is built up. At the end of 
Sync Pulse No. 4, C begins to discharge as before; but now only some 10 fxs elapse 
before Pulse No. 5 arrives. Little charge is lost during this time. 



3.10 Bl7 

Separating the Field Sync Pulses (continued) 

The arrival of Pulse No. 5 allows C to recover not only the small amount of charge 
just lost, but also to acquire an additional charge — and the cumulative build-up of 
voltage across C continues for the remaining half-line pulses, so that at the end of 
Pulse Number 11, a substantial potential has been built up. 

After Pulse No. 1 1, the usual 10 \is period elapses, during which a little of the charge 
on C leaks away; but the next pulse to appear (No. 12) is now an ordinary line sync 
pulse, whose duration is only 9 (j.s. Very little charge is regained by C during its 
presence, but after it has gone a full 100 jxs elapses before Pulse No. 13 arrives. 
During this 100 [is, C loses most of the large charge it acquired during the presence of 
the half-line pulses, and during the remaining line sync pulses (No. 14 onwards) the 
situation reverts to that which existed at the end of Pulse No. 3. 

Thus you will see that a significant voltage is built up in a series of jumps across C as 
each half-line pulse is applied to the integrating circuit, but at no other time. A 
signal is thus produced which is representative of the field sync pulse as a whole. This 
signal, although not yet particularly sharp-sided, can be made quite suitable for 
initiating the flyback operation of the field scanning generator. 

Triggering Differences in the 405-line System 

A field sync pulse such as the one just described is produced by the RC integrating 
circuit at the end of every field period in the 405-line system. But there is an important 
difference between the build-up times of the field sync pulses produced at the end of the 
odd-line and even-line fields. This difference can be seen in the illustrations of the 
output waveforms produced by the field sync pulse integrator circuit on this and 
the next page. 

The difference arises in the following way. At the end of the even-line field, the 
field sync pulse sequence starts one full line period (about 100 jxs) after the last line 
sync pulse has appeared. This period gives the integrator capacitor time enough to 
get rid of the small quantity of charge it has accumulated during the last line sync 
pulse, and ensures that it starts from a state of zero charge when the first of the half- 
line pulses which make up the field sync pulse arrives. At the end of the field sync 
pulse sequence, the first of the line sync pulses of the next (odd-line) field arrives only 
about 10 (as after the last of the half-line pulses, and so delays the decay of the charge 
accumulated by the capacitor; but the next line sync pulse does not arrive until 
1 00 (xs later — during which time the capacitor loses most of its charge. From then on, 
it acquires and loses equal amounts of charge as each line sync pulse arrives — as 
shown in the illustration below. 



TT^IUIMIFTT 




mi* S¥*C PULSE 

produced by the INTEGRATOR 
at the end of the 

evenuNi mi* 



§17] 



3.11 



Separating the Field Sync Pulses (continued) 

But at the end of the odd-line field, the field sync pulse sequence starts half-way along 
the last line, and therefore only about 50 [is after the onset of the last line sync pulse. 
The integrator capacitor has not enough time to get rid of the full charge it accumulated 
during the last line sync pulse, and some charge remains when the first of the half-line 
pulses arrives. The capacitor does not, therefore, start charging from zero when the 
field sync pulse sequence arrives, and this affects the shape of the leading edge of the 
field sync pulse. 

A similar process takes place at the end of the odd-line field sync pulse sequence. 
The first line-sync pulse of the next (even-line) field occurs a full line period (100 ^is) 
after the last half-line pulse, so now the integrator capacitor is allowed enough time to 
lose nearly all the charge it accumulated during the field sync pulse sequence, before 
the first line sync pulse arrives. This difference in time distribution of the first line 
sync pulse following the end of the respective field sync pulses accounts for the 
difference in the trailing edges of the field sync pulse waveforms produced by the 
integrator for the two successive fields. 



oimjuuuulto 




HHD SYNC WlSC 

produced by the INTEGRATOR 
at the end of the 



The extent of the difference in the leading and trailing edges of the pulses in the two 
successive field scans can be seen in the illustration below, which shows the two 
successive field sync pulses superimposed. Though it may not look much, the 
difference in the leading edges is quite enough to cause noticeable displacement of the 
two picture fields on the picture tube, and so to upset interlacing accuracy. (Re- 
member that interlacing is perfect only when the lines from one field lie exactly mid- 
way between those produced by the preceding and succeeding fields.) 




EVEN and ODD-LINE 
FIELD SYNC PULSES 

SUPERIMPOSED 



3.12 



[§17 



Separating the Field Sync Pulses {continued) 

The importance of this difference in the leading edges of the two field sync pulses in 
the 405-line system is shown in the following diagram— which again shows the field 
sync pulses from two successive fields superimposed. 



x/ \ ^Even-Field 
\' Sync Pulse 




M=0 
(Start of 
Field Sync Pulse) 



to the FIILD SCANS of the 405-Um gYSTHI 



Assume that the clipping (or threshold) level of the limiter which follows the 
integrator (and which you will be reading about shortly) is set to point A on the wave- 
form amplitudes. When the field scanning generator is supplied with a sync pulse 
from an even-line field, its flyback will be initiated at time t x . When receiving an odd- 
line pulse, the flyback is not initiated until the later time t 2 . It is this difference in the 
starting time of the generator which causes the displacement of alternate picture 
fields. Sometimes, when the difference between t x and t 2 is large, the interlacing 
becomes so bad that the lines produced from one field are only slightly displaced from 
those produced during the preceding field. This results in pairs of lines separated by 
large gaps — an effect known as line pairing. 

Some field scan generators are also affected if the period allowed for completion of 
the flyback varies from one scan to another. Since this period is dictated by the 
duration of the field sync pulse, the difference between the trailing edges of successive 
pulses can also affect the operation of the generator. There is therefore urgent need 
of some means of minimising the difference between successive field sync pulses. 

Examine the leading edges of the superimposed field sync pulses, and you will see 
that there are certain points on the waveforms (point B is an example) where the 
difference between t x and t 2 is less than it is at point A. To try to operate at such a 
critical point for long periods, however, is not satisfactory, because it relies on the 
sync pulses remaining at a near-constant amplitude— which they are most unlikely to 
do. 

Many circuits have been devised to overcome the dissimilarity between successive 
field sync pulses in the 405-line system— most of them based on making all the pulses 
start from the same amplitude reference level (earth, for example), thereby overcoming 
the primary cause of the difference in the leading edges. Such circuits work quite 
well in practice, but the very fact that they are necessary is a major disadvantage of the 
405-line method of processing the field sync pulse. 



§17] 



3.13 



Equalising Pulses in the 625-line System 

It was to avoid the drawbacks of the 405-line method of reconstituting accurate 
field sync pulses in the receiver that the equalising pulses were introduced into the 625- 
line system. 

You have already learnt about the shape and duration of these pulses in Part 1 (page 
1.78). Summarising, they are made to occur in two clusters, each occupying a time 
interval equivalent to 2\ line periods (about 1 28 u.s). One cluster appears immediately 
before the arrival of the field sync pulse sequence, and the other immediately after it. 
Each cluster contains five pulses which occur at twice line frequency (the same rate as 
the half-line pulses which make up the field sync pulse sequence), but which are indi- 
vidually of only half the duration of the ordinary line sync pulses. 

Being of such short duration (2-5 jxs), equalising pulses have only a small individual 
energy content, and the integrating capacitor of the field sync pulse separation circuit 
therefore acquires only a small quantity of charge during the time each pulse is present. 
The charge on the integrating capacitor at the beginning and end of the field sync 
pulses is thus allowed to build up and leak away at rates that leave it at a level which 
is practically identical for both odd and even-line fields at the crucial moments of 
onset of both the field sync pulse itself and of the first line sync pulse of the succeeding 
field. In other words, the charge content of the integrating capacitor (and therefore 
the voltage developed across it) is quickly equalised for successive fields. 

The illustration shows the field sync pulse sequence for two successive odd and 
even-line fields of the 625-line system. Note how the shapes of the corresponding 
field sync pulses produced by the integrator circuit are affected by the presence of the 
equalising pulses. Although the voltage across the integrating, capacitor is different 
for the two fields at the time of arrival of the first equalising pulse, it has settled down 
to near-zero by the time the field sync pulse sequence starts. At the end of the field 
sync pulse sequence, the voltage is also given time to settle down to near-zero by the 
time the first line sync pulse of the next field arrives. 



Last Line- 
Sync Pulse I Equalising Pulses 
of Odd Field 



Field Sync Pulse 
Sequence 



Equalising Pulses 



VnwniMiunrt 



First Line-Sync Pulse 
of Even Field 




END OF EVEN FIELD 



Field, Sync Pulse produced 
by Integrator, 



THT^^LLI^^n^ir 




END OF ODD FIELD 



The iOM of the WNHiSm HMSiS in the 625-Line SYSTEM 



3.14 



[§17 



The Field Sync Pulse Limiter 

In both line systems, the waveform produced by the field sync integrator circuit 
contains small-amplitude pulses produced by the line sync pulses. It is important 
that these pulses should be removed before the integrator waveform is applied to the 
field scanning generator lest its flyback be initiated before the arrival of the genuine 
field sync pulse. Their removal can be achieved by passing the integrator waveform 
through a simple diode limiter circuit, similar to that shown in the illustration below. 



HT(+) 



C2 
'— ) |— *- Output 




Field Sync Pulse 

Field Scan Generator 



OUTPUT 




The limiter usually consists of a semiconductor diode whose anode is connected to 
a potential which is negative with respect to the potential present at its other electrode. 
The potential to which the anode of the diode is returned is determined by the value of 
the two resistors R 2 and R 3 which form a potential divider across the HT(+) supply. 
The actual value of the potential is determined by the equation: 



V = 



R 2 



R 2 + R; 



xV, 



HT 



Connected in this way, the diode has reverse bias. It will therefore not conduct 
until the amplitude of the negative-going signal applied to its cathode exceeds the 
magnitude of the bias. The bias potential is carefully chosen to be somewhat larger 
than the maximum amplitude of the unwanted line sync pulses present in the integrator 
waveform. These pulses therefore all lie below the bias potential, and so below the 
threshold at which the diode can conduct (often called the "clipping level" of the 
limiter). 

The amplitude of the wanted field sync pulse, however, is considerably greater than 
the bias potential, and the diode is able to conduct for almost the entire duration of the 
pulse save for the small area lying below the bias threshold. 

When the diode conducts, the field sync pulse, minus the unwanted line sync inter- 
ference pulses, is passed to the coupling capacitor C 2 , and thence to the field scanning 
generator. 



§17] 

REVIEW of Sync Pulse Separation 

The sync separator stage accepts the 
negative-going video signal applied to it 
from the video amplifier and extracts from 
it the line and field sync pulses. 

These two pulse chains are then separated 
from one another, and shaped so as to be- 
come suitable for synchronising the fre- 
quencies of the line and field scan generators 
with the corresponding control frequencies 
built into the signal transmitted from the 
studio. 



3.15 




I — /V* — ». To Field Syne S 
L-vV^— *- To Una Sync S 



TElMMJUUir 

1' u 



Field Pulses 
ib Pulses Line 



/ 




run mc mse s&mrm 



Separation of the field sync pulse from 
the train of mixed sync pulses fed into the 
sync pulse separator is achieved by a simple 
RC integration circuit. The time constant 
of this circuit is carefully chosen so that an 
output voltage is generated by it only during 
the presence of the half-line pulses which 
make up the field sync pulse. 



The equalising pulses used in the 625-line 
system effectively obviate the interlacing 
problems which, in the 405-line system, are 
caused by the dissimilarities of alternate 
field sync pulses. 

The equalising pulses ensure that the 
amount of charge retained by the integrating 
capacitor used in the field sync pulse separa- 
tion circuit is identical at the start of every 
new field sync pulse. 



Lost Line- 

of Odd Field ; * 



iTTTTTTlULULJTTnnf 



END OF EVEN FIELD 



innnrmjumwnn" 



END OF ODD FIELD 



Ty*&tMtf\to € W 4US m $ P WStS \*\tom*)MVtm* 



3.16 



§18: THE SCANNING GENERATORS 



You know that every complete picture appearing on the picture tube of a TV 
receiver is built up by modulating the intensity, or brightness, of two interlaced and 
successively presented scanning rasters. Each raster, or half-picture, is known as a 
field and occurs 50 times a second. It is made up of 202^ (in the 405-line system), or 
of 312£ (in the 625-line system), nearly horizontal parallel lines (ignoring in both cases 
the few lines which are lost during the field blanking interval). In the absence of any 
picture signal, the lines which make up the rasters are of uniform brightness — as you 
can see on most TV receivers by removing the aerial, or switching to an unused 
channel, and turning-up the Brightness control. 

A raster is produced by displaying a rapidly-recurring horizontal timebase, which is 
at the same time much more slowly deflected from the top to the bottom of the tube. 
Every line begins at the left-hand side of the tube, and when it reaches the right-hand 
side is rapidly returned to the left ready to start again a little below its predecessor. 
The left-to-right movement of the scanning beam is known as the line scan and the 
fast right-to-left movement as the line flyback. 

The slower vertical deflection from Hop to bottom of the picture tube is called the 
field scan. It starts from the top left-hand corner of the tube and ends midway along 
the last line at the bottom of the raster. (On alternate rasters it starts from midway 
along the first line and terminates at the end of the last line.) Wherever it terminates, 
the scanning beam is rapidly returned to the top of the tube in time for the first line of 
the next field scan to begin. This rapid return to the top of the tube is known as the 
field flyback. 

The horizontal lines, or timebases, of the raster are produced by a circuit known as 
the line scan generator. This is a timebase generator, or oscillator, operating at a 
repetition frequency of 10-125 kHz (in the 405-line system) or at 15-625 kHz (in the 
625-line system). It produces in either case a nearly linear sawtooth waveform of 
current which, after appropriate development, is applied to the horizontal or line 
scanning coils surrounding the neck of the picture tube. 

The slower vertical deflection of the scanning lines is produced by a second timebase 
generator called the field scan generator. This operates at the much lower frequency 
of 50 Hz and also produces a near-linear sawtooth of current which, again after 
development, is applied to the vertical or field scanning coils also situated round the 
neck of the picture tube. 

The repetition rates and starting times of the line and field scan generator wave- 
forms are synchronised with those of the received picture signals by means of the line 
and field sync pulses which you have just seen being extracted from the video signal at 
the sync separator stage. 



The Basic Field-Scan 
Timebase 
Waveform 



Field Flyback 




V 



Scanning 
Raster 



..^\ 



The Basic Line-Scan 
Timebase Waveforms 




Time 



L 'Kar-TTTTTTTT 



§18] 3.17 

The Line and Field Scanning Circuits 

The concern of this Section is to show you how the horizontal (line) and vertical 
(field) scans of the electron beam across the screen of the picture tube are originated, 
and how they are synchronised with the respective timebases used in the studio camera 
by the line and field sync pulses which you now know are available. In the two 
following Sections, you will see how the field scan, and then how the more complicated 
line scan, waveforms are developed until they possess the characteristics of power and 
shape which they need when they are applied to the scanning coils of the picture tube. 

To put the matter more formally, the job of the line and field scanning circuits is to 
generate waveforms of current of the correct shape and at the prescribed repetition 
rates, and to deliver these currents to the line and field scanning coils with sufficient 
power to generate full-width and full-height scans of the picture tube. 

The process normally takes place in three stages. First, a low-power oscillator, 
synchronised with the incoming line or field sync pulses, is used to generate the basic 
scanning waveform. Second, this waveform is raised to the required power level by a 
power amplifier. Third, an impedance-matching transformer is used to couple the 
waveform from the power amplifier to the scanning coils. The power amplifiers plus 
the transformers are generally known as the line and field output stages: the matching 
transformers themselves as the line and field output transformers. 

The basic arrangement is shown in the block diagram below. For simplicity at this 
stage, the sawtooth waveforms at various points are shown as being truly linear. In 
practice, it is often necessary to use sawtooth waveforms which are appreciably non- 
linear in shape. 




Line Sync 
Pulses 



Line Pulse 
Shaper 



Line Scan 
Oscillator 



\ 



Sync 
Separator 




Line Output 



Stage 



Line 

Scanning 

Coils 



Field Sync 
Pulses 




Field Scan 
Oscillator 




Field / 
Scanning 
Coils 



Field Output 
Stage 




%'i:$i$X?$1&&&$&M$ip& l 3$!i 



WtV/«sy*:*"W.W 







3 - 18 [§18 

The Line and Field Scan Generators 

The waveform generator in both the line and the field scanning circuits is a separate 
oscillator, nearly always either of the blocking or of the multivibrator type. In a given 
TV receiver, both oscillators may be of the same type, or oscillators of different types 
may be used to generate the line and field scans respectively. 

The line oscillator circuit is known alternatively as the line scan oscillator, the 
horizontal oscillator or the line scan generator; the field oscillator circuit as either the 
field scan oscillator, the vertical oscillator or the field scan generator. Here they will 
be consistently called the line scan generator and the field scan generator respectively. 

Detailed descriptions of how both the blocking oscillator and the multivibrator are 
used as pulse generators are given in Section 12 of Basic Electronic Circuits (pages 1.69 
to 1.87), so only a brief account of the method of operation of each will be necessary 
here. Let us start with the blocking oscillator. 

The Blocking Oscillator 

This type of oscillator has been used in TV receivers more than any other, because 
of its simplicity and good frequency stability. Only a single valve (usually a triode) or 
a single transistor is necessary. In the valve type the triode often forms half of a 
dual valve of which the other half (typically a pentode) forms the line/field output stage. 

HT(+) 

B 
T1 I 0V- 





The BLOCKIHC OSC/UATOR and its Operating Waveforms 



Over the region A-B, as you can see, anode current in the valve is increasing, and a 
positive voltage is fed back to the grid via the transformer T x and the components C 
and R. This positive feedback causes increasingly more anode current to flow, until 
no more is available from the cathode. At this point the valve is saturated and anode 
current is steady (waveform at point B). 

Also at this point in the operation, however, transformer action ceases (you know 
from Basic Electricity, pages 3.49 to 3.52, that a transformer only works when the 
current through its primary is a varying one). When transformer action ceases, so 
does the positive feedback to the grid. Without this feedback, anode current starts to 
fall — and transformer action once more commences. 



§18] 3.19 

The Blocking Oscillator (continued) 

This time, however, because anode current is falling, a negative voltage is fed back 
to the grid, so that even less anode current is able to flow. The action is thus cumula- 
tive while grid voltage is rapidly driven beyond its cut-off value (point C) and anode 
current flow stops altogether. The energy stored in the transformer inductance, how- 
ever, causes the negative feedback to continue, until a minimum value is reached at 
point D. 

With the valve cut off and not conducting, the charge which the capacitor C 
acquired when the grid was driven positive now begins to leak away through the dis- 
charge path formed by R and the very low resistance of the secondary winding (L s ) of 
the transformer. This discharge is represented by the area D-E-F of the waveform. 
The time comes during the discharge when the voltage across R (which is the same 
thing as the grid voltage) reaches the cut-on point of the valve. Anode current starts 
to flow again, and the whole cycle is repeated. 

The controlling influence in the operating frequency of the blocking oscillator is the 
rate at which C discharges, and hence the time taken by the voltage across R to reach 
the cut-on point of the valve. The faster the discharge rate, s the faster the operating 
frequency, and vice versa. 

In view of the importance of ensuring that the operating periods of the line and 
field scan generators in the receiver should be the same as those in the studio camera, 
it is obviously necessary to be able to adjust "coarsely", as it were, the frequencies of 
the scanning generators in the receiver so that "fine" synchronisation may be applied 
to them by the line and field sync pulses. 

The usual method of "coarse" adjustment of the frequency of a blocking oscillator 
is to vary the value of the discharge resistor R. In practice, this control is only 
infrequently used, and for this reason it is usually of the preset type and situated at the 
back of the receiver. 

When the control is associated with the line scan generator it is known as the Line 
Hold (or sometimes as the horizontal lock). When it is associated with the field scan 
generator, it is known as the Field Hold, or vertical lock. 

The illustration on the next page shows the basic arrangement of a typical frequency 
control, and the effect on the operating waveform of the "fine" adjustment applied by 
the sync pulses as they arrive. 

The negative-going pulses from the line or field sync pulse separator circuits are 
applied (directly or indirectly) to the anode of the oscillator, thence via the transformer 
and the components C and R to the grid, where they are inverted. These now-posi- 
tive-going pulses cause premature discharge of C, and the operating frequency of the 
oscillator is speeded up until it is brought into step with that of the sync pulses. When 
synchronisation is perfect, the sync pulses occur at a point on the waveform coincident 
with that at which it passes through the cut-on value of the grid voltage. 

Work out for yourself what happens when they occur after grid current has started 
to flow, and when the operating frequency of the oscillator needs to be slowed down 
so as to coincide with that of the sync pulses themselves. 



3.20 

The Blocking Oscillator {continued) 

HT(+) 



[§18 




Cut-off - 



~irr 

Sync Pulses 
(Line or Field) 



Line-Hold 

(or Field-Hold) 

Control 




of i MmHrikg tftnYHif r 



The Multivibrator Oscillator 

Unlike the blocking oscillator, the multivibrator is a device which requires the use 
of either two valves or two transistors for its operation. To offset this economic dis- 
advantage, it makes use of simple C and R coupling components to provide the 
necessary feedback, in place of a comparatively expensive transformer. 

HT(+) 




Basic Circuit and Waveforms of the 

MULTIVIBRATOR Oscillator 



§18] 



3.21 



The Multivibrator Oscillator {continued) 

At any point during the operation of the multivibrator circuit shown at the foot of 
the previous page (apart from the exceedingly short change-over periods) one valve is 
conducting heavily at saturation point, and the other is cut off. The rate at which the 
valves change their operating states defines the operating frequency of the circuit. It 
is determined by the values of the C and R coupling components, QR3 and C 2 R 2 
respectively. 

In a "symmetrical" circuit, the time constants formed by QR3 and C 2 R 2 are equal, 
as are the values of the anode load resistors Ri and R 4 . With this type of circuit, the 
"on" and "off" period of each valve is identical, and the waveforms produced at the 
anode of each are of the same duration, but 1 80° out of phase. 

Suppose, however, that the time constant formed by QR3 is made shorter than that 
formed by C 2 R 2 . The multivibrator now becomes "asymmetrical". V 2 will be cut 
off for a longer period than will V 1( and the waveform produced at its anode will be of 
shorter duration than will be that at the anode of V x . 

This form of asymmetry can be brought about either by reducing the values of C lt 
or of R 3 , or of both of them; or by increasing the values of C 2 , of R 2 , or of both. 
However it is done, the Mark-to-Space ratio (Basic Electronic Circuits, page 1.8) will 
be varied, as also will the operating frequency. 

If it is desired to vary the operating frequency without affecting the mark-to-space 
ratio, then either Q andC 2 , or R 2 andR 3 must be varied together by the same amounts. 



HT(+) 



Sync_^_„ 



Pulses 




Line-Hold 

(or Field-Hold) 

Control 



HIGH FREQUENCY (f= 1 /t2) 



C1.R3<C2.R2 



The Asymmetric Multivibrator 



3.22 



[§18 



The Multivibrator Oscillator (continued) 

When the multivibrator is used as the line or field scan generator in a TV receiver, 
it is required to produce a constant-duration waveform at a frequency which can be 
adjusted over a comparatively small range by the line or field hold controls. In other 
words, it must have an asymmetric output waveform, the frequency of which must be 
adjustable. 

This is usually achieved in the multivibrator by making R 3 adjustable, and by 
choosing the value of Q so that the QRg time-constant is different from that of 
C 2 R 2 — as shown in the illustration on the last page, which would be suitable as a 
line or field scan generator. 

Synchronisation of the multivibrator by the negative-going line or field sync pulses 
is achieved by applying them (directly or indirectly) to the grid of the valve which is 
normally conducting during the scan period and cut off during the flyback period. 
The arrival of a sync pulse causes the valve to cut off. This in turn causes the other 
to cut on — and so to initiate the flyback. 

The operation of the synchronised multivibrator, in both its valve and transistor 
forms, is described at greater length on pages 1.78 to 1.81 of Basic Electronic Circuits. 

Shaping the Field Scan Waveform 

As you will see in the next Section, the scanning waveform required at the input of 
the field output stage is of a shape known as an exponential sawtooth. This particular 
shape cannot be obtained directly from either the blocking oscillator or the multi- 
vibrator types of waveform generators, so it is necessary to produce the desired shape 
by other means. It can be done quite simply in either type of waveform generator, 
usually with the aid of a single extra capacitor connected across the output terminal. 
The illustration shows how this is done in the case of the multivibrator. 




Flyback 

Period Field-Scan 
— Period - 



A 



V2 Anode 




HT(+) 



C3 Discharges C3 Charges 



SHAPING the Field Sean Waveform 



If the extra capacitor C 3 were not there, the waveform produced at the anode of V 2 
would be of rectangular shape, as it was in the illustration on the last page. With C 3 
connected, however, the output waveform is modified in the following way. When V 2 
is not conducting, C 3 is fully charged to the voltage of the HT supply. But when V 2 is 
cut on by the cross-coupling from Y lt C 3 discharges very rapidly through the valve 
(which is what gives the sharp vertical edge to the waveform at the beginning of the 
flyback period). 

When V 2 once more cuts off, C 3 is again free to charge up to the HT supply. It 
does so through the anode load of V 2 , on a time-constant of C 3 R 4 seconds. 



3.23 



Controlling Picture Height 

The height of the picture displayed on the picture tube can be controlled by varying 
the amplitude of the exponential sawtooth waveform applied to the field output stage. 
The control which enables this to be done is called, for obvious reasons, the Vertical 
Height (or simply Height) Control. Because it seldom requires adjustment, it is of the 
preset type and is usually situated at the rear of the set. 

The control takes the form of a simple potentiometer connected so that the field 
scan waveform is developed across it. It operates exactly like the volume control in 
a radio receiver. 



HT(+) 




Field Sync 
Pulses . . 

-Ht 

inr 



Field 
Hold 



Vertical 
Height 



? AFIELD SCAN GENERATOR* " 

-'■. '-.:*: " u ^,-<: , ' , ; * rjitaiHfr#r^- ' 



^i-S;a-:;:;«;>:K 



The illustration shows a typical arrangement for a complete field scan generator 
circuit, including the vertical height control just discussed. The capacitor C 4 is a 
large-value blocking capacitor serving to isolate the high potential at V 2 anode from 
the height control, and from the stages which follow. 

Although a multivibrator type of generator has been shown, one of the blocking- 
oscillator type could have been connected in a similar way. 



3.24 



[§18 



Flywheel Synchronisation 

Direct connection of the sync pulses to the oscillator of the field scan generator is 
still standard practice. But in the line scan generator, synchronisation of the operat- 
ing frequency by direct application to the oscillator of the spikes of the line sync 
pulses can give rise to an undesirable effect known as line tearing, and in most modern 
TV receivers a more complicated technique called flywheel synchronisation is applied. 

The trouble with direct application of the line sync pulses to the line scan generator 
is that noise pulses which are often, despite all precautions, still present among the 
sync pulses can have as much effect on'the oscillator as do the line sync pulses them- 
selves — with the result that initiation of the flyback becomes erratic whenever the 
noise pulses (from, e.g., unsuppressed car ignitions, electric drills and some types of 
electric shavers) are present. Disruption of normal synchronisation in this way 
causes random displacement of the scanning lines, or groups of lines, because of 
variation in their respective starting times. This gives the appearance of ragged edges 
to the picture, making it look like the edges of a piece of torn material — whence the 
name line tearing applied to this form of distortion. 




LINE TEARING 



In more modern TV receivers, the frequency of the line scan generator is no longer 
controlled directly by the line sync pulses but by a circuit which compares the frequency 
of the generator with that of the line sync pulses. Whenever the frequency of the 
generator differs from that of the line sync pulses, the comparator circuit produces a 
d.c. voltage which is used to alter the frequency of the generator in such a way as to 
bring it back into step. The greater the difference between the compared frequencies, 
the larger the controlling voltage produced. The polarity of the control voltage 
depends on whether the frequency of the generator is greater or less than that of the 
line sync pulses. 

To prevent a control voltage being produced whenever a burst of interference pulses 
occurs, a deliberate time-delay is introduced so that the control is applied only gradu- 
ally. Application in such a way gives the control a kind of "flywheel" effect and 
enables it to override or ignore sudden, but comparatively minor, changes in frequency 
such as those produced by impulsive interference. 

If the interfering noise is sustained (as it would be if it came from an electric drill), 
flywheel action will lose much of its effectiveness. The resulting line tearing will still 
be less than it would otherwise have been, however; for with flywheel synchronisation 
it is the average frequency variations which are applied to the oscillator, and this 
average tends to even out the sharper variations of frequency caused by pulses of 
noise. 



§18] 



3.25 



Flywheel Synchronisation (continued) 

You will see that two distinct operations are necessary in a flywheel sync circuit. 
The first is the operation of comparing the frequencies of the two waveforms (sync 
pulse frequency with line-scan frequency) and generating a d.c. voltage proportional 
to the difference between them. The second operation is that of controlling the fre- 
quency of the line scan oscillator by means of the d.c. controlling voltage produced in 
the first operation. 

Consider, first, the second operation, and see how the frequency of an oscillator 
circuit can be controlled by means of a variable d.c. voltage. Since the multivibrator 
tends to be more widely used as a line scan generator than does the blocking oscillator, 
start with the basic M/V circuit and see what happens when the resistor R 2 is removed 
from earth and connected instead to a positive voltage (represented in the illustration 
below by a battery connected in series with R 2 ). 



■HTC+) 




V2 Anode Voltage 



V1 Grid Voltage 
(V across R2) 



without V* 
Discharge 
of C2 (with V+) 

V2 Output Waveform 

I Dotted Waveforms show what happens when 
. no V + Voltage is applied. 



V1 Grid Waveform 



t 



(with V+) 



USING A D.C, VOLTAGE 



to Control the Frequency of a 
Multivibrator-Type Oscillator 



3.26 [§|8 

Flywheel Synchronisation {continued) 

At the instant when V 2 suddenly conducts, the fall in its anode potential is com- 
municated to the grid of V 1; and V 1 is cut off. At the instant before V 2 was cut-on, 
C 2 was fully charged, with its right-hand electrode having charged through the anode 
load of V 2 to the potential of the HT supply and its left-hand electrode held at about 
zero volts by the grid current flowing heavily in V^ 

When "V*! is cut off and V 2 starts conducting (i.e., at the moment of switch-over), the 
right-hand electrode of C 2 is reduced from the high HT voltage to a very much lower 
potential. With V + connected, however, instead of its left-hand electrode dropping 
sharply to about zero volts and remaining there for a while, it first drops and then 
immediately tries to "aim" towards the positive voltage which is present at the lower 
end of R 2 . 

In other words, after C 2 has discharged very rapidly to the same voltage as it did in 
the simple circuit, it now seeks to re-charge almost at once towards the additional 
voltage applied to R 2 . But no more time than before is available for this re-charge, 
for the time-constant has not changed and the voltage across C 2 must still rise to 66% 
of the voltage applied to it within a period of time equal to one time-constant. 

Thus the rate at which C 2 re-adjusts its charge must increase; and this means that 
the value of V g0 will be reached sooner. When V KO is reached, as you know, the 
circuit switches over to its other state; so if V g0 is reached more quickly than before, 
it means that the frequency of the circuit has increased. 

It follows that the frequency of an oscillator circuit is increased when a positive 
voltage is applied to the lower end of R 2 ; and the larger the value of this voltage, the 
greater will be the increase in frequency. It is thus possible to control the frequency 
of the line scan oscillator by applying a variable d.c. voltage to the lower end of R 2 . 

Note, by the way, that the same degree of frequency control could be achieved by 
applying the d.c. voltage to the lower end of R 3 , instead of to R 2 . This is not done, 
however, because Q and R 3 are associated with the scan period of the waveform, and 
it is the flyback period which is of importance — as you will shortly see. 

Frequency Comparison Circuits 

Now that you know how the frequency of the line scan oscillator can be controlled 
by means of a variable d.c. voltage, the next step is to see how such a voltage can be 
produced when the frequencies (or phase relationship) of two different signals are 
compared. 

Circuits which compare signals in this way are given a variety of titles, such as fre- 
quency (or phase) comparators, frequency (or phase) discriminators, frequency (or 
phase) detectors, and coincidence detectors. Of the many circuits which have been 
developed over the years to do this sort of job, British TV receiver manufacturers tend 
to favour two particular types. In what follows, these two types will be known as the 
coincidence detector and the phase detector respectively, though you may find both of 
them called by other names in other literature. 

We will begin by looking at the coincidence detector. 



£18] 



3.27 



The Coincidence Detector 

The basic circuit of a simple coincidence detector is shown in-the illustration. The 
circuit consists of a pentode valve to which two separate input signals are applied, one 
to its control grid (GJ and the other to its screen grid (G 2 ). Both signals are positive- 
going; and, for convenience in differentiating between the two, one is shown as being 
of longer duration than the other. 

The screen grid resistor (R s ) and the cathode bias components (R* and C k ) are 
chosen so that the valve passes only a very small anode current when neither signal is 
present. When only Input A pulses are present, the anode current of the valve is 
increased slightly during the period of each pulse. This causes a correspondingly 
small fall in the voltage at the anode, by reason of the increased voltage drop across 
the anode load resistor (R L ). 

Similarly, small increases in anode current (and corresponding falls in anode volt- 
age) occur when only Input B pulses are present. 

When both Input A and Input B pulses are present, their combined influence causes 
a large increase in anode current to flow, but only during the period when both pulses 
are actually coincident. Outside the limits of coincidence the anode current of the 
valve reverts to its normal single-pulse-present values. 

The increase in anode current flow which results from the coincidence of the two 
pulses causes a negative voltage pulse to be produced at the anode of the valve, the 
duration of which equals the period of coincidence. 

You must now see how such a simple coincidence detector needs to be developed 
for use in flywheel synchronisation. 

HT (+) v::„,:,.^,,.,.,.«...,,,, W . W ,« fi 



Output 



.vJLJL- 



Input A 




fin. 



Input B 



G1 Pulses 



sic 



G2 Pul 



DETECTOR 



Output 
Pulses 



zILJLJLJL 

dHririi - 



Coincidence 
Periods 



3.28 



[§18 



The Coincidence Detector (continued) 

When the coincidence detector is used for flywheel synchronisation, the Input A 
pulses to Gi are derived from the line scan oscillator and are coincident in time with 
the flyback period of the line scan waveform. These pulses are truly rectangular in 
shape, having sharp leading and lagging edges. The Input B pulses to G 2 are derived 
from the normal line sync pulses (with which they are exactly coincident in time) 
through a pulse shaping and inverting circuit which will be described later on. This 
latter circuit, as you will see, gives the G 2 pulses a sharp leading edge and a flat top, 
but a sloping lagging edge. 

When the TV receiver is functioning normally, the phase relationship between the 
two waveforms is such that the leading edge of the G x waveform arrives at a point 
about one-third of the way down the slope of the lagging edge of the G 2 waveform. 
The resulting coincidence pulse at the anode of the valve is negative-going, and has the 
same shape as that of the coincidence area (which corresponds to the period of dura- 
tion of anode current flow, and is shown shaded in the illustration below). 



G2 Pulses 




i G1 Pulses 
i 
J 



Coincidence Pulse | X/' 
at Anode | ' 



G2 Pulses 



lTTNl 



i G1 Pulses 
i 

J 



Coincidence Pulse 
at Anode 



T^~" 



NORMAL OPERATION 
G1 & G2 PULSES 

FREQUENCY OF G2 PULSES 

Cretttr 

THAN THAT OF G1 PULSES 



G2 Pulses 



G1 Pulses 




Coincidence Pulse 
at Anode 



FREQUENCY OF G2 PULSES 

less 

THAN THAT OF G1 PULSES 



If the frequency of the line sync pulses tends to increase, the leading edges of the G 2 
waveform will occur earlier than before, and the G ± waveform will appear to occur 
later than before — which corresponds to a reduction in the frequency of the line scan 
oscillator. This has exactly the same effect on the detector circuit as does an increase 
in the frequency of the line sync pulses. So the two waveforms tend to move further 
apart in phase, with the leading edges of the G ± waveform moving further down the 
lagging edges of the G 2 waveform. The area of coincidence becomes smaller than 
normal, resulting in a reduction in the duration of anode current flow and in the 
amplitude and duration of the coincidence pulses produced at the anode. 

If, on the other hand, the frequency of the line sync pulses tends to decrease with 
respect to the frequency of the line scan oscillator, the two waveforms move closer to- 
gether in phase, and the leading edges of the Gi waveform move further up the lagging 
edges of the G 2 waveform. This causes an increase in the coincidence area, with a 
consequent increase in the period of anode current flow and in the amplitude and 
duration of the coincidence pulses at the anode. 



§18] 



3.29 



Generating the Control Voltages in the Coincidence Detector 

Before the coincidence pulses can be used to control the frequency of the line scan 
oscillator, they need to be converted into a steady d.c. voltage whose amplitude must 
vary in accordance with either gradual or sustained variations in the degree of coinci- 
dence. This is done by feeding the coincidence pulses into a simple integrator circuit 
formed by a resistor and capacitor. These components are labelled R^Ci in the 
illustration below. 



HT (+) 



T 

— w 



Coincidence Pulses 



G1 Pulses 



PRODUCING THE 




Control 
Voltage 




COINCIDENCE 
DETECTOR 



IN THE 

Coincidence Detector 



,HT (+) 




G1 < G2 Frequency 



G1, G2 Frequencies 
Coincident 



G1 > G2 Frequency 



When the coincidence pulses are first applied to the integrator circuit, a d.c. voltage 
gradually builds up across Q equal to the average value of the coincidence pulses, and 
proportional to the areas of the individual pulses. (The area of a pulse is equal to 
the product of its amplitude and its duration.) Provided the frequencies of the G x and 
G 2 waveforms remain the same, this voltage will remain steady — as will therefore the 
frequency of the line scan oscillator. 

Should the frequency of the line sync pulses increase, the areas (and therefore the 
average value) of the individual coincidence pulses applied to the integrator will 
become smaller. This will cause an increase in the d.c. level appearing across C r , and 
therefore an increase in the controlling voltage applied to the line scan oscillator. 



3 - 30 [§18 

Generating the Control Voltage in the Coincidence Detector (continued) 

You know that an increase in the control potential causes an increase in the fre- 
quency of the oscillator. This, of course, brings about a greater degree of coincidence 
between the Gi and G 2 pulses, which in turn causes an increase in the area of the 
coincidence pulses at the anode of the coincidence detector. The circuit is trying to 
restore the control potential to what it was before the change in line sync frequency; 
and eventually, the frequency of the line scan oscillator is adjusted to match the higher 
frequency of the line sync pulses. The control circuit then becomes quiescent. 

The same thing happens, save that the control potential is decreased instead of 
increased, when either the frequency of the line sync pulses is decreased or the fre- 
quency of the line scan oscillator is increased. 

The time constant of the integrator circuit (Ci x R x ) is chosen so that it is only 
capable of responding to slow changes in the shape of the coincidence pulses. It is 
this deliberate sluggishness of response which makes the circuit insensitive to sudden 
bursts of noise or interference pulses, or to short-term variations in frequency. 

The value of the time constant is fairly critical. If it is made too long, the con- 
trolling voltage will be unable to react sufficiently quickly to genuine changes in the 
frequencies of the line scan oscillator or of the line sync pulses — such as may occur, for 
instance, when changing channels. If made too short, the circuit will respond too 
readily to interference pulses, and line tearing will not be prevented. 

The Coincidence Detector — Full Circuit Diagram 

The illustration opposite shows a coincidence circuit which has found wide accep- 
tance in lower-priced TV receivers. It lacks high sensitivity and to that extent is less 
effective in operation than are the more complex circuits found in more expensive 
receivers. The circuit employs a double (triode-pentode) valve, of which the triode 
section functions as an inverter-amplifier and pulse shaper; and the pentode section as 
the coincidence detector. 

Negative-going line sync pulses from the anode of the sync separator stage are 
applied to the grid of the triode through a differentiating circuit formed by Q and the 
parallel combination of R x and R 2 . This circuit produces a pair of negative and 
positive spikes for each line sync pulse, corresponding to the leading and trailing edges 
of the pulse respectively. 

The resistors Ri and R 2 form a potential divider across the HT supply, and bias the 
triode so that its anode current flow is near saturation. Anode voltage is thus nearly 
"bottomed", as low as it will fall. The positive spikes applied to the grid therefore 
cause little increase in the already high anode current of the triode, and equally little 
reduction in the already-low anode voltage. The negative spikes, on the other hand, 
drive the grid of the valve beyond cut-off, with the result that anode current is momen- 
tarily reduced to zero and anode voltage is allowed to rise to the value of the HT 
supply. Anode voltage remains at this value until the negative voltage of the spike 
rises to within the grid base of the valve — whereupon anode current once more flows 
and anode voltage starts to fall. 

The value of anode current, and therefore of anode voltage, varies according to the 
shape of the trailing edge of the negative spike at the grid. Since this is basically 
exponential, the fall of anode voltage will be exponential also. So every sync pulse 
applied to the triode produces a single flat-topped positive-going pulse at the anode, 
this pulse having a sharp leading edge but a sloping trailing edge such as you saw in 
the illustration two pages back. 



§18] 



3.31 



The Coincidence Detector— Full Circuit Diagram 

The positive-going pulses produced at the anode of the triode are applied to the 
screen grid of the pentode via the coupling capacitor C 3 . At the same time, positive- 
going rectangular pulses from the line scan generator are applied to the control grid 
of the pentode via the long time-constant formed by Q and R 8 . (The purpose of R 7 
is to reduce the effect of amplitude variations in this waveform.) 

The pentode thus receives input signals from two sources, and the magnitude and 
duration of its anode current pulses will vary according to the degree of coincidence 
between these two signals. 

The coincidence pulses produced at the anode of the pentode are smoothed by the 
integration circuit formed by C 5 and R 5 , and a steady positive control potential (for a 
given degree of coincidence) is developed across C 5 . 

The components R 9 , C 8 connected across the control potential line provide low- 
frequency damping of the circuit, thereby preventing "hunting" by the frequency of the 
line scan generator. Without them, the frequency of the line scan generator could 
swing beyond that set by a change in control potential, with the result that it might 
never quite settle down to its correct value. 



HT<+) 




Input Pulses 
from Sync Pulse 
Separator 



*-(+) 

Control Voltage 
to Line -Scan 

R9 Generator 

i 8K 



Input Pulses from 
Line-Scan Generator 



T*tfS«» 











3.32 

The Phase Detector 



[§18 



The second main type of frequency comparison circuit is the phase detector. Its 
full circuit diagram is illustrated below. 



HT(+) 



:juul 



:innr*|L_- 



Line Sync Pulses from 
Sync Pulse Separator 



:mnr 




Control Voltage to 
Line Scan Generator 



C4.O05 

From Line Scan 
Generator 



C6 
*04 



Full Circuit Diagram of the ^SK : M. 



Negative-going sync pulses from the sync separator stage are applied to the grid of 
Vl This valve has two load resistors of equal value, one (R 4 ) in its cathode and the 
other (R 2 ) in its anode. R 3 is a cathode biasing resistor, and Ri the grid resistor. 
Since the anode and cathode potentials move in opposite directions when a signal is 
applied to the grid, sync pulses of equal value but opposite polarity will be produced 
across the anode and cathode load resistors whenever sync pulses are applied to the 
grid. The valve therefore functions as a phase splitter, in the same way as does a 
transformer having a centre-tapped secondary winding. Indeed, a transformer is 
often used in this role instead of a valve or its equivalent transistor. 

The anti-phase sync pulses from the phase splitter are applied via the capacitors C 2 
and C 3 to the opposite ends of two series-connected semiconductor diodes (Dj and 
D 2 ). The positive sync pulses are applied to the anode of Dj and the negative pulses 
to the cathode of D 2 ; both diodes are therefore forward-biased by the sync pulses, 
and are free to conduct. But a sawtooth waveform from the line scan generator is 
also applied to the anode-cathode junction of the two diodes, via the CR coupling 
circuit C 4 -R B ; and it is the time-phase relationship between this waveform and the 
anti-phase sync pulses which controls the conduction period of the diodes. 

The coupling circuit C 4 -R 5 gives the sawtooth waveform equal positive and negative 
areas above and below a mean base-line value of zero — because C 4 cannot pass a 
direct current. This means that the anode-cathode junction of the two diodes will be 
alternatively positive and negative. 

The time-phase relationship between the sawtooth and the anti-phase sync pulses is 
made such that, when the circuit is functioning correctly, the sync pulses occur at the 
point of time coincident with the centre point of the flyback period, and straddling the 
zero-voltage base line. This condition will now be considered. 



§18] 



3.33 



The Phase Detector (continued) 

The illustration shows the time-phase relationship which exists between the line 
sync pulses and the sawtooth waveform from the line scan generator in the normal 
operating condition when the line scan generator is running at the correct frequency. 



. Flyback 
Period 



Line Sawtooth Waveform applied to 
D1 and 02 Junction 




At the instant when the leading edge of a particular sync pulse occurs, the flyback 
period of the sawtooth waveform is of negative polarity, rising towards zero (and 
beyond it). The junction of D x and D 2 in the circuit opposite follows this polarity, 
and at the instant considered is also negative. D x is therefore forward-biased by the 
negative potential on its "cathode" and the positive potential (derived from the sync 
pulse at the anode of V^ on its "anode". D 2 is reverse-biased by the sawtooth poten- 
tial, and is therefore not conducting. With D x conducting, C 2 charges up through 
R 2 , D x and R 5 , its right-hand electrode thus acquiring a negative charge of a value 
governed by the voltage present at the junction Di-Da. 

When the flyback period of the sawtooth waveform passes through zero, the junction 
Di-D 2 takes up a positive potential, rising towards the maximum value of the saw- 
tooth. D 2 now becomes forward-biased by the positive potential on its "anode" and 
the negative potential (derived from the sync pulse at V x cathode) on its "cathode". 
D 2 therefore conducts and allows C 3 to charge through R 4 , D 2 and R 5 until its right- 
hand electrode acquires a negative charge equal in magnitude, but opposite in polarity, 
to that already possessed by C 2 . During this time D 1; reverse-biased, is not conducting. 

At the end of the duration of the sync pulse (which comes while the sawtooth flyback 
is still increasing positively), both diodes are non-conductive. They remain in this 
condition during the scan period of the sawtooth until the next sync pulse arrives — 
whereupon the whole process is repeated, and C 2 and C 3 replenish the charge which 
leaked away during the scan period. 

This leakage of charge is quite small and takes place through the two equal-value 
resistors (R 6 , R 7 ) connected in series across the two diodes. Since the remote ends 
of these two resistors are connected to the right-hand electrodes of C 2 and C 3 res- 
pectively, the potentials at these ends will be opposite but equal. The polarity of the 
voltage at their centre-point connection will therefore be fluctuating about zero. By 
connecting an integrating circuit (R e , C 5 ) to this point, the fluctuations can be 
smoothed out, and a steady d.c. level (in this case zero) obtained which is used as 
the control potential for the frequency of the line scan generator. 

The time constant of the integration circuit R 8 -C 5 is chosen so that it will respond 
to gradual changes appearing at the junction of R 6 and R 7 , but will ignore such short- 
term variations as may occur between individual sync pulses. C 6 and R 9 are the 
normal damping components used to prevent "hunting" by the frequency of the line 
scan generator. 



3.34 



[§18 



The Phase Detector {continued) 

Should the frequency of the line scan generator start to rise above that of the 
incoming line sync pulses, the commencement of its flyback period relative to the 
commencement of the line sync pulses will appear to occur earlier. Conversely, should 
the frequency of the generator be reduced, its flyback period will appear to occur 
relatively later. The illustration shows the time-phase relationship between the line 
sync pulses and the line scan generator sawtooth for three operating frequencies of 
the generator: A = normal, B= higher than normal, and C = less than normal. 




Line Sync Pulses 
at steady frequency 



The PiMSi MSOUmma*: Control Voltages 

Consider Situation B. With the flyback period of the sawtooth waveform oc- 
curring earlier than normal, the line sync pulses arrive at a time when the flyback is 
passing through its positive region. Consequently, instead of T> 1 and D 2 conducting 
in turn, only D 2 will now conduct because only it is forward-biased by the flyback 
voltage appearing at the junction of the two diodes. Only C 3 is now enabled to 
charge up — and it does so this time to a higher positive voltage than normal because 
the conduction period of D 2 occurs at a higher point on the flyback waveform. 

With Dj not conducting, C 2 cannot replenish the charge it lost during the preceding 
scan period of the sawtooth waveform and would, if something did not happen to 
reverse the process, lose even more of its charge on successive scan periods. What 
happens to prevent this is that the average value of the voltage appearing at the 
junction R 6 -R 7 , and therefore across the integrating capacitor C 5 , increases above its 
normal value of zero and is used (in a way you will shortly see) to correct the frequency 
of the line scan generator by slowing it down to its normal value. As this happens, 
the conduction periods of both diodes gradually become equal again, and the charges 
acquired by C 2 and C 3 result in a mean control voltage of zero across C 5 . 

A similar sequence of events takes place when the frequency of the line scan genera- 
tor falls below normal (Situation C). The commencement of the flyback period now 
appears to occur later than usual, so that the line sync pulses occur during the period 
when the flyback is wholly negative. Because of this, only D x can conduct during the 
sync pulse periods and only C 2 is able to replenish (and more than replenish) the 
charge it lost during the preceding scan periods. A negative control voltage is 
quickly built up across the integrating capacitor C B and is used to increase the fre- 
quency of the line scan generator until it is again in step with that of the line sync 
pulses. 



§18] 



3.35 



The Phase Detector (continued) 

You have just seen that the controlling voltage developed across the integrating 
capacitor C 5 is normally zero but that it swings positive when the frequency of the line 
scan generator increases above normal and negative when the frequency falls below 
normal. You know (from the description of the coincidence detector and from earlier 
discussions on the basic multivibrator) that this is exactly the opposite of the polarities 
required. Moreover, the swinging of the polarities positive and negative about zero 
(chassis potential) is quite different from the unidirectional (either wholly positive or 
wholly negative) polarity actually required. 

Both of these difficulties are overcome by connecting into the circuit one more triode 
in such a way that it functions both as a d.c. amplifier (thereby increasing the sensi- 
tivity of the detector itself) and as a signal inverter which also shifts the average d.c. 
level. 

HT(+) 



Positive Control Voltage 
to Line Scan Generator 



Input from C5 
in Phase Detector 
Circuit 




•;!llff|i 

JRI0DE*m 




The triode has an undecoupled cathode-bias resistor (R K ) whose purpose is to 
improve the signal-handling capability of the valve and to reduce distortion. The 
signal from the integrating capacitor C 5 is applied to the grid of the valve, and the 
output is taken from its anode. Under normal operating conditions, the input signal 
is at zero potential, and the valve passes an anode current the magnitude of which is 
determined by the value of the cathode bias. A steady voltage drop across the anode 
load resistor (R L ) results, and a positive control voltage is applied to the line scan 
generator. 

Should the frequency of the line scan generator increase, the input signal will 
become positive and even more anode current will flow. An even greater voltage 
drop across R L will cause a less positive controlling voltage to be applied to the line 
scan generator, so reducing its frequency as required. If, on the other hand, the 
frequency of the line scan generator falls, the input signal to the valve will also fall. 
Less anode current will flow through R L ; a more positive controlling voltage will be 
applied to the line scan generator, and its frequency will correspondingly increase. 

Note that the controlling voltage, although varying in magnitude, remains wholly 
positive at all times, and that it now always moves in the right direction to control the 
frequency of the line scan generator. 



3.36 [§ l 8 

Pull-in Time 

Whatever the nature of the detector circuit used to control their frequency of opera- 
tion, all timebase circuits require time to settle down to correct synchronisation after 
signals of a different frequency have been applied to them — as, for instance, when you 
switch channels on your TV receiver, or when you switch it on "from cold". The 
length of this pull-in time, as it is called, is governed by the sensitivity of the phase 
detection circuit (the amplitude of the control voltage produced for every degree of 
phase difference) and by the length of the time-constant of the integration circuit from 
which the control potential is derived. 

Ideally, pull-in time should be as short as possible so that corrections are made to 
the frequency of the line scan generator as soon as a genuine and sustained difference 
exists between its frequency and that of the incoming line sync pulses. Unfortunately, 
however, as pull-in time is made shorter, so the circuit becomes progressively more 
susceptible to the effects of impulsive interference. 

A problem also arises during the period of the field sync pulse when the flywheel 
circuit receives pulses recurring at twice-normal line pulse frequency (the half-line 
pulses) ; for these can upset the operational balance of the circuit. Given a very short 
pull-in time, however, the circuit will correct itself just as quickly as it became upset as 
soon as the normal line sync pulses are restored at the end of the field pulse. 

If pull-in time is made longer than the duration of the field sync pulse, of course, no 
problem will arise; for the circuit will be too sluggish in the first place to respond to 
the "error" presented by the appearance of the half-line pulses. So a long pull-in 
time is required to make the flywheel circuit insensitive to impulsive interference, and a 
short pull-in time is required to ensure r,apid correction to the frequency of the line 
scan generator when genuine frequency differences arise. 

The length of pull-in time actually built into a given receiver is (as so often in TV 
design) a matter of compromise. A typical length would be the duration of 20 to 
200 lines, though it may be much longer than that in some makes of receiver. 

The frequency range within which the flywheel circuit is capable of controlling the 
line scan generator is called the pull-in range of the circuit. . A typical operating range, 
for either line system, would be approximately ± 1 % about the normal operating 
frequency of the system. 



Some Drawbacks of the Flywheel Sync System 

Though the introduction of flywheel sync has undoubtedly improved the overall 
picture quality of the modern TV receiver, particularly in regions of fringe-area 
reception, its complexity has certainly added to the cost of the set. And as a large 
proportion of all TV receivers are used in regions close to the transmitter where 
reception is generally good and signal-to-noise ratio high, it is not universally accepted 
in the industry that the extra performance is worth having at the price. No doubt the 
solution to this particular dilemma will vary in different parts of the world according 
to whether a critical proportion of potential customers live in areas of generally good, 
bad or "middling" reception. 



§18] 



3.37 



Some Drawbacks of the Flywheel Sync System (continued) 

Two particular faults to which some types of flywheel sync circuit are prone should 
be mentioned. The first of them manifests itself during switch-on. If for some 
reason (generally mis-adjustment of the line-hold control) the frequency of the line 
scan generator is considerably different from that of the incoming line sync pulses, it 
can happen that the flywheel circuit will be unable to cope with the difference at all. 
In such circumstances the viewer must alter the frequency of the line scan generator 
himself by manipulation of the line-hold control, so as to bring the frequency of the 
generator sufficiently close to that of the line sync pulses for the control voltage to be 
able to take over and lock the picture by eliminating all frequency difference. 

The other problem arises when a picture appears on the picture tube which is 
perfectly stable but which is displaced horizontally by up to one half-line across the 
tube. The result is two half-pictures side by side, with the blanking interval positioned 
between them near the centre of the screen instead of at its ends. 




An Effect of 



PHASi 
MISALIGNMENT 



in some 



Flywheel Sync Circuits 



Such a picture can occur in a flywheel sync circuit which compares the frequency of 
the line scan generator with that of the line sync pulses directly, rather than indirectly 
by measuring their phase relationship. The flywheel circuit is maintaining the 
frequency of the line scan generator exactly in step with that of the line sync pulses, 
but the two are out of phase. In other words, the line sync pulses are occurring, in 
time, halfway along the scan period of the line scan waveform. 

The fault is usually caused by incorrect adjustment of the line-hold control, or by 
incorrect setting of an internal line-phase control. (Indeed, correct initial setting of 
the line-hold control is considerably more critical in flywheel sync circuits than it is in 
circuits employing direct synchronisation.) 

Once correctly set, however, the line-hold control can be adjusted over a con- 
siderable part of its range without upsetting synchronisation, functioning rather like a 
horizontal shift control by means of which the picture can be made to move bodily 
from left to right, or vice versa. 

What happens is that the control, being unable to change the frequency of the line 
scan generator because every adjustment is counteracted by a corresponding change in 
the control voltage from the flywheel circuit, alters instead the phase of the waveform 
it produces. The effect is obviously to shift the picture bodily towards one side or 
other of the screen of the picture tube. 



3.38 

REVIEW of the Scanning Generators 

The line and field scanning generators 
produce waveforms which, after subsequent 
shaping and amplification, are used to con- 
trol the scanning currents in the line and 
field scanning coils. These coils are thereby 
caused to produce the scanning raster on the 
screen of the picture tube. 

The operating frequencies of the two 
generators are synchronised with those of 
corresponding circuits in the studio camera 
by pulses extracted from the video signal in 
sync pulse separator stage. 



[§18 





Lint- Hold 

(or Fltld-Hold] 

Control 



C1.R3<C2.R2 



The Asymmetric 

Multivibrator 



The multivibrator makes an inexpensive 
waveform generator much used in both line 
and field scan circuits. Its operating fre- 
quency is determined by the time constants 
of two sets of cross-coupled capacitors and 
their associated grid resistors. 

In many TV sets, the operating frequency 
is controlled by varying the value of one of 
the two grid resistors. A control of this 
kind is called a line hold control if the 
circuit is being used to generate line-scan 
waveforms, and a field hold control if it is 
being used for field-scan waveform genera- 
tion. 



Flywheel synchronisation is a technique 
employed to keep the line-scan synchronisa- 
tion consistently stable despite the presence 
of interfering noise pulses. In the absence 
of such a technique, the picture could be 
subjected to line tearing. 




§18] 

REVIEW of the Scanning Generators {continued) 



3.39 



The coincidence detector is a frequency- 
comparison circuit much used in flywheel 
synchronisation. It produces a surge of 
anode current flow when the frequencies of 
the line scan oscillator and of the line sync 
pulses extracted from the incoming video 
signal coincide, and flows of lesser and lesser 
value as the two waveforms move further 
apart in phase. The coincidence pulses 
produced as a result of these changing anode 
current flows are used to control the frequen- 
cy of the line scan oscillator. 





:tmit 



Full Circuit Diagram of the 



The phase detector is a widely used 
alternative to the coincidence detector. It 
compares the phase of the pulses produced 
by the line scan oscillator with that of the 
line sync pulses extracted from the incoming 
video signal, and produces an output voltage 
proportional to the difference. This out- 
put pulse is used to control the frequency of 
the line scan oscillator. 



3.40 



§19: DEVELOPING THE FIELD SCAN 



Your next job is to see how the line and field scan waveforms are developed until 
they are large enough and of the correct shape to activate the line and field scanning 
coils clustered round the neck of the picture tube. It is best to begin with the field 
scan waveform, since it is considerably the simpler of the two. 

First, a word about the electrical characteristics of the field scanning coils, because 
these characteristics largely determine the shape of the current waveforms which need 
to be applied to the coils to produce the shape of scan required across the picture tube. 
(For reasons which you will see in a moment, this waveform is not the perfectly linear 
one you would expect, though it is very close to it.) 

The field scanning coils of a modern TV receiver consist of two identical coils wound 
into special shapes and positioned 180° apart (i.e., directly opposite one another) 
round the neck of the picture tube. The coils are electrically connected in series and 
have a combined inductance of about 90 mH. 

The combined resistance of the coils (caused by the resistance of the fine copper 
wire from which they are made) is typically about 40 ohms. The coils are supplied 
from the field output stage with a scanning waveform which has, in all British and 
European TV systems, a repetition rate of 50 Hz (the field frequency). 

You know from Basic Electricity, page 3.57, that the inductive reactance, in ohms, 
of a pair of coils connected in series is given by the equation X u =2wfL, where/is the 
frequency in Hz and L the inductance in henries. Substituting 0.09 (=90 mH) for L 
and 50 for/, you get 

X h = 2x3-1416x50x0-09 = 28 ohms (approx.) 

Thus the inductive reactance of a typical field scan coil assembly is seen to be rather 
less than is its resistance. In other words, the coil assembly is predominantly resistive, 
although possessed of a substantial inductive reactance. 

Shown below is a bird's-eye view of a basic field scan coil assembly, with its electri- 
cally equivalent circuit on the right. 



Scanning 
Current 




Picture 
Tube 



Scanning 
Current 



g X L * 28<x 




R 
40.a 



The F EO SCAN Coils 



Equivalent Circuit 



§"»] 



3.41 



~^§J 




The Shape of the Field Scan Waveform 

You read on the last page that the shape of the current waveform which needs to be 
applied to the field scan coils to produce a satisfactory scan down the picture tube is 
nearly, but not quite, linear. 

In the earlier TV tubes whose screens 
were only about 13" wide and markedly ^ ^^^ 

convex (see diagram opposite), the length 
of the scanning beam measured from the 
centre of the scanning coil assembly was 

nearly the same whether the beam was at the beginning, middle or end of its scan 
(Li = L 2 = L 3 ). Given a linearly increasing angular deflection of the beam, therefore, 
the rate at which the beam moved down the face of the tube was pretty well constant. 

The requirement, therefore, was for the magnetic field set up by a pair of field scan 
coils to be linearly related to the magnitude of the scanning current flowing through 
them— so that a doubling, for example, of the scanning current would double the 
strength of the magnetic field, which in turn would double the distance travelled by the 
scanning spot down the face of the picture tube. 

The modern picture tube, however, is nearly flat-faced and has a much wider 
scanning angle— typically 110°. The illustration below shows the sort of difficulty 
which this introduces. 




Scanning / / 

Coils / /L 2 







Correction 



Scanning 
Beam 



The distance travelled by the scanning spot in traversing half the face of the picture 
tube is shown as d. Li is the length of the scanning beam when the spot is moving 
down the centre of the tube face; L 2 and L 3 its length when the spot is at the beginning 
and end of the scan respectively. <t> is the scanning angle of the tube. Clearly, both 
L 2 and L 3 are greater than Li, which means that the rate at which the spot would 
move down the face of the tube, given a linearly increasing angular deflection of the 
beam, would be greater at either extremity of the tube face than it would be in the 
middle. 

In other words, the rate at which the spot moves down the face of the tube would be 
"quick-slow-slow-quick". The result would be that the picture would appear to the 
viewer to be stretched-out for a few lines at the top and bottom of the picture. Heads 
would appear egg-shaped, and feet and ankles too long for the legs to which they 
belonged — rather as in those convex mirrors which face you in the Tunnel of Horrors in 
seaside amusement arcades. There would also be some decrease in brilliance at these 
extremities. 



3.42 



[§19 



The Shape of the Field Scan Waveform (continued) 

The method adopted to compensate for this effect is to introduce some small 
degree of curvature into each end of the scanning waveform in order to lessen the 
velocity of the scanning spot at the beginning and end of the scan. As the scanning 
waveform increases at a less-than-linear rate at the beginning of the scan, so the 
intensity of the magnetic field produced by the scanning coils at that point increases 
at a less-than-linear rate also. For the great bulk of the scan period, the rate of 
increase in current flow through the coils then becomes linear, but it falls off again to 
less-than-linear as the scan nears the bottom of the screen. 

The shape of the field scan waveform required in the modern picture tube thus 
resembles a much elongated capital letter "S", tilted over at an angle of 45° from the 
vertical and with all its curves sharply flattened out. The technique is for this reason 
known as S-Correction of the field scan waveform. It is illustrated below. 



S- Correction 

in the 
FHU> SCM WM&OXM 







J 


c 

0) 




A 






/' Linear 


O 






CD 


















c 






<J 






w 







Time 



You will see in the next Section that S-correction of the waveform is also needed 
in the line scan, for exactly the same reasons. 

The methods used to introduce S-correction vary considerably from receiver to 
receiver. In the field scan it is generally applied in two stages. The first stage, which 
affects the initial part of the scanning waveform, relies on the inherent curvature of 
the IJV g characteristic of the valve used in the field output stage when it is connected 
to the rather special anode load which you will read about on the next page. 

The second stage of correction is applied mostly to the final part of the scanning 
waveform, and is introduced by a special RC network forming part of the Vertical 
Linearity control, which will also be explained later. 

Note that the degree of S-correction needed for the field scan is somewhat less than 
is that required for the line scan. This is because the width of the screen is, as you 
know, one-and-a-third times greater than its height (you will recall that the aspect 
ratio of the picture is 4:3). 

The Field Output Stage 

This consists of two elements: — a power amplifier valve delivering some 3 or 4 
watts of power to the field coils, and an impedance-matching transformer connected as 
the anode load of the valve and serving to couple it to the coils. Both the valve and 
transformer operate very much as do the PA valve in a radio set and the output 
transformer which matches it to the loudspeaker (see Basic Electronics, page 2.73). 

The essential features of the field output 
stage are shown in the illustration on the 
next page. Pictured opposite is the shape 
of current waveform needed at the anode 
of the valve to produce the S-corrected 
waveforms in the field scanning coils. 




§l»] 



3.43 



The Field Output Stage (continued) 

The illustration shows, in outline form, the essential features of the field output 
stage, together with the voltage waveform received as input from the field scan 
generator and the current waveform required to be delivered as output to the field 
scanning coils. 

HT{+) 



Input 
Scanning 
Voltage 



Picture 
Tube 




You have seen that the field scan coils are predominantly resistive but also have 
substantial inductive reactance. To drive a linearly rising current through such coils 
requires an applied voltage which contains (a) a linearly rising component to overcome 
the resistance of the coils, and (b) a rectangular component to overcome their inductive 
reactance. Such a composite waveform is a step sawtooth of trapezoidal shape such 
as was explained in greater detail on pages 2.60 to 2.63 of Basic Electronic Circuits 
(which you would do well to re-read at this point). In the case of the field scan coils 
of a TV set, as you also know, the step sawtooth requires to be further modified to 
allow for S-correction of the field scan current waveform. 

The required waveform must be derived, of course, from the secondary of the output 
transformer. This winding, too, will possess both resistance and inductive reactance 
— as will also its primary, which receives its input from the anode of the PA valve. 
Thus this valve, "looking into" the primary of the output transformer, sees as its 
anode load an impedance composed of certain values of resistance and inductive 
reactance derived from the characteristics of two stages — the transformer itself and the 
load connected to its secondary (i.e., the scanning coils). Given the types of scanning 
coil and output transformer used in a modern TV receiver having a wide-angle picture 
tube, the effective anode load of the PA valve possesses an impedance such that the 
waveform of current flowing in the primary of the output transformer — which is also 
the anode current of the valve — must have a shape like that pictured at the foot of the 
last page. 

This shape is called parabolic, and the waveform is said to have zero initial slope. 
This zero initial slope assists in providing the required S-correction for the beginning 
of the scan. You will shortly see how S-correction for the end of the scan is provided 
by the Vertical Linearity control circuit which alters the shape of the input voltage 
waveform applied to the grid of the valve. 



3.44 [§|9 

The Field Output Stage (continued) 

The full circuit diagram of a practical field output stage is shown in the illustration 
opposite. 

The grid bias of the output valve (V^ is determined by the cathode bias components 
C 3 and R 3 , whose values are carefully chosen to give the correct operating point and to 
ensure the correct shape of anode current waveform. The peak-to-peak value of 
anode current required for the full vertical scan of a 19" picture tube is about 100 mA, 
which corresponds to a peak-to-peak current of some 500 mA flowing through the 
field scan coils. 

The output transformer (T x ) has a primary-to-secondary turns ratio of about 5:1, 
and is connected to the field scan coils through the thermistor R x . A thermistor is a 
temperature-sensitive resistor made from a carbon composition whose resistance 
becomes less as its temperature increases. It is thus said to have a negative tem- 
perature coefficient of resistance. 

The resistance of R x at normal room temperature (about 25°C) is 8 ohms, and the 
resistance of the field coils at the same temperature is, as you know, some 40 ohmns. 
Since the resistance of the Secondary winding of the output transformer is about 15 
ohms, the total resistance in the secondary circuit, at 25°C, is about 8+40+15 = 63 
ohms. 

When the TV receiver has been running for some time, the temperature inside the 
cabinet will have risen considerably and will have affected the resistance of the 
transformer secondary and of the scanning coils. Since these are both wound from 
copper wire which has a positive temperature coefficient of resistance, their individual 
resistances increase as the set warms up. This increase in resistance, unless counter- 
acted, would cause a reduction in the magnitude of the scanning current and a 
shrinkage in picture size. But the thermistor is also affected by heat — only as the 
temperature rises, its resistance decreases. So if its value is chosen to be such that its 
resistance decreases by approximately the same amount (within a limited temperature 
range) as the resistance of the other two components rises, the overall resistance of the 
secondary circuit will be maintained constant at about 63 ohms. 

The incoming signal from the field scan generator (which you know to be of 
exponential sawtooth shape) is coupled to the grid of the output valve via C x , and is 
developed across the grid leak resistor R x . After amplification, it appears in the 
normal way at the anode of the valve, where it is developed across the primary of the 
output transformer and applied to the scanning coils. 

Vertical Linearity and S-Correction 

The waveform at the anode of the PA valve also appears across C 4 , R 4 and RV 2 . 
Together with R 2 , RV^ C 2 and R lf these components form the Vertical Linearity 
control shown in block outline in the illustration on the last page. C 4 , R 4 and the 
adjustable RV 2 enable the shape of the waveform developed across Ri to be varied by 
filtering out some of the harmonic frequencies contained in the waveform at the anode. 
The filtered waveform is then fed back to the grid of the valve, in opposite phase to the 
incoming signal, via R 2 , RV X and C 2 . 

The actual amount of current so fed back is controlled by the setting of RV X (which 
also influences the shape of the waveform). This method of linearity correction 
operates by feeding back a negative waveform such that the shape of the input wave- 
form appearing at the grid of the valve (i.e., across R x ) is distorted so as to produce a 
waveform of current in the field scan coils which possesses the required amount of 
S-correction over the last 30 lines or so of the scan. 



§19] 

The Field Output Stage (continued) 



3.45 







< 
O 
CO 



&0Q 



§&: 



IMl 



M 



mm 






3.46 



[§l» 



The Field Output Stage (continued) 

In a practical receiver, RV 2 would be a viewer-operated control labelled Vertical 
Linearity on the outside of the set. It would be used to correct the picture if the 
leading lady started to develop that Hall-of-Mirrors effect in head or feet! RVi 
would be mounted within the set and normally preset at the factory so as to produce, 
in conjunction with RV 2 , the required linear scan. It should require further adjust- 
ment only to compensate for ageing of the output valve. 

The components R 5 , C B and C 6 form a flyback-suppression circuit which helps to 
prevent the field flyback lines from becoming visible on the picture tube when the 
viewer adjusts his Brilliance control for a brighter-than-usual picture. (You saw on 
page 1.74 the type of line pattern you might otherwise get superimposed on the normal 
picture.) R B and C 5 form a potential divider across the secondary of the output 
transformer, and C 6 couples the voltage developed across C 5 to the grid electrode of 
the picture tube. The reactance of C 6 to the comparatively slow rise of the scanning 
waveform is high, so little voltage is fed to the picture tube during this period. At the 
end of the scan, however, the current in the field coils is rapidly reduced. The resis- 
tance of C 6 to the sharpsided flyback waveform appearing across C 5 is very low, and a 
large negative pulse is applied to the grid of the picture tube sufficient to black-out the 
scanning beam during the flyback period. 

It should be noted that not all receivers employ this type of flyback suppression 
circuit. Some derive the necessary negative pulses from the scanning generator 
itself. 

The two resistors R 6 , R 7 shown connected across the field scan coils belong more 
logically to the picture tube stage; but because their equivalent components in the line 
scan circuitry play an important part in the line output stage, it will be convenient to 
cover their function and mode of operation at this point. 

The purpose of R 6 and R 7 is to damp down the ringing oscillation which would 
otherwise occur across the coils every time the current flowing through them is rapidly 
reversed during the flyback period. If this ringing were not suppressed, there could be 
distortion of the picture at the start of every new field — especially if the ringing per- 
sisted after the cessation of the flyback period. 

.Flyback g ^Flyback 





UNDAMPED RINGING 

(No Damping Resistors) 



DAMPED RINGING 

(With Damping Resistors) 



Another dangerous possibility if ringing were not suppressed would be the injection 
of an interference signal into the line scan coils, which are mounted in very close 
proximity to the field coils. 

In addition to damping down ringing oscillations, R 6 and R 7 serve also to absorb any 
unwanted pulses which could be injected into the field scan coils from the line coils 
close beside them. This mutual interference between the line and field scanning coils 
is called cross-talk. It is caused by the stray capacitances and mutual inductance 
which exist between the two sets of coils. Its effect on the picture is apt to be a fixed 
pattern of wavy striations running horizontally across the whole screen. 



§l»] 

REVIEW of Field Scan Development 



3.47 



The two scanning coils used for producing 
the field scan are predominantly resistive in 
nature, though they also possess substantial 
inductive reactance. The coils are usually 
connected in series with one another. They 
form part of the line and field scanning coil 
assembly situated around the neck of the 
picture tube. 



ft 



n.nBD8GM 



90mH 5? X L * 28A 



Mquhnlmnt Gkvutt 




£Htto 



iflwP^^'Wl 



The shape of the scanning current wave- 
form supplied to the scanning coils from the 
field output stage resembles an elongated 
and much flattened letter "S". Such a 
shape of current waveform helps to com- 
pensate for the differences in distance which 
the beam has to travel when scanning the 
centre, and the top and bottom areas res- 
pectively, of the screen. This does much to 
ensure a properly proportioned picture. 



Variations in the resistance of the Scan- 
ning coils are temperature-compensated 
with the aid of a series-connected thermistor, 
selected to have a temperature coefficient of 
resistance which matches that of the scanning 
coils, but is of opposite sign. 

The compensation provided in this way 
prevents the height of the picture from being 
affected by changes in temperature which 
occur within the receiver. 




Field Scan 
Coils 

R7 



Picture Tube 



3.48 



§20: DEVELOPING THE LINE SCAN 



The function and basic layout of the line output stage is similar to that of the field 
output stage. In its simplest form, it too consists of a power amplifier valve with an 
impedance-matching output transformer in its anode circuit coupling the valve to the 
line scan coils. 

In practice, however, the line output stage is also required to fulfil two other impor- 
tant tasks. One of these is to produce a "boost" voltage of some 500 to 800 V to be 
used by the line output valve for its own HT supply— and also (as you will see) by the 
first-anode electrode of the picture tube. The second task is to produce an EHT 
voltage supply as high as (typically) 18 kV for the picture tube. Not unnaturally, 
these two high-voltage requirements introduce complications into the basic circuit of 
the line output stage; but it is still worth while looking at the latter in outline form as a 
first step. 



Scan 



Flyback 



HT(+) 

500 V 



f ^* ^i -\ Line Scan 
£ is' [sj Current 




Line Scan 
Coils 



^uH§m^T^^mmf^^0M 



w&m 



mm 



imMMM-^m^^jmmsm^ 



The valve is supplied from the line scan generator with a sawtooth voltage which 
causes a current of similar shape to flow through the primary of the matching trans- 
former, connected as the anode load of the valve. The secondary of the transformer 
couples the valve to the scanning coils so that, during the "scan" period of the input 
waveform, the beam of the picture tube carries out a linear scan of the screen. At 
the end of the scan period of the input voltage, anode current in the valve is reduced 
to a minimum, the scanning current in the coils is reversed, and the scanning beam is 
rapidly returned to the left-hand side of the picture tube. 

Note that at all times the magnitude of the anode current flowing in the valve, and 
therefore of the scanning current in the coils, is under the control of the input voltage 
sawtooth, exactly as were the corresponding current flows in the field output stage. 



§20] 



3.49 



The Line Scan Coils 

The line scan coils are essentially similar to the field scan coils, though they are 
arranged to have considerably different values of combined inductance and resistance. 
As you know, the two pairs of coils are clamped together to form a single assembly 
round the neck of the tube. If they could be viewed through the front of the picture 
tube, the line coils would be above and below the neck of the tube, with the field coils 
on either side. 

The line coils may be connected either in series or in parallel, but present-day practice 
tends to favour the parallel connection because it demands fewer associated compo- 
nents. 

The combined inductance of a pair of line scan coils is generally made much lower 
than that of a pair of field coils in order to reduce the ringing oscillations which arise 
in the line coils (and, as you will see, in some other coils in the output stage also) when 
the onset of line flyback sets up large back-e.m.f.'s across the coils. Because of the 
very short line flyback period, such oscillations would, if left to themselves, have 
insufficient time to die away before the initiation of the next line scan. They would 
resonate with the self-capacitance of the scanning coils, and would produce a series of 
alternately bright and dark horizontal striations down the left-hand side of the picture. 

Typical values of combined inductance and resistance in a line scan coil assembly are 
therefore made as low as 3 mH and 4 ohms, respectively. You know that the scanning 
waveform has a repetition rate of 10-125 kHz (in the 405-line system) and of 15-625 
kHz (in the 625-line system). So, using again the formula X u =2irfL, you get inductive 
reactances for the coil assembly as follows: 

405-line system 2 x 3-1416 x 10-125xl0 3 x3x 10" 3 ohms 
/. X L = 200 ohms 

625-line system 2 x 3-1416 x 15-625 x 10 3 x3x 10 -3 ohms 
.-. Ai. = 300 ohms 



The inductive reactance of the line scan coils is thus about 60 times the value of 
their d.c. resistance — a far greater difference than in the field scan coils. The line 
coils are therefore predominantly inductive, and much more strongly so than the field 
coils are predominantly resistive. 



Scanning 
Current 




Picture 
Tube 
(side view) 





-. Scanning Current 


L jc 

3mH K 


► X L = 200 ii (405- Line) 

► = 300X1 (625-Line) 


R < 
4n < 




' 


' 



ii 



Equivalent Circuit 



3.50 [§20 

The Line Output Stage 

If a line scan coil assembly with parallel-connected coils, an effective inductance of 
3 mH and a d.c. resistance of 4 ohms were to be used in a basic circuit for scanning 
a modern 19" picture tube, a peak-to-peak scanning current of about 1-5 amperes 
would be required. Since the coils are fed from a transformer — which cannot transfer 
d.c. — the peak-to-peak value of the required scanning current would extend from 
-0-75 A to +0-75 A, and its average value would be zero. 

Now when a changing current is passed through any inductor, there is developed 
across that inductor a back-e.m.f. whose polarity is opposite to that of the voltage 
which is driving the current through the coils (Basic Electricity, page 5.50). The mag- 
nitude of this e.m.f. (F L ) is proportional to the value of the inductance (Z,) and to the 
rate at which the current through the coil is changing. It can be quantified by using 
the formula: — 

Fl (in volts) = L (in henries) x the rate of change of current flow 
(in amperes per second). 

You know from page 1.71 that the duration of the line period in the British 625-line 
scan waveform is about 52 jxs (after allowing some 12 [is for completion of the flyback). 
During this period, the current in the scanning coils is increasingly linearly through a 
maximum of 1 -5 amperes ; the rate of change of current flow is therefore 1 -5 -f- 52 x 10 ~ 6 
A/s. Using the formula in the last paragraph, the back-e.m.f. developed across the 
coils during the scan period is found to be: 

^ = 3xl0 " 3x 52x : i^= 86V 
Note that since the scanning current increases linearly (as it should do to produce a 
linear scan), its rate of change will be constant throughout the entire scan period — 
which means that V h will also be constant over the same period, at 86 V. 

During the flyback period, however, the back-e.m.f. developed across the coils 
becomes much greater because of the much higher rate of change of current (remember 
that the scanning current falls through 1-5 amperes in a period of only 12 (is). Using 
the formula again, you get: 

1-5 
12xl0" 6 

If the fall of current during the flyback period were strictly linear (which in practice 
is not always the case), the value of V u throughout flyback would remain constant 
at 375 V. This is shown in the illustration on the next page, which demonstrates the 
relationship between the scanning voltage waveform applied to the grid of the line 
output valve in the basic circuit and the voltage and current waveforms which sub- 
sequently appear at the coils. 

The voltage developed across the resistive component of the scanning coils (4 ohms) 
increases linearly during the scan period of the waveform, and decreases in a similar 
manner during the flyback. The magnitude of this voltage (V R ) may be determined 
at any moment during the period of the waveform by application of Ohm's law, 
multiplying the value of the scanning current at the chosen moment (/,) by the resis- 
tance of the coils. Thus, V R =I s x R. 

Since the peak-to-peak value of the scanning current is 1-5 A, the corresponding 
peak-to-peak value of V B will be 1-5 x 4= 6 V. This voltage is so small compared to 
that which is developed across the inductive component of the coils that it can for most 
practical purposes be ignored. 



F L = 3 x 10- 3 x ,„ ,„_ B = 375 V 



§20] 



3.51 



He Line Output Stage {continued) 

The illustration below pictures the relationship which exists between the waveforms 
of voltage applied to the grid of the line PA valve, and of voltage and current which 
then appear at the line scan coils. 



GRID SCAN 
VOLTAGE 



COIL SCAN 
CURRENT 



BACK-E.M.F. 
ACROSS COILS 

(Volts) 




0-75A 



0-75 A 



86V 



The back-e.m.f.'s which are developed across the scanning coils during the scan and 
flyback periods also appear, of course, across the secondary winding of the output- 
transformer which feeds them. Since the turns ratio of this transformer is typically 
4:1, any voltage appearing across the secondary will appear across the primary four 
times as great (assuming, theoretically, that both coils and transformer are free of 
both resistance and capacitance). 

Thus, if 86 V is developed across the secondary during the scan period, 4 x 86 = 344 V 
will appear across the primary during the same period. And if 375 V is developed 
across the secondary during the flyback period, 4 x 375 = 1,500 V will appear across the 
primary. This is a very considerable voltage. 

But you must also consider what happens to the voltage at the anode of the line 
output (or PA) valve. During the scan period, the valve is passing anode current, so 
its anode voltage will be equal to the HT voltage (in this case 500 V) less the reflected 
voltage appearing across the primary of the transformer which forms its anode load. 
Effective anode voltage during the scan period is therefore 500 — 344 = 1 56 V. This is 
a fairly modest value ; but during the flyback period anode current flow in the valve is 
rapidly reduced to zero and its anode voltage rises until it equals the HT voltage plus 
the reflected voltage (which is now of opposite polarity to that produced during the 
scan). Effective anode voltage is therefore 500+ 1,500 — or the even more formidable 
figure of 2 kV. 

The need to handle these high voltages clearly calls for major adaptations to the basic 
circuit of the line output stage, and helps to explain why the stage itself presents more 
problems than did the field output stage examined in the last Section. 



3.52 



[§20 



The Line Output Stage (continued) 

You were assumed to be dealing on the last page with a circuit in which resistance 
and self-capacitance in the scanning coils and in the output transformer were both 
non-existent. In practice, of course, both components always possess significant 
amounts of each, and you must now see how their presence affects the performance of 
the stage. 

Of the two characteristics, the self-capacitance is the more important, because it 
reacts with the natural inductance of the circuit to form a tuned circuit. 

At the beginning of the flyback period of the scanning waveform, anode current 
flowing in the tuned circuit from the line output valve is suddenly reduced to zero. 
This rapid change of current gives rise to the formidable back-e.m.f. already discussed, 
and it also causes the tuned circuit to resonate at its own natural frequency — which 
may be anywhere between 20 and 60 kHz depending on the physical construction of the 
transformer and the coils. This ringing, once initiated, will die away at a rate governed 
by the resistive losses of the circuit, which consist of the winding resistances of the 
transformer and coils. The lower the resistance (and therefore the greater the 
efficiency) of the circuit, the greater will be the time taken for the ringing to die away. 
In a modern TV receiver of efficient design, it would persist well beyond the flyback 
period, so impairing the start of the scan period and distorting the picture at the 
beginning of each new line. 

You will recall that when a similar situation arose in the field output stage, ringing 
was reduced to an acceptable level by the addition of damping resistors across the 
coils. In the line output stage, where very large voltages are involved, this method 
of damping is unacceptably inefficient because the resistors would absorb too much 
energy from the scanning waveform during the period of the scan itself. (You can 
get an idea of the quantity of energy wasted from the fact that a damping resistor 
dissipated more than 5 watts of power in the scanning circuits of even the small 9* 
picture tube used in very early TV receivers.) Something much better is obviously 
required for a modern receiver using 23" picture tubes. 

Some improvement in efficiency could be brought about by connecting a capacitor 
of carefully chosen value in series with the resistor. The reactance of this capacitor 
would appear large to the slowly rising scan portion of the scanning waveform, so 
reducing the amount of current absorbed by the damping resistor, but very low to the 
much faster flyback portion of the waveform. The damping resistor would therefore 
be able to exert maximum effect. 



Line Output 
Transformer 



Damping 
Components 





Line Scan 
Coils 



Though better than the single resistor, this method of damping is still not good 
enough for modern TV circuits, where something much more efficient is required. 



§20] 3-53 

The Efficiency Diode 

A simple but efficient method of controlling the flyback-generated oscillations in the 
line output stage is to connect across the line scan coil circuit a diode in series with a 
parallel RC circuit. This arrangement can also be made to play a useful additional 
role in helping to produce the line scan itself— as will be seen from the illustration 
below. 




Line Output 
Transformer 



When the beginning of the flyback causes the powerful back-e.m.f. across the line 
scan coils (it is, you will recall, positive-going), the sudden burst of energy which 
would otherwise have appeared as a ringing oscillation causes the diode to become 
forward-biased. Its anode conducts, and the energy of the oscillations is stored in the 
capacitor C in its cathode circuit. 

The diode continues to conduct for some time after flyback has been completed. 
While it does so, the anode current flowing in the scanning coils is used to produce the 
first half (or thereabouts) of the scan itself. It is only when C has given up its charge, 
and the diode has ceased to conduct, that the line output valve takes over and supplies 
the energy required for the second half of the scan. 

It is because of its performance in this double role (first, providing "emergency 
storage" for unwanted energy and then putting this energy to good use) that the 
diode, so connected, is often known as the efficiency diode. 



The Line Output Valve Operated as a Switch 

The presence in the circuit of an efficiency diode makes possible a different method 
of operating the PA valve which greatly reduces the amount of power dissipated within 
the valve when it is conducting, and so makes it a more efficient power amplifier with a 
longer useful life. 

It has been assumed hitherto that the line output valve is required to operate as a 
Class A amplifier (Basic Electronics, page 2.29) in which anode current flows during 
the whole cycle of the input signal. With the addition of the efficiency diode, this is 
no longer necessary. A waveform capable of giving a good linear scan can be pro- 
duced when the output valve itself is only conducting for the short period of some 25 jxs 
constituting the second half of each line scan. 

This means that the valve can effectively be operated as a mere electronic switch, 
with great resultant saving of power loss. You should now see how this is done. 



3.54 



[§20 



The Line Output Valve Operated as a Switch (continued) 

When a valve is operated as an electronic switch, it has two states only. It is 
either cut-off altogether when the switch is "open", or it is conducting heavily at 
saturation point when the switch is "closed". The sort of waveform which needs to 
be applied to the control grid of the valve to achieve this kind of operation is obviously 
rectangular in shape, and extending in amplitude from a value near zero volts when the 
valve is conducting, to a negative value beyond cut-off when it is not. 

The highly simplified illustration below may make it easier to understand how a 
linear scanning current can be produced by operating the line output valve, not as a 
power amplifier at all, but as a simple electronic switch. 



HT(+) 



Line Output 
Transformer 




Valve 



300 n. 



Line Scan 
Coils 



- 



Grid Voltage (V„) 



Cut-Off 
Level 

H - 



- Scan 
(Valve On) 



^ Flyback 
(Valve Off) 



mm£mWW& Operate*! as a 




When the waveform at the grid rises through cut-off to zero, anode current in the 
valve builds up and anode voltage quickly falls to saturation value, which is a few 
volts above zero. When this happens, the primary of the output transformer in the 
anode circuit of the valve is placed, through the valve, across the HT supply. The 
current which thus flows in the transformer, and therefore in the scanning coils, 
builds up at a constant rate and gives a linear scan of the picture tube. 

At the end of the scanning period, the grid waveform is again reduced to a value 
below cut-off and the current in the transformer is reduced to zero. 

When an efficiency diode is connected in a circuit in which the line output valve 
is operated as a switch, the two valves conduct in sequence and each makes a contri- 
bution to the scan. The valves are in effect operating in a manner analogous to two 
parallel-connected switches, one being closed while the other is open. 



§20] 3 - 55 

The Boost Diode 

There is another, and probably more widely used, way of connecting the efficiency 
diode so that, instead of acting in parallel with the line output valve, it acts in series 

with it. 

In this mode of connection, the energy received by the diode from the ringing 
oscillation during flyback is still used to charge up a capacitor (typically of some 
0-1 f*F in value); but now the voltage built up across the capacitor during flyback is 
connected in series with the HT supply to the line output valve, so boosting the anode 
voltage of the valve by as much as 250-650 V. Once again, the energy contained in the 
unwanted oscillations of the line output transformer is put to good use; and it is not 
surprising that an efficiency diode connected in this way is known as a boost diode. 

The basic arrangement of such a circuit is shown below, and its associated wave- 
forms in the illustration overleaf. Briefly, ,the circuit works as follows. 



HT(+) 



Line Output 
Transformer 




(-) 




7L/U" 



Line Scan 
Coils 



Input from 
Line Scan 
Generator 






^^^^^^^^tt^Hgliii^i^iiHi^ 



m 



At the moment when the latter part of the scan is being produced, the line output 
valve is conducting heavily and the diode is cut off. Anode current flowing in the 
pentode, and thus in the transformer primary, is rising linearly towards its maximum 
value reached at the end of the scan — the current being drawn as a discharge current 
from the boost capacitor connected in series with the HT supply and the transformer 
primary. 

At the end of the scan the pentode is cut off by its grid waveform, and its anode 
voltage is suddenly driven to a large value by the creation of the back-e.m.f. which 
results. After this rise has reached its maximum value, anode voltage swings nega- 
tively in the opposite direction and starts the first half-cycle of a ringing oscillation. 
This negative excursion makes the cathode of the diode negative with respect to its 
anode (the latter being held steady at HT value); and at the peak of the oscillation, the 
diode conducts. 

As it does so, diode current re-charges the boost capacitor. At the same time it also 
causes the scanning beam to start to move towards the right-hand side of the screen, 
because the diode current is flowing also through the secondary of the transformer, and 
so through the scanning coils connected to it. When the boost capacitor has become 
fully charged, current in the diode ceases to flow and its contribution to generating the 
line scan ceases simultaneously. 



[§20 



3.56 

The Boost Diode (continued) 

During the period when the diode is conducting, the pentode is held below cut-off by 
the negative portion of the waveform applied to its grid. But the trailing edge of this 
waveform is deliberately shaped so that, as it rises towards zero, it reaches the cut-on 
point of the pentode just before the diode ceases to conduct. In this way a smooth 
transition from diode to pentode conduction takes place, the linearity of the scan 
being unaffected by the change in the agency by which it is produced. The effect will 
be clearly seen by studying the I a curves of the pentode and of the diode (waveforms 
3 and 4 below). 

When the pentode begins to conduct, it once more draws its anode current from the 
boost capacitor, and the cycle of operation is repeated. 



PENTODE Vg 

Cut-off — 

(-) — 



PENTODE V a 



PENTODE I 



DIODE I a 




(+) 
Scanning 
Coil Current 



Ring which would occur if Boost Diode were absent. 

BOOST DIODE e/*ewT mm**** 

The point to grasp is that the first half of the scan is produced by the anode current 
of the diode and the second half by the anode current of the pentode. By reason of 
inevitable losses in the circuit as a whole, the proportions are modified in practice so 
that the pentode usually produces about 60% of the scan, with only the first 40% being 
produced by the diode. 

Nevertheless, with even 40% of the scan being derived with its aid from an unwanted 
and potentially tiresome ringing oscillation, it is easy to see how a boost diode adds 
to the overall efficiency of the line scan circuit. 



§20] 



3.57 



The Auto-transformer 

You may have noticed in the illustration two pages back that one end of the 
primary winding of the line output transformer is shown as being directly connected 
to one end of the secondary winding. This means, of course, that complete isolation 
between the two windings no longer exists, and that the transformer is operating as 
an auto-transformer {Basic Electricity, page 4.80). You will find this easier to 
appreciate if you redraw the circuit in another way. 

HT(+) 



HT<+) 



Boost 
C Diode 



Line Output 
Pentode 




Line Output 
Pentode 




Scan 
Coils 



EQUIVALENT 
CIRCUIT 



Transformer 



The auto-transformer is widely used in the line output stage of modern TV receivers. 
One advantage it gives is that the leakage flux between the two windings of the double- 
wound transformer — that part of the flux which fails to link the two windings — is much 
reduced (though not altogether eliminated). This is valuable, because this flux makes 
a significant contribution to the ringing oscillation which occurs during flyback. This 
oscillation is admittedly put to good use in generating the second half of the scan; but 
leakage flux nevertheless represents an energy loss in the circuits, so should be mini- 
mised where possible. 

With the scanning coils now directly connected to the anode circuit of the pentode, 
one could reasonably expect that a d.c. current would be flowing through them; and 
this would cause a permanent displacement of the scanning spot on the screen. But 
you have just seen that, if an efficient boost diode circuit is employed, the current 
flowing in the transformer reverses its polarity at a point in time midway through the 
scan. This effectively makes the scanning current a.c. in nature; and since the mean 
value of an a.c. current is always zero, no displacement of the spot will occur. 

Note that, in less efficient boost circuits, the point at which the scan ceases to be 
produced by the anode current of the diode and begins to be produced by the anode 
current of the pentode no longer occurs midway through the scan. This effectively 
means that the 7 a of the pentode is greater than the /„ of the diode, and the average 
value of the waveform of current delivered to the line scan coils will no longer be zero. 
For this reason, it is sometimes necessary to couple the output transformer to the 
scanning coils through a large-value d.c. blocking capacitor, in order to block the 
resultant d.c. element in the scanning waveform. 



3.58 

Width Control of the Picture 



[§20 



You saw in the last Section that adjustment of the vertical size (height) of the picture 
can be simply made by adjusting the amplitude of the scanning waveform applied to 
the control grid of the field output stage. Unfortunately, control of the horizontal 
size (width) of the picture is not so simply achieved. The reason is that any variation 
in the amplitude of the waveform applied to the control grid of the line output valve 
would be likely to disturb both the boost and the EHT voltages (of the latter, more 
anon), and so to impair the linearity of the scan. Other methods of width control 
must therefore be found. 

One such method is based on a switching, plug-and-socket, arrangement which 
enables the turns ratio of the line output transformer to be varied. 

HTW 



Ganged 
Switches 



Line Scanning 
Coils 



Line Output 
Valve 





f±\ 



<y 



■& 



WIDTH CONTROL: 

A Typical Arrangement 



It will be seen that the scanning coils can be connected at will to a number of different 
taps on the transformer winding, and the viewer can switch from one to another of 
them until the correct width of picture is obtained. A ganging arrangement enables 
shunt capacitors of appropriate value to be automatically connected across the coils 
every time a different transformer tap is selected. This ensures that a constant total 
current is supplied to the selected shunt-capacitor-scanning-coil combination, whatever 
the width of the picture. In this way, a constant voltage is maintained across the 
transformer, and steady EHT and boost voltages are achieved. 

(Ignore D 2 for the time being. You will see what it does on the next page.) 

Transformer Whistling 

The cause of the characteristic whistle which may be heard coming from most TV 
receivers is mechanical vibration of the line output transformer core at the line fre- 
quency. It is brought about by what is called the magnetostrictive effect. The 
vibration occurs at about 10 kHz when the receiver is operating on the 405-line 
standard — a frequency well within the audio-frequency range, though few older 
people will agree! On the 625-line standard, the vibration occurs at the higher fre- 
quency of about 16 kHz, and is therefore inaudible except to those with exceptional 
hearing. 

Vibration of the transformer core can be reduced by mounting the transformer on 
sound-absorbing material and enclosing it in a metal container lined with foam 
rubber. This container also serves as a screen against r.f. radiation from the trans- 
former which could cause interference at harmonics of the line frequency, and so in the 
lower operating bands of some radio receivers. Particularly vulnerable in this respect 
are small receivers tuned to the medium waveband and having a built-in aerial. 



§20] 



3.59 



EHT Voltage Generation 

The very large back-e.m.f.- (typically of more than 2 kV) which is created at the 
anode of the line output valve every time it is cut off during the flyback period can be 
exploited by adding some extra turns to the secondary of the output transformer. The 
magnitude of the back-e.m.f. is in this way deliberately stepped up even higher, to a 
value which, after rectification, can be used to supply the picture tube with its required 
EHT of 15 to 20 kV. (You will see in the next Section why so large a voltage is 
required to operate the tube.) 



HT(+) 



Autotransformer 



Line Output 
Valve 




^- EHT to Picture Tube 



lOOOp 

~i — Smoothing 
_J Capacitor 



Heater Winding 



tow EHT * Generated from the Line Output Trar>sform«r 

■%?\ X--9:k .'•' '••" •■' ■ "■'•■*' ''•.'.''.'■ ; ; '•' - ■ X'.' '.-r'-f !■'■'■ ■"'/}V^-"'-'--'^ v lHk-^ 

In the illustration above, the extra turns (they are usually called the EHT overwind) 
are shown as an additional secondary winding on the auto-transformer. When the 
line output valve is cut off, the back-e.m.f. pulse which appears at the anode of the 
valve is amplified 8-10 times by reason of the presence in the auto-transformer of these 
extra turns on the secondary winding. 

Thus amplified, the pulse appears at the anode of the EHT rectifier (D 2 in the illus- 
tration). This is a special thermionic diode capable of handling very high voltages. 
The pulse is rectified in the normal way (Basic Electricity, page 3.19) and appears at 
the cathode of the diode as a positive d.c. potential of 15-20 kV. 

Remember that the rate at which the back-e.m.f. pulses recur is very high indeed — 
10-125 times a second in the 405-line system and 15-625 times a second on 625 lines. 
The ripple frequency of the rectified d.c. voltage coming from the rectifier diode is 
correspondingly high, and the voltage therefore requires little smoothing to make it 
usable by the picture tube. The type of smoothing capacitor commonly used (it 
would have a value of about 1000 pF) actually forms part of the physical construction 
of the picture tube itself, and you will see how it works in the next Section. 

A valve such as the EHT rectifier diode requires a heater supply of its own. In the 
diagram, this supply is shown to be derived from a very small extra winding, of between 
2 and 6 turns only, added to the auto-transformer. It develops some 6-3 V. 



3.60 [§20 

EHT Voltage Generation (continued) 

An EHT supply developed in the manner described is said to be of the "flyback- 
derived" type. Such supplies possess very high internal impedances — which means 
that they are capable of delivering only small load currents. In a TV receiver, this is 
actually an advantage ; for the picture tube itself needs only a few hundred uA of 
current, and the fact that the supply cannot deliver a large current means that it is 
unlikely to kill you if you accidentally touch it. 

Never forget, however, that the EHT supply in your TV receiver is carrying so high a 
voltage that it can still give you a very nasty shock indeed. It must never, in any cir- 
cumstances, be touched while the set is switched on. Indeed, so dangerous is its potential 
that you should make it a rule never to poke about in the inside of your set — even with a 
well-insulated screwdriver — without pulling out the wall-plug which feeds your set with 
HT from the mains. 

One major drawback of the large output impedance associated with an EHT circuit 
of this type is that the supply will have very poor regulation. This means that the 
EHT voltage will vary in magnitude every time the load current (which is the beam 
current drawn by the picture tube) changes. This happens frequently — as when a dark 
scene is replaced by a bright one, or when the mains supply to the receiver varies. 

Variations in EHT voltage cause the picture size to change (picture shrinks when 
EHT is increased, and vice versa) ; and the extent of the changes may be sufficient to 
become objectionable to the viewer. The effect is less pronounced, however, in circuits 
in which the line output valve is used as an electronic switch, because the EHT is then 
dependent on the absolute magnitude of the HT voltage derived from the mains. 

Third Harmonic Tuning 

The magnetic coupling between the EHT overwind and the remaining sections of the 
line output transformer cannot be made as tight as that between the other windings 
themselves because of the very high voltage involved and the consequent need for 
extra-good insulation. In other words, the overwind cannot be wound too close to 
the primary for fear of a voltage breakdown between them. As a result, there is a 
considerable leakage inductance associated with the EHT overwind; and this, in 
conjunction with its stray capacitance, can give rise to severe ringing. 

The most usual modern way to prevent ringing from this source is to design the 
output transformer in such a way that the ringing from the overwind occurs at a 
frequency equal to the third harmonic (actually, it is the 2-8th harmonic, but the dif- 
ference is not important) of the line frequency. The phase relationship between the 
anode voltage of the line output valve and the ringing oscillation across the overwind 
then becomes such that the ringing is passing through a minimum at the time the 
anode voltage is reaching its maximum. The ringing is thus caused to be of insignifi- 
cant amplitude at the critical moment of the scan. 

Line Output 
Transformer 



Line Output 




Standard Selection 
Switch 

^7 



Valve ™ g Extra Winding flf 

E H 0v e;nd ^- LIKE OBTWfT TOUTS* ORHER 



§20] 



3.61 



Third Harmonic Tuning (continued) 

Because of the difference in line scan frequency between the 405- and 625-line 
systems, it is necessary to re-tune the transformer every time you switch standards so 
as to ensure that the overwind is always tuned to the third harmonic of the particular 
line scan frequency being employed. This can be achieved in several ways, one of 
which makes use of another extra winding of about ten turns on the autotransformer 
which is physically situated underneath the EHT overwind. 

When the receiver is set for 405-line operation, this extra winding (L 3 in the illus- 
tration overleaf) is open-circuited by switch S ls and the transformer is tuned to the 
third harmonic of the 405-line operation (i.e., to 3 x 10-125 = 30-375 kHz). When the 
receiver is set for 625-line operation, the extra winding causes an increase in the 
coupling to the overwind, which effectively changes the magnitude of its leakage 
inductance. This change is arranged to be of such a value that the overwind is retuned 
to the third harmonic of the 625-line scan frequency (i.e., to 3 x 15-625=46-875 kHz). 



S-Correction in the Line Scan Circuit 

S-correction in the line scan circuit is often achieved by connecting in series with the 
line scan coils a capacitor whose value has been carefully selected so that it distorts the 
shape of the scanning current into the elongated and flattened "S" shape required. 
The effect of introducing into the line scan circuit of a modern flat-faced 110° picture 
tube a 0-1 jxF capacitor is to increase the peak value of the scanning current by about 
7%. This cancels out part of the inductance of the line scan coils and so introduces 
the desired degree of S-correction; but an explanation of exactly what happens would 
take you some way beyond the scope of this Basic TV series. 



Line Output 
Valve 



£ 



<y 




-t- HT (♦) 



o 

5 



S 



-© 



Line Scan 
Coils 



0-35)1 

dj-ll-o* 25 



SB 



"Hi-oior 



OlSfi 



lS-C0fTO^ 
iaH* 



In dual-standard receivers, a different S-correction capacitor (Q-Ca in the illus- 
tration above) is brought into circuit every time the Standard Selection switch is 
operated, because of the different scan periods of the two systems. 



3.62 

The Line Output Stage— Full Circuit 



[§20 



HT(+) 
200V 



>- = 600V for First Anode of 
Picture Tube 




■01 

Input — | 
from 

Line Scan 
Generator 



Line Scan 
Coils 



V3 

DY86 



- 17kV EHT for Final Anode 
of Picture Tube 
[1000p 

~~J" Picture Tube Capacitance 



ii 



Mte 



v.ri;i-;i^;ai:i^va'-c-: 



LINE OUTPUT STAGE 






■'•'i$:Wi ...... 



§20] 



3.63 



The Line Output Stage— Full Circuit 

The illustration opposite shows the full circuit diagram of a line output stage rep- 
resentative of modern design techniques. 

The main winding (section A-B) of the output auto-transformer is labelled L l5 the 
EHT overwind (section B-C) L 2 . L 3 is the third-harmonic tuning inductance, switched 
into circuit only when the receiver is set for 625-line operation; and L 4 is the heater 
winding for the EHT rectifier diode V 3 . The 1-ohm resistor in series with this winding 
is used to trim the heater voltage to the correct value. To get this voltage directly 
would require a fractional number of turns on L 4 . It is much easier to use a whole 
number of turns and then trim the voltage produced by adding a small resistor. 

The line output valve Vj. is supplied at its control grid with a rectangular waveform 
having an exponential trailing edge, coming from the line scan generator. Its screen 
grid is connected to HT, which is of the order of +200 V. 

The boost diode V 2 appears upside-down compared with the circuit you studied a 
few pages ago, but it works in exactly the same way and supplies charge to the 0-l-(xF 
boost capacitor C 3 connected in series between the HT line and point A on the output 
transformer. The boost voltage (typically, 600 V) appearing at point A is also used to 
provide the potential required by the first anode of the picture tube, via the smoothing 
circuit formed by the 1 M resistor and 0-1-fi.F capacitor shown. Smoothing is 
necessary in this case because the boost voltage is fluctuating at the line frequency. 

Width control is provided by the transformer tap selection method you already 
know about; and S-correction of the line scan current waveform by the alternative- 
capacitor arrangement labelled Q-CV The switches S x and S 2 are, as usual, sections 
of the Standard Selection switch. 

The Line Output Transformer— Physical Appearance 

The typical line output transformer is of unusual shape. The EHT overwind, 
which is rather like a catherine-wheel in shape, is wound over the main anode winding. 
It is kept narrow so as to maintain good insulation, and is usually impregnated with 
wax to stop moisture getting into it. The heater winding for the EHT rectifier diode 
is wound on the opposite leg of the transformer core, and consists of wire having 
thick polythene insulation. 

The transformer core itself is made of a ferrite material (minute particles of iron 
oxide fused together to form a ceramic) and consists of two U-shaped pieces clamped 
tightly together by securing bolts and a clamping frame. The mating faces of each 
U-piece are insulated from one another by thin pieces of paper, to prevent core 
saturation. 

The EHT rectifier diode is mounted either on a tag strip on top of the transformer 
or.in a valve holder alongside; and the solder joints associated with it (heater and anode 
connections) are carefully rounded off so as to make what is known as corona discharge 
from these points less likely. All sharp-pointed projections tend to concentrate the 
electrostatic field from the EHT voltage and so to encourage corona discharge. 

If you want to see what corona discharge is like, look into the back of any well-worn, 
dust-encrusted TV receiver when the room is in total darkness (and preferably when 
humidity is high). TOUCH NOTHING— but watch out for a faint crackling noise 
and for small areas of bluish-white light coming from regions associated with the EHT 
voltage. You may even smell the ozone which is created when air is ionised by the 
corona discharge. 

Discharges of this kind result in less EHT being available for the picture tube, and 
therefore in reduced picture brightness. They can also give rise to an r.f. radiation 
which may interfere with the receiver itself, or with others situated nearby. 



3.64 

REVIEW of Line Scan Development 



[§20 




The line scan output stage accepts the 
waveform generated by the line scan genera- 
tor and uses it to produce a powerful 
scanning current which it supplies to the line 
scan coils. These coils are predominantly 
inductive, and have comparatively low 
impedance. They are coupled to the line 
output valve by an impedance-matching 
transformer. 



Very large back-e.m.f.'s are produced by 
the line scan coils during the short-duration 
flyback periods of the line scan. These 
e.m.f.'s, passing back through the impedance 
matching transformer, would (if allowed to 
do so) appear at the anode of the line output 
valve in the form of a ringing oscillation 
having a peak amplitude of more than 
2000 V. 




Diode ^ 




f) a"( 


A 


\^ jof 


to 




to 




N" 


: 1 ȣ cf! 


n 


T f 


Line 
Co 



They are prevented from reaching the anode by means of a diode connected in series 
with a capacitor which not only stores much of the energy contained in these dangerously 
high voltages, but puts it to good use in providing the first 40% or so of the following 
line scan. 

In its simplest form, this diode is known as an efficiency diode. 






Grid Voltage (V B ) 


Cut-Off 
Level 


"" (Valve On) " 




""*~ Flyback 
(Valve OH) 



vtttm 



mm Spar** as a 5WITQS 



When an efficiency diode is in use, the line 
output valve is only required to produce the 
last 60% of the complete line scan. With 
the requirement it must fulfil reduced in this 
way, the valve can be operated as an 
electronic switch instead of as a "Class A" 
linear amplifier. 

Its input waveform has a sharply rec- 
tangular leading edge, but a trailing edge 
with a pronounced exponential curve. 



§20] 

REVIEW of Line Scan Development (continued) 



3.65 



Efficiency diodes of improved design use 
the energy of the back-e.m.f.'s generated 
during line flyback to produce a positive 
voltage which can be applied so as to add to 
the value of the HT voltage delivered to the 
line output valve, so further improving the 
efficiency of the line output stage. 

When the efficiency diode is operated in 
this way, it is known as a boost diode. 




7L/^i 



Autotransformer 




The large back-e.m.f.'s created by the 
line output stage are put to yet another use 
by means of an additional winding on the line 
output transformer. This overwind pro- 
duces a very high voltage which, after 
rectification, is used to supply the picture 
tube with the EHT voltage of 15 to 20 kV 
which it needs for efficient operation. 

The rectification is done by a small diode 
capable of handling very high voltages, 
which in turn derives its heater supply from 
yet another winding (of a few turns only) on 
the line output transformer. 



Because of the high voltages involved, the 
magnetic coupling between the EHT over- 
wind and the other windings of the line output 
transformer cannot be made tight. In con- 
sequence, there is a considerable leakage 
inductance associated with the overwind. 




EHT OverwirxJ 



This inductance is caused to resonate with the self-capacitance of the circuit at a 
frequency equal to about three times that of the line scanning frequency. This ensures 
that the ringing produced by the inductance occurs at moments when minimum voltage is 
present at the anode of the line output valve. 

This further way of reducing the damage which could be done by over-large back- 
e.m.f.'s is known as third harmonic tuning. 



3.66 



§2 1 : THE PICTURE TUBE 



You have now followed out the different routes by which the vision signal and the 
synchronised field and line scans are brought to the picture tube. The time has at 
last come to take a closer look at the component which provides the picture you see on 
the screen of your TV receiver, and to learn how it is activated and modulated by the 
different signals applied to it. 

In all essentials, the picture tube in a TV receiver is little different from the types of 
tube used in a radar set or in an oscilloscope. All are cathode-ray tubes operating in 
the same basic way, and containing a similar internal-electrode structure. The 
essential differences between them are merely matters of size, shape and external 
appearance. 

You know from page 1.29 that the operating principle of the TV picture tube is for 
the received signals to be caused to modulate the intensity of an electron beam, which 
has been accurately synchronised with the scanning beam in the camera tube in the 
studio, as it scans the fluorescent face of the picture tube. Before seeing what is 
involved in putting this principle into practice, you would do well to re-read the 
Section on the cathode ray tube in Basic Electronics, pages 5.100 to 5.110, for most of 
it applies to the TV picture tube as well. 

The modern TV picture tube has a nearly rectangular-shaped screen whose dimen- 
sions approximately match the 4 : 3 aspect ratio of the televised picture. It is customary 
to denote the size of a particular receiver by quoting the corner-to-corner diagonal 
measurement of its screen face— 17", 19", 23" and so on at the time of writing, though 
the metric equivalents will doubtless soon be quoted instead. 



Screen 




Cone 



Neck 



Deflection 
Angle 




4* If-Ml PWTURE T&K 



Note that the depth of the tube from the face of its screen to its plug-in base is quite 
short compared to the dimensions of the screen itself. The advantage of this is, of 
course, that the overall depth of the receiver cabinet can be kept reasonably shallow. 
This has not always been possible. Tubes in earlier receivers had circular, convex 
screens which were then covered by a rectangular plate-glass mask in the front of the 
cabinet. They also had long necks, which meant either that the cabinet had to be of 
great depth or that part of the tube neck stuck out of the back of the cabinet covered 
by a "top-hat" kind of extension to protect it. 



§21] 3.67 

The Picture Tube (continued) 

With continuing improvements in tube manufacturing technology over the years, 
picture tube screens eventually became almost perfectly rectangular in shape; and the 
length of the neck was progressively shortened by increasing what is called the de- 
flection angle. This is the angle through which the scanning beam must be deflected 
for it to reach the outermost (i.e„ extreme left-hand to extreme right-hand, and 
extreme topmost to extreme bottom-most) edges of the screen. The angle is measured 
from a point known as the deflection centre, which is close to where the flared-out cone 
of the tube joins the neck. 

As a matter of historical interest, picture tubes of 1946 vintage had a deflection angle 
of 52° and an overall depth (from front to back) of about 18" for a 10" screen. As 
deflection angles were progressively increased through 65°, 85° and 90° (in 1953) to 
the present-day 110° and 114° tubes, so tube depths decreased in step until a 23" 
screen in 1971 is little more than 14" deep. 

Further significant increases in scanning angle seem unlikely, for tube depth is no 
longer the most important limiting factor in cabinet design. The size and shape of 
other receiver components, and of the chassis itself, are nowadays the predominant 
considerations. 




PICTURE TUBE 



The internal electrode structure of a picture tube consists of two basic parts. The 
first, situated in the neck of the tube, is an electron gun assembly. The second part, 
situated on the flared edges of the cone itself, is the final anode. You will now see 
how they both work. 



3.68 [§2 , 

How the Picture Tube Works 

The gun assembly used in a modern picture tube is similar to those described in 
Basic Electronics, pages 5.100 to 5.110. The diagram below illustrates the electro- 
static method of beam focusing, though the electromagnetic method is also used. 

When the cathode is heated, electrons are liberated from it and forced to pass 
through the hole in the end of the cylindrical grid surrounding the heater-cathode 
assembly by high positive potentials (typically, 200 to 800 V) on the first and second 
anodes. A negative charge on the grid repels the electrons as they pass through the 
hole in its end and concentrates them into a narrow beam. The video signal (as you 
learnt in Section 17) is applied to the cathode of the gun and modulates the number of 
electrons leaving it at any one instant of time, and so the intensity of the beam for 
that instant. You will recall that the polarity of the signal is such that its most 
negative excursions represent the highlights of the scene by supplying more electrons 
to the stream emitted from the cathode. 



Base 



Grid 



1st Anode 




Final Anode 
Connector 



Heater Cathode 2nd Anode 



The £UCT*0H CUff Assembly 



Final Anode 
Aquadag 




Fluorescent 
Screen 



The HNAl AH0M 



In addition to accelerating the electron stream away from the grid by the positive 
voltages placed on them, the first and second anodes (as you will shortly see) prevent 
the beam from spreading and focus it into a tiny spot on the inside face of the screen. 

After leaving the electron gun assembly, the beam comes under the influence of the 
line and field scan coils which deflect it across and up-and-down the face of the screen 
respectively; and it then receives its final acceleration from the final anode. This 
electrode is formed from a very thin coating of a graphite composition known as 
Aquadag applied to the inside surface of the glass cone. It carries a very high positive 
voltage which, for a 23" tube, would be in the region of 18 kV. External connection 
to it is made by a small metal plug mounted on the outside of the tube, passing through 
the glass and bonded to it to maintain a hermetic seal. 

External connections to the heater, cathode, grid and first and second anodes are 
made through a 12-pin (duodecal) valve-type base bonded to the far end of the tube 
neck. Connection to the final anode cannot be made through this base for fear of 
flash-over between the closely-spaced pins, so the metal plug is used instead. 

The screen of the picture tube consists of a thin layer of fluorescent material de- 
posited on the inside surface of the glass face, the material used (generally a mixture of 
zinc sulphide and zinc-beryllium silicate) having a high conversion efficiency in that it 
produces a high light output when bombarded by electrons, with the right fluorescent 
colour for displaying a black-and-white picture. It also has an after-glow short 
enough to prevent smearing when fast-moving objects appear in a scene. 



§21] 



3.69 



Focusing the Electron Beam 

The stream of electrons leaving the electron gun is accelerated towards the screen 
by the highly attractive force of the large positive potential on the final anode. But 
before it can be used to trace out a raster consisting of hundreds of very fine lines on the 
screen, it must first be shaped into a narrow beam having a very small cross-sectional 
area. 

This can be done by using either an electrostatic or an electromagnetic field to exert 
a force on the electrons as they travel towards the screen. The effect of this force 
is to deflect every electron in the beam slightly inwards towards the axis of the tube 
so that they all eventually cross the axis at a single point. By careful adjustment of 
the strength of the field, the cross-over point where all the electrons converge can be 
made to occur exactly at the surface of the screen. The result is the illumination of a 
tiny area of the screen (often no more than a few tenths of a millimetre in diameter) 
by a spot which, when the beam is made to scan, is able to trace out thin crisp lines and 
a clear picture. If beam focusing is poor, it will not be able to resolve fine detail, 
and the result will be a picture with fuzzy outlines throughout. 



Scanning 
Beam 



Focusing 
Field 




Electron 
Gun 



Picture Tube 
Screen 



FIELD TOO WEAK 



CORRECT FIELD FIELD TOO STRONG 



FOCUSING the Scanning Beam 



The focusing field is applied along the neck region of the tube, and is created either 
by an external magnet surrounding the neck of the tube (electromagnetic focusing), 
or by electrodes placed within the tube itself (electrostatic focusing). Both methods 
are used in modern TV picture tubes, and must therefore be examined. Part 5 of 
Basic Electronics provides a good introduction to them both. 



Electrostatic focusing 

In this method of focusing, internal electrodes are used to produce between them an 
electrostatic field such that an electrostatic lens is formed. By varying the strength of 
the field, the focal length of the lens can be adjusted, and with it the point at which the 
electrons in the stream passing through it converge. The electrodes used are the 
first and second anodes in the electron gun assembly — the same electrodes as are used 
to accelerate the electrons towards the screen. 



3.70 [§2 , 

Electrostatic focusing (continued) 

The illustration shows a simplified form of one electrode arrangement for electro- 
static focusing, but there are a good many other arrangements in common use. 

2nd Anode 

1st Anode 

L Scanning \ 

^— ^^ m ^T"~ mm / Beam \ 

tla- ■ \s~*i —j?Ljr jS / Screen 

Tube Axis 




Control " *\ U \ rau ffo BAStC 



(+)100V 



ELECTROSTATIC LENS 



Remember that you are looking at a two-dimensional view only. You know that 
the first and second anodes are both cylindrical in shape, so that the electron beam 
has depth "through the paper" as well as the length and breadth shown. 

The potential on the first anode is set by the focus control, and is maintained at a 
value permanently lower than that on the second anode. The difference in voltage 
between the two electrodes causes an electrostatic field to exist between them, with the 
lines of force running in the direction shown. When electrons in the scanning beam 
encounter the fields created by the two electrodes, those which are divergent from the 
axis of the tube (and therefore trying to cut the lines of force) are deflected by them 
back towards the axis. Those already moving along the direction of the axis run 
parallel with the field and are therefore merely accelerated, rather than deflected, by it. 

By varying the potential on the first anode, it is possible to make the point of 
convergence of the electron beam occur at the surface of the screen. The setting of the 
focus control is done by the manufacturer; it should rarely require attention during the 
life of the receiver. 

It may have occurred to you to wonder how focusing is affected by the differing 
distances which the beam has to travel when scanning the different areas of a modern 
flat-faced screen having a wide deflection angle. When you were studying S-correction 
in the two preceding Sections, you saw that this distance was greater when the beam 
was on the periphery of the screen than it was when it was scanning its central areas. 
For this reason, an electrostatic field of such strength that the electron beam it 
controlled focused at a point when the beam was in the centre of the screen would be 
too strong when the beam was scanning an outside edge, and the picture at that point 
would tend to become fuzzy. 

The solution adopted is, as usual, a compromise. Most manufacturers set the 
focus control on the first anode to a value which will cause the electrons on the beam 
to converge to a single point when the beam is approximately one-quarter of the way 
across a line scan one-quarter of the way down a field. In this way, the degree of 
fuzziness caused by the focal length of the beam failing to coincide exactly with the face 
of the screen at every point on its surface is reduced to an overall minimum. It would, 
in fact, not be great enough to cause much impairment of the picture even if the focal 
length of the beam was set at dead centre of the screen. 



§21] 



3.71 



Electromagnetic Focusing 

In this form of focusing, the electron lens is formed by an electromagnetic field 
rather than by an electrostatic one; but its effect on the electron scanning beam is 
precisely the same. 

In the earlier TV receivers, the magnetic field was created by a ring-type coil 
assembly placed round the neck of the tube. The strength of the field, and hence the 
control of focus, was adjusted by varying the amount of current flowing through the 
coil. This method was simple to adjust; but the coil assembly tended to be bulky, 
and a degree of stability higher than could be maintained irt practice was demanded of 
the voltage supplying current to the coil. When this voltage "wandered" from its 
intended value, focusing efficiency suffered. 

In later receivers, the focusing coils were replaced by a pair of permanent magnets, 
also ring-shaped, slipped over the neck of the tube. Focusing adjustment was less 
simple than it had been before, having to be effected by mechanical movement of the 
whole magnet assembly; but there were compensating advantages. The essential 
parts of such an assembly are shown in the illustration below. 



Ring-type 
Magnets 



Adjustabfe 
Disk 



Screen 





Ring-type Magnets 



Layout of a typical 

ELECTROMAGNETIC FOCUSING AssemW, 



The two ring magnets are made of a Ferrox-type amalgam of iron and barium oxide, 
and one of them is adjustable lengthwise along the tube on which they are both moun- 
ted. The magnetic fields on the two rings are caused to be always in mutual opposition ; 
and focusing adjustment is made by varying the position of one magnet with respect to 
its neighbour, usually with the aid of a small lever projecting from the magnet housing. 

When the magnets are moved farther apart, the strength of the focusing "lens" is 
reduced, and the point at which the electrons in the scanning beam converge is moved 
forward towards the screen. When the magnets are moved closer together, the 
increased magnetic forces brought to bear on the lens cause it to focus the electrons 
to a spot at a point nearer the magnets, and so further back from the face of the screen. 

Electromagnetic focusing has been widely used in TV picture tubes for some 
considerable time, but the space which the magnet assembly takes up along the neck 
of the tube, and its bulk, has led to increasing use of the electrostatic method of 
focusing in recent years. 



[§21 



3.72 

Deflecting the Scanning Beam 

Now that you have seen how the scanning beam can be focused to a tiny spot at the 
exact point where it strikes the screen, the next thing is to see how it is deflected many 
times (and very rapidly indeed) across the face of the screen and at the same time 
(though much more slowly) down it from top to bottom. 

Of the two means of deflecting the scanning beam across the face of a cathode ray 
tube mentioned on pages 5.104/105 of Basic Electronics, the electrostatic method is 
used extensively in CRO's and precision radar displays, but very seldom in TV 
receivers. The reason is that it is much easier to generate from reasonably low voltages 
the high scanning currents required in electromagnetic deflection than it is to generate 
the high voltages at low current load which are needed in .electrostatic deflection. 
Almost all TV receivers therefore achieve deflection of the scanning beam by electro- 
magnetic means. 

Each of the two scanning coil assemblies required in the TV receiver— one for the 
horizontal (line) scan and the other for the vertical (field) scan — consists of a pair of 
coils situated 180° opposite one another around the neck of the picture tube, close to 
where it joins the flared section of the cone. Viewed from the front of the screen, 
the line coil pair for the horizontal scan (call them YL X and H 2 ) is mounted above and 
below the neck of the tube, whereas the field coil pair for the vertical scan CV\ and V 2 ) 
is mounted on either side of it. Remember that the horizontally-positioned coils 
deflect the beam vertically, and the vertically-positioned coils deflect the beam 
horizontally. 

All the coils (H and V alike) are wound from many hundreds of turns of wire, and 
in their simplest form are shaped to resemble a hollowed-out saddle. This shape 
ensures that the coils fit snugly round the neck of the tube, and so helps to maintain a 
uniform magnetic field within it when the scanning currents are applied. 



Field 
Coils 




M 



how the SCANNING COILS 



§21] 3.73 

Deflecting the Scanning Beam (continued) 

Any pencil-like beam of electrons behaves just as does a thin conductor carrying a 
direct current. As you learnt on page 1 .45 of Basic Electricity, it will therefore have 
a magnetic field surrounding it; and the stronger the current the more intense will be 
the field. 

Now imagine that you are able to look right through the screen of the picture tube 
as you sit in front of it, and that you can see the scanning beam coming towards you 
from the electron gun. Represent the cross-sectional area of this beam by a circle, 
in the centre of which is a spot which might be the tip of an arrow coming very fast 
indeed straight for your eye. Cast your mind back to the "Left-Hand Rule" for 
determining the direction of the lines of forces which are set up around a conductor 
when an electric current is caused to flow through it, and you will realise that the lines 
of force in the field surrounding the beam are as they are shown at A in the illustration 
below. In other words, they are running clockwise round the tip of the arrow. 



Neck of 
Picture Tube 




Scanning Beam 




Neck of 
Picture Tube 



Imagine, next, that the poles of two magnets are placed one above and one below 
the "arrow", the top magnet being the N pole and the bottom one the S pole. The 
lines of force existing between these two poles will travel from N to S (i.e., from top to 
bottom). In the absence of the "arrow", the field from the magnets would exist as a 
series of parallel lines of force, as shown at B in the illustration; and the stronger the 
magnets, the greater would be the strength (flux density) of the field. 

Now consider what happens when the "arrow" of the scanning beam comes slicing 
through the beautifully straight field produced by the magnets. 

Because of the relative directions of the two sets of flux lines, those on the right-hand 
side of the scanning beam (as you look at it) will tend to reinforce one another, since 
they are moving in the same direction; whereas those on the left-hand side will tend 
to cancel one another out, since they are moving in opposite directions. The effect 
will be that a resultant magnetic force will be applied to the beam in such a direction 
that the beam will be forced towards the 
left-hand side of the picture tube. 

Remember, though, that all the time the 
distorted field of the two magnets will be 
straining to return to its normally undis- 
torted condition — just as the string of a 
bow would seek violently to return to 
normal once the constraint of the archer's 
fingers was removed from it. 



Direction of 
Movement -■ 
of Beam 




Neck of 
Picture Tube 



3.74 



[§21 



Deflecting the Scanning Beam (continued) 

If the field produced by the two magnets is weak, there will be few lines of force to 
react with the scanning beam, and it will be only slightly deflected. But if the 
strength of the deflecting field is gradually increased from zero to a high value, the 
scanning beam will be completely deflected to one side, its rate of movement being 
governed by the rate of increase of the deflecting field. 

If the poles of the two magnets were to be reversed, of course, the scanning beam 
would be deflected in the opposite direction. So if it is desired to deflect the beam 
completely from one side of the screen to the other, all that is needed is to start with 
the deflection field at maximum strength with the magnet poles in one position (when 
the beam will be completely to one side), and then gradually increase/reduce the field 
through zero (when the beam will be central) to maximum strength once more, but this 
time with the polarities of the magnetic poles reversed. 

Take the line scan as an example, for it is rather the easier to understand. With 
zero current flowing through the scanning coils, the beam will be stationary in the 
centre of the screen. When a large current is passed through the line scan coils in 
such a direction that the upper magnet in the illustration becomes a North pole, the 
beam is forced over to the left-hand side of the tube— the current at this moment being 
said to have "maximum value in the negative direction". 

Now suppose that current flow through the coils is made gradually less and less 
strong until it becomes zero, and is then, without any pause, made to flow more and 
more strongly in the opposite direction. (In electrical terms, it is said to increase 
linearly from maximum value in the negative direction, through zero, to maximum 
value in the positive direction.) The polarities of the magnet poles alter, with North 
becoming South and vice versa; and the electron beam will be forced over to the right- 
hand side of the picture tube. 

These two states are pictured in the illustration below, the left-hand diagram showing 
the beam at the start of a line scan and the left-hand one showing it at the end of the 
scan, with current flow now maximum in the positive direction and the polarities of 
both magnets reversed. 





Current Maximum in 
Negative Direction 



Current Maximum in 
Positive Direction 



How Altering the Direction of Current flow through the 

Scanning Coils causes a Scan to be Traced Out across the Picture Tube 



§21] 



3.75 



Deflecting the Scanning Beam (continued) 

Now, with the beam hard over to the right of the tube, the direction of current 
flow is suddenly made to collapse from maximum in the positive direction, through 
zero, and (very quickly indeed) right over to maximum in the negative direction once 
again. This time the change in the direction of flow is not so much linear as practically 
instantaneous, and the beam zips back to the left-hand edge of the tube ready to begin 
another line scan. 

Reduced to its essentials, this is what is done to produce both the line and the field 
scans of a TV picture tube. The two magnets are replaced by electromagnets, con- 
nected to one another either in series or (less frequently) in parallel, and currents are 
made to flow through them in such a way that they increase smoothly (though at very 
different rates) from a maximum value in the negative direction to a maximum value 
in the positive direction, and then collapse very rapidly indeed to their original states. 

The Line and Field Scan Coils 

The four electromagnets used — two of them (Hj and H 2 ) for the horizontal line scan 
and two (V! and V 2 ) for the vertical field scan — are shown diagrammatically in the 
illustration below. Note that those controlling horizontal movement are positioned 
above and below the beam, and those controlling vertical movement are placed on 
either side of it. By increasing or decreasing the current in the line coils at a very high 
rate while at the same time much more slowly increasing the current in the field coils, 
a series of horizontal lines can be traced out one below the other on the screen — and 
a scanning raster is produced. 




Line Scan Current 



\-^\^^\^ 



V-T^ 



Hi 






^^t&fifEid"^WL^mi 



mm 




3.76 [§2 , 

Corner Cutting 

The coil assemblies used in present-day TV receivers differ greatly from the simple 
arrangement shown on the last page, principally because of the large screens and wide 
scanning angles of the modern picture tube. 

You will realise, in the first place, that the larger scanning distances demanded of 
the line scan (remember the 4:3 width-to-depth ratio of the screen) mean that a 
considerably greater power output is required from the line output stage than from the 
field output stage. For this reason alone it is important to maintain the efficiency of 
the line scan at as high a level as possible. 

The effect of a wide scanning angle has further importance as regards the line scan, 
because if the scanning coil assembly is positioned too far back along the neck of the 
tube, the deflected beam will be prevented from reaching the far corners of the screen 
by striking the neck of the tube and being absorbed by the highly positive potential on 
the final anode. The result would be a loss of picture at the edges of the screen. 

The efficiency of the line scan circuit is improved, and corner cutting (as it is called) 
is simultaneously prevented, by so shaping the line deflection coils that for part of their 
length they follow the curvature of the picture tube cone. When the ends of the coils 
are flared in this way, they can be moved closer to the screen, and the deflection centre 
is moved forward with them. The problem of corner cutting no longer arises; and 
another advantage is that the magnetic field produced by the flared section now extends 
well into the cone area of the tube, so increasing its controlling influence on the 
scanning beam and improving the efficiency of the scan. , 



Corner 






Cutting 


^ Scanning 




Horizontal 1 / /*\ 


Beam 




Deflection 1 / \ 




Flared 


Coils \ 1/ 1 




Coils 



Deflection 
Centre 




CORNER CUTTING and its Prevention 



The problem of corner cutting is not nearly so serious with the vertical scan because 
of the smaller distance through which the vertical scanning beam needs to be deflected. 



3 77 

§21] 

A Practical Scanning Coil Assembly 

Pictured below are two views of a typical scanning coil assembly for a modern 110° 
or 1 14° picture tube. You will notice that the amount of flaring on the line coils is 
greater than that on the field coils, for reasons already explained. The two pairs of 
coils are clamped round a two-piece ferro-magnetic core which fits round the neck of 
the tube and is shaped to follow the degree of flaring in the coils. 

Picture Correction Magnets 



Field Coil 




Clamping 
Ring 



Coils 




Picture 

Centering 

Magnets 



A SCANNING COIL ASSCMBIV 

for a modern Picture Tube 

The coils are wound on special trumpet-shaped formers, and are made of plastic- 
covered enamelled wire. After the winding process is completed, a current is passed 
through each coil sufficient to heat it enough to cause the plastic surrounding the wire 
to melt and fuse together throughout the coil. Once this has happened, the current 
is switched off and the coil is allowed to cool. The result is a firmly bonded coil 
whose exterior surface shows the ridged pattern observable in the photograph. 

The two small picture correction magnets shown are positioned one on either side of 
the picture tube cone, each supported by two thin strips of aluminium. The magnetic 
fields they produce can be made to add to, or subtract from, the extremities of the 
field generated by the line scan coils, so compensating for the imperfections which 
sometimes impair the desired shape of the scanning waveform. The picture correction 
magnets thus assist the linearity coils (of which more anon) in straightening up the 
raster and improving the general efficiency of the scanning coil assembly. 

The magnets are moved either by rotating them round the cone, or by bending the 
supporting strips. Once a good, rectangular raster has been obtained, no further 
adjustment of the magnets should be necessary until the coil assembly itself is changed. 
It is not an operation which is normally carried out by the viewer. 



3.78 [§2| 

Picture Centering 

Vertical and horizontal alignment of the picture on the screen is achieved, as you 
know, by rotating the scanning coil assembly round the neck of the picture tube. 
Imperfections in the scanning system, however, make it additionally necessary to 
provide a special control whose job it is to enable the picture to be positioned sym- 
metrically about the centre of the screen. This picture centering control may take 
several forms, varying basically with the method of focusing the electron beam which 
is being used. 

With the electro-magnetic method of focusing, the control usually takes the form 
of a thin steel disk, capable of being revolved, which is attached to the rearmost 
focusing magnet and is therefore magnetised by it. The field it produces, though very 
weak, is enough to alter the overall direction of the scanning beam sufficiently to centre 
the picture in the screen. 



Magnet 
Housing 



Neck of 
Picture Tube 




T-/ 

Focusing Magnets 



Neck of 
Picture Tube 



ELECTROMAGNETIC FOCUSING 

/ 




Magnet 2 



Magnet 1 



ELECTROSTATIC FOCUSING 

The PICTURE CENTERING CONTROL 

With the electrostatic method of focusing the beam, it is usual for the picture 
centering control to take the form of two thin ring-type magnets. These magnets are 
situated immediately behind, and are supported by, the scanning coil assembly. 
Either magnet can be rotated independently of its neighbour. 

The pattern of the magnetic field produced by the two magnets can alter the direc- 
tion taken by the scanning beam, and therefore the position of the picture on the 
screen. 



521] 3.79 

Linearity Correction 

Since no scanning system is perfect and no method of S-correction faultless, some 
means of correcting imperfections in both the line and field scans is needed. The 
normal means of adjusting the linearity of the line scan is to place round the neck of the 
picture tube, during manufacture, a special pre-set type of control which requires little 
further attention during the life of the receiver. 

A method of linearity control used in earlier receivers consisted of a small coil having 
an adjustable core, connected in series with the scanning coils. A small permanent 
magnet affixed to the outside of the coil gave the core a magnetic bias which caused 
it to approach saturation as the scanning current increased towards maximum. This 
caused the inductance of the coil to vary in sympathy; and appropriate adjustment of 
the core utilised this variation in inductance to improve the linearity of the scan. 

But this method was costly, and introduced losses into the circuit. Moreover, an 
extra damping resistor was needed to prevent undesirable ringing during flyback; 
so a search was made for something better. 

The latest method of controlling linearity consists of two thin metal loops made from 
a cheap copper pressing and stuck to a thin tube made from insulating material. This 
tube is slipped over the neck of the picture tube into a carefully calculated position, 
and fixed there before the scanning coil assembly itself is slid into position partially 
(but not exactly) on top of it, with one copper loop lying almost underneath each coil. 



COPPER FOIL LOOPS MADE 
FROM FLAT PRESSINGS Insulating 

Tube 




\^^ 





Linearity 
Loop Assembly 





Ho* 


r Linearity Correction 

*r m Una Sean 


is Achieved 











The linearity-loop/assembly-coil combination functions as follows. Each loop 
gives rise to eddy currents which interact with the magnetic field produced by the coil 
under which it is lying. By Lenz's law, the fields of these eddy currents oppose the 
fields set up by the coils themselves. The electrons in the scanning beam on their way 
to the screen encounter, first, the field produced by the projecting loops, and then 
shortly afterwards the main field, opposite in direction, produced by the coils. By 
careful calculation of the correct axial position for the loops, the correcting field pro- 
duced by them can be made to linearise the scan. 



3.80 [§2 | 

The Problem of Ion Burn 

When electrons are emitted from the cathode, they are usually accompanied by a 
number of other negatively-charged particles called ions. A negative ion is an atom 
in which an extra electron has been captured by, and locked into, its atomic structure. 
Since the normal atom is neutral in charge — the positive charge of the nucleus being 
exactly balanced by the sum of the negative charges on its orbital electrons — the 
acquisition of an extra electron gives the atom an overall negative charge. 

(Note, by the way, that it is also possible for an atom to acquire a net positive 
charge, when it is known as a positive ion. But since only negative ions are attracted 
by the positive accelerating potentials of the anode electrodes in the picture tube, only 
these ions are of present interest to us.) 

An atom is made up of heavy protons and neutrons, and its mass is many thousand 
times heavier than the mass of a single electron. Negative ions are not therefore 
affected by the comparatively weak deflecting forces which are enough to make the 
electron beam scan the screen of the picture tube. They are indeed accelerated towards 
the screen along with the electrons in the beam itself; but then, too heavy and travelling 
too fast to be influenced by the line or field scanning coils, they charge straight ahead 
and impinge in a group at a single point in the centre of the screen directly opposite 
the cathode from which they came. 

The result is a continual bombardment of the central area of the screen, and the 
rapid build-up of a burn area from 10 to 25 mm in diameter on its face. This appears 
as a dirty brown smear marring the centre of the picture displayed. 

For many years, the problem of ion burn was tackled by a technique of mounting 
the electron gun assembly so that it was pointing just enough off-centre for the 
emitted electrons, and their accompanying ions, to be directed towards the side wall 
of the picture tube. A small magnet (called the ion-trap magnet) was placed round 
the neck of the tube so that its field re-directed the electrons, and the electrons alone, 
towards the centre of the neck of the picture tube. The heavy ions, unaffected by this 
small field, smashed into the side of the tube, where they were either captured by the 
final anode or distributed as a fine shower over the whole area of the screen. 



Ion-Trap 
Magnet 



Heavy 
Negative 
ions i 




Non-Magnetic 
Clamping Pieces 



Magnetic 
Field 



Off-Centred 
Electron Gun 



Scanning 

Beam 

Electrons 



Final Anode 




Adjustment 
Clamping Screw 



Soft- Iron 
Pole Pieces 
(N &S) 



Magnet 
Clamp 



Magnet 



$!wW$:V''W# : & : ''.«&$ 



i 1 ^ 1 '?* ■■v^'^\tf*V : ^ 



|p|ION-TRAiii^|:; 



PI] 



3.81 



The Problem of Ion Burn (continued) 

The problem of ion burn is solved nowadays by another technique whose intro- 
duction represented a significant breakthrough in picture-tube manufacturing tech- 
nology. 

You know that, when the electrons of the scanning beam strike the inside surface 
of the screen, that area of the screen which is being bombarded is made to fluoresce. 
What you may not have realised is that much of the light generated in this way is 
radiated backwards towards the cathode, and not only forwards through the screen. 
Indeed, because of the finite thickness of the screen material and the partial reflection 
from the inside surfaces of the glass face of the tube (try looking out into the darkness 
through a clear glass window from a well-lit room and you will appreciate the re- 
flecting properties possessed by an ordinary sheet of glass), as much as 60% of the 
light is actually radiated backwards. 

This is obviously a formidably wasteful way of operating a picture tube, and the 
need to improve matters led to the invention of aluminising. 

The screens of aluminised picture tubes have a very thin layer (usually only a few 
molecules thick) of aluminium deposited on their inside surfaces. This layer is thin 
enough to be almost completely permeable to the high-energy electrons in the scanning 
beam, but thick enough to be almost opaque to the light emitted from the screen. 
Since only the back of the screen is covered in this way, the aluminised layer acts as a 
mirror to the backward-radiated light and reflects it out of the front of the tube to 
add to that already being radiated in that direction. The resulting improvement in 
light output is astonishing. 

Another advantage conferred by the aluminised screen is the increase in contrast 
it gives to the picture being displayed. With the backwards radiation from the screen 
greatly reduced, the internal reflection which used to be cast by the polished glass 
walls of the picture tube itself can no longer impair the quality of the picture presented 
to the viewer. These reflections tended to throw unwanted extra light from the 
brighter regions of the picture on to the areas intended to be darker, thereby noticeably 
reducing picture contrast. 

Lastly, and most important in the context of ion burn, the aluminised layer offers 
good resistance to ionic bombardment. Although offering minimum opposition to 
the passage of the fast-moving (and therefore energetic) electrons, it offers a much 
more effective barrier to the slower-moving negative ions. The need for an off- 
centred electron gun and ion-trap magnet is no longer felt, and most modern picture 
tubes dispense with them altogether. 

To make certain of adequate protection of the screen in such cases, however, the 
aluminising layer is usually made somewhat thicker than normal, particularly in the 
central areas of the screen. 



The Principle 
of 



Aluminising 
Screen 



Reflected Light 

Direct 
Light 




To Electron 
Gun Assembly 



Fluorescent 
Screen 




TO THE 
VIEWER 



viv 



3.82 [§21 

The EHT Smoothing Capacitor 

On the outside of the flared portion of most modern picture tubes is an external 
coating of Aquadag covering about the same area of the cone as the internal coating 
which forms the final anode. This external coating is electrically isolated from the 
inside coating, but is connected to chassis (earth) by spring contacts. 

The two coatings separated by the glass wall of the tube (glass is one of the most 
efficient dielectric materials) together form a capacitor having a capacitance of some 
2,000 pF. This is the capacitor which, as you saw in the last Section, acts as a 
simple but efficient means of smoothing out the ripples in the EHT voltage applied to 
the final anode. 

The cathode of the EHT rectifier diode is connected to the inner conductive film of 
Aquadag, which thus acts as one of the electrodes of the smoothing capacitor as well 
as forming the final anode itself. 

The Direct-View Picture Tube 

Until about 1964, TV receiver cabinets were fitted with a sheet of thick plate glass 
positioned in front of the face of the picture tube. Its purpose was to protect the 
viewer from flying fragments of glass and metal in the event (which in fact was rare) 
of a picture tube imploding. 

An implosion (as opposed to an explosion) is the violent disintegration of an evacu- 
ated vessel when its outer surfaces are subjected to an atmospheric pressure greater 
than' they can withstand. When a picture tube implodes, air rushes into the tube at 
the point of fracture, and the walls of the tube collapse with such force that fragments 
of glass and metal electrodes are sent flying in all directions. 

The disadvantages of the shield were its cost (plate glass is not cheap — nor is the 
fitting of it), the dust which often accumulated between the plate and the front of 
the picture tube, so reducing the light output of the tube, the increased cabinet depth 
required and poor distribution of cabinet weight. From the technical point of view, 
an even more important disadvantage was the multiple reflections between the surface 
of the shield and the face of the picture tube face. 

In 1964, a new concept in picture tube technology was brought to market. It 
consisted of wrapping a strong mild-steel band round the periphery of the tube face — 
the area which is subjected to the greatest tensile stresses — and bonding the two 
together. Normally, the band is free from 
stress ; but if a break occurs in the tube it 
acts at once in opposition to the atmos- 
pheric pressures, and serves as a support 
to the screen and walls of the tube. 

An additional advantage of the metal 
band is that it can be fitted with four 
fixed lugs, and so be used to anchor the 
picture tube to the cabinet. This is a 
much neater arrangement than were the 
complicated fixing methods needed earlier. 
A thin plastic mask slipped over the peri- 
phery of the tube face protects the corners Th A/A//*r VIEW 
of the faceplate when it is tightened ,ne *"***'' • r '** r 
against the cabinet and, being flexible, PJCtllPG TllbG 

ensures a good fit. 




§21] 3.83 

Test Card "F" 

You must have seen the picture below on the screens of scores of TV receivers 
displayed in retailers' windows, and you will assuredly have obtained it on your own 
screen if you have ever switched on outside normal broadcasting hours. Do you 
know what it is? 




«C» 



TEST CARD "F 



::::;:>: ;:: : :.: : : : : : : : ::;:; : :; 



Test Card "F" is a white card bearing the photograph of a very young lady playing 
noughts and crosses on a blackboard, and surrounded by an odd-looking pattern of 
circles, lines, rectangles, diagonals, triangles and squares. It is used by both the BBC 
and the Independent Television Authority, on both line standards, and for both 
colour and black-and-white receivers. The card itself is in a miscellany of colours 
(though you see it above as it appears on a black-and-white-only receiver screen). It 
is placed before one of the studio cameras and transmitted outside normal programme 
hours to help the service engineers, or the viewer, to adjust the various controls on the 
receiver so that they are set to give a stable, well-defined and properly proportioned 
picture. 

(The age of the young lady chosen as the model, by the way, was governed by the 
possibility that changing fashions in make-up used by a more grown-up rival could 
confuse a viewer trying to assess the colour quality of the picture he was receiving!) 

The dimensions of the card provide the correct aspect ratio of 4:3; and its edges 
carry a series of "castellations" which, in monochrome, shade off from black through 
various shades of grey to white. Look at the top of the test card and observe how the 
first few lines of it are made to consist of half-tone gradations varying from peak 
white in the top left-hand corner to black in the top right-hand corner. These 
gradations appear as separate colours on a colour receiver and are intended for use 
with such a receiver only, to help in colour assessment. 



3.84 [§2 | 

Test Card "F" (continued) 

When the test card is being used, the viewer should adjust the Width, Height and 
Picture Centering controls on his set until the card is centrally positioned on the 
screen, with the centre points of its sides just reaching the edges. Note that because of 
the slight curvature of the screen and its mask, parts of the card may extend beyond the 
screen and therefore be out of sight. 

The Vertical (Field) and Horizontal (Line) Linearity controls are next adjusted 
until the circle in the centre of the card is correctly shaped — though the horizontal 
control, an internal one, will normally be adjusted by a service engineer only. More- 
over, since some of the controls (of which Height and Vertical Linearity are examples) 
are so inter-related that adjustment of one will often demand adjustment of the other, 
the correction of picture linearity can be a skilled operation. 

Focus should be set for overall sharpness of picture detail rather than for perfect 
focus at any given point. This is particularly important in picture tubes using the 
electro-magnetic focusing technique, for this method is rather less effective over large 
areas than is the electrostatic method. Uniformity of focus is best achieved by 
observing the diagonal areas of black and white stripes in each corner, and the picture 
in the centre of the card. 

Contrast and Brightness are set by watching the column of six half-tones situated 
to the left of the centre circle. The overall contrast range of the column is about 
30:1, and a properly adjusted receiver should make every half-tone visible with a 
constant difference in brightness between each. The small lighter-shaded spots in 
the top and bottom half-tones in the column must be clearly visible, with no tendency 
to merge into surrounding areas. 

The presence of reflections of the received signal caused by hills, large buildings, 
etc. near the receiver is often revealed by "ghost" images on the test card. These 
generally take the form of displaced images of the black and white vertical lines, 
particularly where they are close together. The noughts and crosses on the black- 
board are especially good "ghost detectors". 

Line synchronisation is checked by observing the castellations running down the 
right-hand side of the test card. Faulty synchronisation shows up as a horizontal 
displacement of those parts of the picture which lie on the same level as the white 
castellations, and it also makes the central circle look like a cog-wheel. 

The resolution and bandwith of the receiver are adjustable with the aid of the 
manufacturer's instructions and suitable test equipment (signal generator, test-meter, 
etc.), and by observing the column of six rows of gratings which appear to the right of 
the central photograph. (Note, however, that this is a major operation seldom 
necessary during the life of the average receiver.) Every row of gratings consists of a 
number of vertical stripes, the spacing between them representing a particular funda- 
mental frequency. The stripes are equivalent to square-wave signals whose amplitudes 
vary from black to peak white, and the fundamental frequencies (in MHz) which they 
represent are as follows, reading from top to bottom: 

On 625-line: 1-5; 2-5; 3-5; 4-0; 4-5; 5-25. 
On 405-line: 1-0; 1-6; 2-25; 2-6; 2-9; 3-4. 

Lastly, the low-frequency response of the receiver can be checked by reference to 
the black rectangle lying within the white rectangle at top centre of the card. Poor . 
low-frequency response shows up as streaking at the right-hand edges of both these 
rectangles — as well as on the castellations running round the borders of the card. 



§21] 

REVIEW of the Picture Tube 



3.85 





The modern TV picture tube has a nearly 
rectangular screen. Its size is specified by 
quoting the length of its corner-to-corner 
diagonal. Typical present-day sizes are 
17% 19" and 23". 



The deflection angle (nowadays normally 
110°) is measured from a point known as the deflection centre, located close to the 
beginning of the flared section of the tube. A wide deflection angle makes it possible 
for the neck section of the tube to be kept short, so reducing the overall front-to-back 
length of the tube. 




n*mmc 
ELECTROSTATIC LENS 



Focusing of the scanning spot on the 
screen is usually achieved by electrostatic 
means. The difference between the poten- 
tials on the first and second anodes of the 
electron gun assembly is made variable, the 
object being to form the shape of the field 
which exists between them into an electro- 
static lens. 



l^Mb-^ 1 



The four coils which together make up the 
line and field scanning coil assembly are 
located around the neck of the picture tube, 
close to the beginning of the flared section 
and extending round the first part of it. The 
coils are connected in pairs, and are so 
arranged that the line (horizontal) pair is 
situated above and below the scanning 
beam, whereas the field (vertical) pair is 
situated on either side of it. 



Every coil, together with its core material, 
forms an electromagnet; and both pairs of coils are electrically connected in such a way 
that at any point in time during the scanning period one coil of a pair is operating as a 
North pole and the other coil of the pair is operating as a South pole. 

As the value of the current flowing through a coil pair varies from maximum in the 
positive direction, through zero, to maximum in the negative direction, so the polarity of 
any given coil changes from North to South, or vice versa; and the electron beam is 
diverted up and down the screen, or across it, accordingly. 




3.86 

REVIEW of the Picture Tube (continued) 



K2I 



COPPER FOIl LOOPS MADE 
FROM FLAT PRESSINGS 

W/ <&J 




Loop Assembly 



L JMBrity C ott9ttiOB 

«r on lin Sen h AoMmri 



Linearity control of the line scan is 
exercised by means of two closed-loop coils, 
fabricated from a thin copper-foil pressing 
and so positioned beneath the scanning coil 
assembly that eddy currents are induced in 
them by fields produced by the line scan coils. 



These eddy currents give rise to magnetic 

fields which oppose those of the line coils 

themselves; and by careful positioning of the copper loops with respect to the coils, the 

degree of opposition can be made to compensate for any non-linearity which may be 

present in the raster produced by the coils. 



The technique of aluminising the inside 
surface of the picture-tube screen makes 
possible a large increase in the total light 
output from the front of the screen. Dis- 
tortion of the picture caused by multiple 
internal reflections is reduced, and a much 
greater contrast range is achieved. 



ALUMINISING 




Aluminising also protects the screen from ion burn, and obviates the need for an 
off-set electron gun and ion-trap magnet. 



§22: VISION INTERFERENCE 



3.87 



Most viewers will be familiar with the effects of electrical interference on a TV 
receiver. Unsuppressed car ignition is a major cause of trouble, the pulses of inter- 
ference recurring at regular intervals determined by the speed of the engine. Its 
effect is to cause bands of white or black spots to move up and down the picture, 
and to cause exhaust-type noises to accompany the normal sound from the 
loudspeaker. 

Another form of interference is caused by arcing at the brushes of some types of 
electric motor, e.g., in some drills and electric shavers. This type of interference is 
continuous, and causes black or white spots to appear over the entire picture. It 
is usually accompanied by an intense hissing noise from the speaker. Interference of 
the same general kind can be caused by arcing at the contacts of thermostats in electric 
irons, etc., although this type of interference is usually periodic. 

Car ignition interference is transmitted to the receiver directly as an r.f. wave 
from the spark plug or distributor leads, and should be suppressed at these points by a 
suppressor generally consisting of a single resistor (of, typically, 10 k value) connected 
in series with the coil or distributor lead. Interference from electric drills, etc. may 
be radiated either directly from the point of arcing or indirectly by way of the mains 
supply which feeds both the TV receiver and the offending apparatus. Such inter- 
ference also can be easily suppressed at source. 

Unfortunately, it is frequently not so suppressed; and most TV receivers incorporate 
some means of reducing the worst effects of sound and vision interference. The 
circuits which do this are known as sound and vision interference limiters. You learnt 
about sound interference limiters in Section 14. Now for their vision counterparts. 

The Vision Interference Limiter 

The first thing is to understand how vision interference manifests itself in the 
voltage waveforms which control the operation of the dual-standard receiver. The 
illustration below shows a few lines of a 405- and a 625-line vision signal immediately 
after detection. 

_ White (100% Modulation) 



V 






Black 



Sync 



405-LINE 




Sync (100% Modulation) 



- - Black 



- White 



625-LINE 



INTERFERENCE PULSES 



Both waveforms are distorted by strong interference pulses, but note the important 
differences. 



3.88 [§22 

The Vision Interference Limiter (continued) 

In the 405-line signal, the interference pulses cause an increase in the amplitude of 
the vision carrier. Because the 405-line system employs positive modulation of the 
vision carrier, this causes an increase in the picture signal content of the video wave- 
form, driving it into and beyond the peak-white region and so causing white spots to 
appear on the picture tube. The effect is made worse by the fact that these pulses 
overdrive the picture tube and cause partial loss of focus. This causes the white spots 
to enlarge into unfocused blurs, so exaggerating the initial distortion and making it 
even more objectionable. Note, however, that the sync pulse content of the video 
waveform is not affected by the interference — a most important point. 

In the 625-line waveform, the interference pulses again tend to increase the ampli- 
tude of the vision carrier; but since this system employs negative modulation, they 
serve to reduce the amplitude of the picture signal content of the video waveform, and 
extend into and beyond both the black level and the sync pulse level. The interference 
pulses therefore cause small black spots to appear on the picture tube, which are much 
less objectionable than the large white spots produced in the 405-line system; but the 
intrusion of the interference spikes into the sync pulse region is serious. In bad 
cases, it can impair the synchronisation of the receiver and strain the ability of the 
flywheel sync circuit to keep this essential function going. 

Vision Interference Limiters — 405-line System 

A form of vision interference limiter much used in 405-line receivers consists of a 
diode connected either across the output circuit of the video amplifier or, sometimes, 
across the cathode circuit of the picture tube. 

In the former case, the cathode of the diode is connected to the anode of the VA, 
and the anode of the diode is taken to a variable resistor forming part of a potential 
divider connected across the HT supply. This resistor enables the limiting action of 
the diode to be controlled by the viewer, whose action adjusts the positive potential 
applied to the anode. 



HT (+) 



Video 
Amplifier 



Limiting 
Level - - 





405- Line Video Signal 
to Picture Tube 



An Adjustable VISION INTERFERENCE LIMITER 



§22] 3.89 

Vision Interference Limiters — 405-line System (continued) 

The anode potential of the diode on the last page is normally set a little below the 
level of the (negative-going) video signal representing peak white. Under normal 
operating conditions the diode is reverse-biased and therefore non-conducting; but 
when an interference pulse appears superimposed on the video signal, of sufficient 
amplitude to extend beyond peak white level, the cathode of the diode will become 
negative with respect to its anode, and the diode will conduct. 

When this happens, the capacitor connected between the diode anode and chassis 
is placed across the video amplifier, and the negative spike of the interference pulse 
is absorbed as its charging current. 

In this way, the amplitude of the interference pulse is limited to the operating level 
of the diode. The limiter diode thus makes no attempt to eliminate the interference; 
it merely restricts it to a pre-set level. In doing so, it prevents the picture tube from 
being overdriven, so preventing the white spots produced by the interference from 
becoming defocused (and highly objectionable) on the screen. 

One obvious disadvantage of this particular circuit is that the limiting action 
depends on the Contrast setting of the receiver. If the operating level of the limiter is 
set close to the peak-white level of a particular video signal and the contrast is then 
increased, the limiting action will take place earlier, and part of the picture signal itself 
will be clipped. If, on the other hand, the limiter is set to operate at a higher level, 
a correspondingly greater level of interference must be tolerated. 

The only other solution, of course, is for the viewer to reset the limiter control 
potentiometer every time he operates the Contrast control. This he does by rotating 
the control until the picture begins to turn milky-white, and then turning it back again 
until the picture just returns to normal. 

Many circuits have been devised to overcome the disadvantages of the simple 
limiter circuit described, some of them completely self-adjusting in operation. Cir- 
cuits of this type use the peak-white level of the video waveform to provide the anode 
potential for the limiting diode, usually with the aid of a simple RC integrating circuit. 
Whenever an interference pulse arrives, the diode conducts because its anode potential 
cannot change fast enough, and the interference pulse is short-circuited through the 
diode. When the Contrast setting is altered, the integration circuit supplying the 
diode anode re-adjusts automatically to the new peak-white level. 



405-Line Video Signal 

from 

Video Amplifier 




Limiter 
Diode 



k 

mm 

HiTi*f£*£HC£ 



A disadvantage of this type of circuit (an example of which is shown, with the limiter 
diode this time connected across the cathode circuit of the picture tube) is that, in 
order to maintain charge on the capacitor C, the diode must conduct slightly on the 
whitest part of the video signal. The result, of course, is that the quality of the 
highlights of the picture will be somewhat impaired. 



3.90 



[§22 



405-line Vision Interference Limiting — The Black Spotter 

The best it is possible to do with the 405-line vision interference limiter is to restrict 
the interference pulses to a level not much greater than peak white; but this does not 
prevent the white spots from appearing on the screen. This is where the black spotter 
comes in. 

A black spotter is a circuit which accepts those interference pulses which exceed 
peak white, amplifies them, and then applies them to the picture tube in opposite phase 
to that of the video signal itself. In this way the white spots are converted into black 
spots, without affecting picture focus; and the interference becomes much less notice- 
able. In some circuits, the amplification of the spotter (and therefore the amplitude 
of the anti-phase interference pulses) is made adjustable so that the black spots can 
be made to show up as grey — thus becoming even less noticeable. 

There are two basic methods of black spotting. In the first, the interference pulses 
which exceed peak white are amplified, inverted and applied (now positive-going) to 
the cathode of the picture tube along with the normal negative-going (and noise- 
carrying) video signal. This causes the picture to black-out every time an interference 
pulse occurs. 

In the second method, the interference pulses are amplified but not inverted, and 
are then applied (still negative-going) to the grid of the picture tube — the video signal 
being applied to the cathode of the tube as usual. As you know, putting a negative 
potential on the grid has precisely the same effect as putting a similar positive potential 
on the cathode; so the result of this type of connection is the same as that of the first. 
A black spotter circuit based on this second method is shown in the illustration below. 




HT(+) 



Video 
Amplifier 



C1 



Spotter 
Control 

500 K I 



I 



J 



£> 



A 

\ 



Picture Tube 



The BLACK SPOTTER 



§2Z] 3.91 

The Black Spotter (continued) 

The spotter itself in the circuit shown on the last page is a triode valve functioning 
as a grounded-grid amplifier. Its input signal is applied to its cathode directly from 
the anode of the video amplifier, and its grid is earthed, insofar as a.c. is concerned, by 
the capacitor Q. 

The triode is normally reverse-biased, by a positive potential set on its grid by the 
spotter control, to a voltage slightly lower than the voltage which represents peak 
white at its cathode. Thus the valve can only conduct at times when the signal 
extends beyond peak white. At all other times its presence has no effect at all on the 
picture signal. Whenever an interference pulse causes the video signal to exceed 
peak white, however, the cathode of the spotter becomes negative with respect to its 
grid, and the valve passes anode current for the duration of the pulse. This causes an 
amplified version of the pulse to appear across the anode load (R^), which is applied 
to the grid of the picture tube through the coupling capacitor C 2 . 




■ HT(+) 

CATHODE 
OF SPOTTER 

Spotting 
Threshold 

- OV 



WmtMm 



HT(+) 



ANODE OF SPOTTER 

and Grid of Picture Tube 



It is theoretically possible, by careful adjustment of the spotter control for a given 
contrast setting, to make the amplitudes of the interference pulses fed to grid and 
cathode of the black spotter exactly balance each other — thereby achieving perfect 
cancellation. In practice, however, adjustments of this sort can seldom be maintained 
for long because the unpredictable and varying nature of the interference calls for 
constant contrast adjustment. Black spots are therefore usually accepted as the 
lesser of two evils. 



The 625-line Vision Interference Limiter 

Interference pulses on the 625-line video signal cause, as you know, only black 
spots to appear on the picture tube and are therefore much less objectionable than are 
similar pulses on the 405-line signal. There is obviously no need for black-spotting, 
and all that need be done as far as picture quality is concerned is to limit the inter- 
ference pulses to the level of the sync pulse tips, so as to avoid overloading the video 
amplifier. This can be done by simple limiter diode circuits like that used for the 
405-line signal. 



3.92 



[§22 



The 625-line Vision Interference Limiter (continued) 

What is really important, however, is the effect that 625-line interference pulses 
could have on the synchronisation of the receiver. These pulses, as you know, extend 
into the sync pulse region of the video signal and so could cause faulty triggering of the 
scanning generator circuits. This would be particularly serious if the interference was 
sustained, because the flywheel sync circuit would then be unable to maintain control. 

A circuit which does much to overcome the problem uses the interference pulses 
themselves to control the conduction time of the sync pulse separator valve. When 
controlled in this way, the separator is said to be noise-gated. 




91 Waveform 



The HOISi-GAUD 

Sync Pulse Separator 



93 Waveform 



Anode 
Waveform 



The valve used in the circuit is called a pentagrid, or heptode. It has seven electrodes 
— a cathode, an anode and five grids. Grids 1 and 3 are control grids, isolated from 
one another by two screen grids (2 and 4) which are linked together internally. Grid 5 
is the suppressor grid. 

Interference pulses of greater amplitude than the tips of the sync pulses are amplified 
and inverted with the aid of a biased amplifier such as that used for black-spotting, 
and are then applied to Grid 1 of the sync pulse separator. Their amplitude is sufficient 
to cut off anode current flow in the valve whenever they occur. 

The video signal, together with its interference if present, is applied to Grid 3, 
and its sync pulse content is removed from the picture signal in the usual way by grid 
current flow. But whenever an interference pulse occurs, the valve is immediately 
cut off by the negative pulse reaching Grid 1, so the positive-going interference pulse 
on Grid 3 is effectively neutralised. (In practice, the mutual cancellation is somewhat 
less than perfect and a small residual interference pulse often occurs at the anode, but 
it is too small in amplitude to cause any trouble.) 

Note that Grid 1 is taken to a small positive potential of about 1 V so that the valve 
is forwarded-biased in the absence of an interference pulse. 

Some circuits carry a further refinement. They apply the negative gating pulse to 
Grid 3 through a variable resistor so that its amplitude can be adjusted for best 
cancellation of the interference. 



§23: AUTOMATIC GAIN CONTROL 



3.93 



The purpose of automatic gain control (AGC) in a TV receiver is to maintain the 
amplitudes of the sound and video signals applied to the loudspeaker and picture tube, 
respectively, at a pre-determined level set by the viewer by operating the Volume and 
Contrast controls. 

To do this, the AGC circuit must continuously monitor the amplitudes of the 
received sound and vision r.f. carriers, and so control the gain of all the sound and 
vision sections of the receiver that the gain of any section is automatically increased 
when the received signal is low, and automatically reduced when the received signal 
is high. To be perfectly effective, the response time of the AGC circuit should be 
short — so that, in addition to responding to gradual signal variations such as fading, 
it is also capable of reacting quickly to any rapid change in the strength of the received 
signal — as may occur when the set is switched to another channel. 

Most AGC circuits control the gain of a TV receiver by varying the bias voltages 
applied to the^grids of either or both of the r.f. or i.f. amplifying stages; and they do so 
in much the same way as AVC voltages are used to control the gain of a broadcast 
radio receiver. 

You might think at first that a single AGC circuit, functioning from either the sound 
or the vision signal, would be all that is needed to control the common level of both 
sound and vision signals in a TV receiver. This would indeed be true if you could be 
sure that both sound and vision carriers— two completely separate signals of different 
frequency — were always affected during transmission in exactly the same way, and 
that their relative amplitudes on leaving the transmitter were always the same. In 
practice, of course, you could seldom be sure of either. 

Nevertheless, some relatively low-priced receivers do indeed operate on this 
principle, controlling the overall gain of both the sound and vision sections of the 
receiver by means of an AGC voltage derived from the strength of the vision carrier. 
Note that the common AGC in such receivers is never derived the other way round — 
from the strength of the sound carrier. This is because the human eye is much less 
tolerant of distortion in a reproduced image than is the human ear of distortion in 
reproduced sound. (If you want proof of that statement, consider all those young 
people who cheerfully accept atrocious distortion on their portable transistor radios, 
but would object strongly to a fraction of such distortion on the displayed picture of 
their favourite TV programme). 

When separate sound and vision AGC circuits are employed, it is usual for the 
vision AGC to be made to control the gain of the tuner and of any common i.f. 
stages, in addition to controlling the gain of the vision i.f. stage. The principal reason 
for this is to preserve the quality of the picture signal; but this method of operation 
also exerts some degree of control over the sound signal, and so gives the sound AGC 
circuit less work to do. 

AGC circuits usually measure the amplitudes of the two r.f. carriers either at the 
sound and vision detector outputs (where you will remember that the output waveforms 
are caused to be exactly proportional to the carrier amplitudes) or (in the case of 
vision AGC) at a point immediately following the video amplifier. Various methods 
of AGC will be described later in this Section. 



3.94 



[§23 



A Typical AGC Layout— 405-line System 

The block diagram below shows the layout of a typical 405-line AGC system in 
which separate sound and vision AGC circuits are employed. Note, nevertheless, 
that even here the sound AGC circuitry lies "within the control loop" of the more 
important vision AGC circuit. 



Video 
Amplifier 



Vision 
AGC 



Aerial 

N/ 



Vision AGC 
Voltage -^ 



I— Tuner 



Picture 
Tube 



A 



Vision 
Detector 



Vision i.f. 



Common i.f. 




ileal 

AGG SYSTEM 

for a 



Sound i.f. 



Sound AGC 
Voltage 



Sound 
Detector 



Sound 
AGC 



Loudspeaker 



A.F. 
Amplifier 



^ 



AGC of the Sound Carrier 

Only in the 405-line system, of course, is AGC for the sound carrier required ; for 
the 625-line sound carrier is frequency-modulated, and carefully limited to a constant 
amplitude before detection. 

In the 405-line system, where the sound carrier is amplitude-modulated, AGC is 
usually derived from the d.c. output of the sound detector. This voltage is then used 
to control the gain of the sound i.f. amplifying stages in the manner described in Sec- 
tion 13, when the functioning of IF Amplifier B of the British Dual-Standard Receiver 
was discussed. 



§m 



3.95 



AGC of the Vision Carrier 

One of the biggest problems associated with vision AGC in any system is that of 
obtaining an accurate indication of the amplitude of the received vision carrier. It is 
no good trying to measure the amplitude of the actual r.f. carrier before detection, 
because its average value is always zero (as is the average value of any truly a.c. 
waveform). Nor can a satisfactory indication be obtained by measuring the ampli- 
tude of the detected modulation (as is done in sound AGC systems), because of the 
peculiar nature of the modulation. As you know, video modulation of the vision 
carrier is either wholly negative (as in the 625-line system) or wholly positive (as in the 
405-line system). 

The difficulty is best realised by considering the illustration below, which shows a 
few lines of the video modulation of the same (405-line) carrier at three different 
instances in time. At A, the picture signal content of the modulation is predomi- 
nantly white — as it would be if a sunlit snowscape were being televised. At B, the 
picture signal is predominantly black, as it would be if a night scene were being shown, 
or if no picture signal at all were present during a slow scene change or pause. The 
video waveform at C is typical of that representing a complex scene containing widely 
contrasting tones, e.g., a snow scene containing dark shadows thrown by sunlit 
trees. Here the average value of the video waveform is dark grey. 

In all three of the cases illustrated, the average value of the video waveform is quite 
different. Which of them is the correct one to use ? 

Nevertheless, despite this demonstration that perfect results cannot be obtained in 
the estimation of the amplitude of the vision carrier by measuring the average value 
of its video modulation, AGC circuits which rely on just this method are in fact 
widely used in the cheaper TV receivers— especially those designed for 405-line 
reception only. The reason is their extreme simplicity. 



T 



Picture 
Signal 

_1 



Sync 




100% (White) 
Average Value (Light Grey) 

30% (Black) 

— 



O 



1" 

Picture 

Signal 

Sync I 



100% (White) 




30% (Black) 




Average Value 
(Blacker than Black) 



© 




100% (White) 



— Average Value (Dark Grey) 
30% (Black) 







3.96 



[§23 



Mean-level AGC 

Apart from its simplicity, the main technical argument in favour of using the 
average (mean) level of the de-modulated video waveform to produce an AGC 
voltage is the fact that abrupt changes in tonal content between scenes are compara- 
tively rare — save sometimes when changing channels, in which case it is reckoned that 
the viewer will accept some adjustment of the Contrast control as an acceptable chore. 
Moreover, the method is equally suitable for both 405- and 625-line systems, without 
the need for special switching. 

Much the most widely-used method of producing a mean-level AGC voltage is to 
make use of the negative voltage appearing at the grid of the sync separator valve. 
You will recall that this voltage is caused by leaky-grid action when the positive-going 
sync pulses of the video waveform from the anode of the video amplifier are applied 
to the grid. The greater the amplitude of this waveform, the greater the magnitude of 
the negative voltage produced. It follows that the average value of this voltage gives 
an indication of the mean amplitude of the video waveform, and therefore of the 
average magnitude of the vision carrier (bearing in mind the limitations just discussed). 

Since no polarity inversion is required, all that is needed before this negative voltage 
can be used to control the gain of the r.f. and i.f. stages is to find some means of 
smoothing the variations in its amplitude. This is done with the aid of a simple 
integration circuit (low-pass filter) such as that formed by Q-Rj in the illustration 
below. 

HT(+) 



How a mmitm 

AGC VOlTACt is Produced 



(+)- 



Video Signal 
from Video 
Amplifier 




OV 



Negative 
AGC Voltage 

The time constant of the integration circuit, C x x Rx seconds, needs to be at least 
as long as five line periods lest the AGC voltage developed across d should progressive- 
ly leak away between the intervals of being "topped-up" by the next succeeding sync 
pulse. In practice, a much longer time constant is required because, during the 
period of the field sync pulse, the picture signal content of the video signal is sup- 
pressed for at least 1 5 lines (20 lines in the 625-line system) — during which time the 
AGC voltage, if left to itself, would fluctuate considerably between each field pulse 
period, with unsteadying effect on the gain of the receiver. 

For this reason, the time constant of the integration circuit is generally made equal 
to about fi\z field periods, typical values being 100-200 ms. Taking the components 
Ci, Ri shown in the illustration, their time constant is 0*05 f*F x 2-2 M = 0-1 1 seconds, 
or 110 ms. (Remember that microfarads x megohms give a time constant measured 
in seconds.) 



§23] 



3.97 



Delayed AGC 

All AGC circuits suffer from the disadvantage that a negative control voltage is 
produced from all signals received, not only from the strong ones. The trouble about 
this is that as much gain as possible is needed to boost weak signals reaching the r.f. 
and i.f. stages, and anything which controls or limits this gain is undesirable. Since 
much less gain is required for the stronger signals, some form of lag or delay is clearly 
required to prevent the AGC circuit from operating until the received signal has 
exceeded a certain threshold. Once this level has been exceeded, the circuit can come 
into play to control receiver gain in the normal way. 

A popular way of introducing a delay into an AGC circuit is to connect a diode 
across the AGC line with its anode connected through a load resistor to a positive 
potential, such as HT( + ). When weak signals are being received, the negative voltage 
built up on the AGC line is quite small; and the reverse bias which this voltage 
applies to the diode anode is much less than is the forward bias offered by the HT 
line through the anode load. The diode therefore conducts through its anode load 
and places a near-short-circuit across the AGC line, holding its potential at a little 
above zero. The lack of a negative AGC voltage therefore allows the r.f. and i.f. 
amplifiers to operate at maximum gain — which is what is required for the weaker 
signals. 

HT(+) 

iiiiiiiiiiiiiiiiiiiiiiiiiiiiiiiii» 



DELAYED 
AGC 



Video 



Sync 
Separator 




Negative 
AGC Voltage 

As the strength of the received signals increase, however, the negative AGC voltage 
becomes progressively more effective in its opposition to the forward bias offered by 
the HT line, until it is sufficient to cut off all current flow through the diode. The 
diode then loses its delaying effect, and the AGC potential resumes control of the 
amplitude of the signal passing to the r.f. and i.f. amplifiers. 

The presence of the diode serves another useful function in ensuring that the AGC 
line can never rise positively above zero, as it might do if no signals were present at 
the grid of the sync separator. Should a positive voltage ever be applied to the grids 
of the amplifying valves, they would at once pass larger-than-normal anode currents ; 
and damage to themselves or their associated components could occur. In this 
particular role, the diode is said to function as a clamp. 

The purpose of the potential divider R 2 -R 3 at the grid of the sync separator is to 
attenuate the AGC voltage before it is applied to the stages whose gain is to be 
controlled. The point of this is that the amplitude of the AGC voltage created at 
the grid of the sync separator is often too great to be fed direct to the valves to be 
controlled. This is particularly so in the case of modern high-gain (high slope) valves 
which have a restricted grid base, and are therefore capable of accommodating only 
small changes in the amplitude of signals reaching their grids. 



3.98 



[§M 



Double-delayed AGC 

The maintenance of a good signal-to-noise ratio within a TV receiver makes it 
important that the r.f. stages — in particular, the mixer stage in the tuner — should be 
operated at a fairly high level of signal. For this reason, it is usual in a modern TV 
receiver to provide a second delay in the AGC circuit, operating for some time after 
the first one has ceased to be effective. 

The first delay is generally introduced into the circuit immediately after the point 
at which the AGC voltage is produced, as you have just seen. The second is inter- 
posed between the first delay and the r.f. amplifying stages. The simplified arrange- 
ment shown in the illustration works as follows. 



DOUBLE 
OCLAYS 

in the 

AGC 
glftCUi? 

AGC 



to r.f. Stages 




Sync 
Separator 



AGC 
to i.f. Stages 



Say that a signal is being received which is just sufficient to overcome the delay 
potential offered by the diode D 2 . A negative AGC potential is built up as this 
happens, and appears across the capacitor C 2 (C 2 and R 4 forming the integration 
circuit already described). A negative AGC potential is applied to the i.f. stages, 
thereby reducing their gains. 

The same AGC potential now has to overcome a similar delay potential offered 
by Di (the purpose of R 2 is to isolate Delay 1 from Delay 2). Until this is achieved, 
the i.f. stages only are reduced in gain and the r.f. stages are permitted to operate at 
maximum gain. But should the strength of the received signal continue to increase, 
the AGC potential developed across C 2 will eventually become large enough to over- 
come the delay offered by D x also. A negative AGC potential will be built up across 
Q and applied to the r.f. stages, reducing their gain accordingly. 

Should the strength of the received signal now fall, AGC will be removed from the 
r.f. stages first, and only if it continues to fall further, from the i.f. stages as well. 

When two AGC delays are employed, it is common practice to provide means of 
rendering the second delay inoperative in case the receiver is to operate in an area of 
very high signal. This is generally effected by a wire-and-plug arrangement set by a 
TV dealer — the second delay being effectively removed altogether by disconnecting 
its anode resistor R ± from the HT supply. This causes r.f. gain to be reduced at the 
same time as i.f. gain is reduced. 

If, however, the receiver is to be used in a fringe area where much higher r.f. gain is 
required, the delay is re-connected into circuit by again connecting the diode load to 
the HT supply. 



§23] 



3.99 



The Contrast Control 

A component which in many less expensive TV receivers is incorporated in the 
mean-level vision AGC circuitry is the contrast control. It takes the form of a 
potentiometer (typically, of some 500 k) connected between HT( + ) and the anode 
of the delay diode (or, if there are two such diodes, of the first of them). The slider 
of this potentiometer is activated by the viewer manipulating the Contrast control 
knob on the outside of his receiver. 

You know that the upper limit of the signal reaching the Video Amplifier is deter- 
mined by the value(s) of the potential(s) on the delay diode(s) in the AGC circuit. 
Once the incoming signal starts to exceed this value, the AGC control potential in- 
creases in proportion, and immediately reduces the gain of the r.f. and vision i.f. 
stages to compensate for the attempted rise in signal amplitude. 

If, then, the AGC delay potential is increased (made more positive), the incoming 
signal will be allowed to reach a higher level before it triggers off AGC action. This 
means that a larger video signal will be applied to the picture tube, and there will 
therefore be an increase in the contrast of the picture being displayed. Conversely, 
if the delay potential is reduced, AGC action will be initiated at a lower signal 
amplitude, a smaller video signal will be applied to the picture tube, and the contrast 
of the picture will be reduced. 

It follows that a simple method of allowing the viewer to control picture contrast 
is to enable him to adjust at will the positive voltage applied to the (first) delay diode. 
In the circuit illustrated below, contrast will be increased {i.e., video signal will be made 
larger) as the slider on the potentiometer is moved towards HT( + ), thus making the 
voltage applied to the diode more positive 



Contrast , 

500K 



Increase 



-4n 



1 



Video 
Signal 



,05 



1M ^ 




HT(+) 



Sync 
Separator 



H 



2-2M 



Delay 
Diode 



1111 

VoSWIm 





CONTROL 



Disadvantages of this type of AGC contrast control are, first, that it affects the 
amplitude of the video signal fed to the video amplifier, and therefore that of the signal 
fed to the sync separator. This means that it could in certain circumstances impair 
picture synchronisation. A second disadvantage is that, in varying the gain of the r.f. 
stages, the contrast control also varies the amplitude of the signal fed to the sound i.f. 
stages. If the sound AGC circuit is unable to accommodate these variations, the 
Volume control will have to be adjusted every time the Contrast control is used. 



3.100 [§23 

The Contrast Control {continued) 

Neither of these disadvantages, of course, applies to the high-level type of contrast 
control which you learnt about in Section 15 on the Video Amplifier. This control in 
no way affects either the r.f. or the i.f. gain of the receiver, but operates (you will recall) 
to maintain the video signal at a near-constant amplitude at the point where it is 
applied to the grid of the sync separator. This method of achieving contrast control 
is, however, more expensive than is the simpler, though less efficient, method just 
described. 

Note that in some receivers an internal preset "sensitivity" control may be found 
incorporated into the tuner, to set the maximum gain of the receiver. This type of 
control is particularly useful in areas of high signal strength to "take the weight off" 
the AGC circuits and to ensure that they are always operating within the range they 
can handle. 

A more general point about contrast controls is perhaps worth making. Since any 
control capable of varying the gain of either the r.f. or the vision i.f. amplifying 
stages will automatically affect the amplitude of the video signal applied to the picture 
tube (and hence the tonal contrast of the reproduced picture), it would seem at first 
sight that the contrast control circuitry could be put almost anywhere in the video or 
vision sections of the receiver. This is not so, however, because if such a control were 
used, e.g., to control the gain of the mixer stage in the tuner, both the vision and 
sound AGC circuits would react immediately to every variation of the control by 
trying their best to maintain the amplitude of the incoming signal constant. Since no 
AGC circuit can distinguish between wanted and unwanted variations in the signal 
reaching it, any form of contrast control which affects the gain of the r.f. or i.f. stages 
of the receiver must either work in conjunction with the AGC circuits or else be 
situated at a point in the circuit after the r.f./i.f. stages have done their work. This 
latter, of course, is the case with the efficient high-level contrast control situated in the 
Video Amplifier. 

Mean-level AGC— A Typical Circuit 

The illustration on the opposite page shows the essential components of a mean- 
level vision AGC circuit for a modern TV receiver (it is, in fact, the Baird 620). It 
will be seen that the AGC potential is here derived directly from the anode of the video 
amplifier instead of from the grid of the sync separator; and also that the "diode 
action" typical of the grid and cathode electrodes of the sync separator valve is 
performed instead by a separate triode connected to function as a diode, with its 
anode and grid electrodes strapped together. This valve is labelled V^B) and forms 
the triode section of a triode-pentode multiple valve. 

The video waveform at the anode of the VA is coupled to the anode/grid of V^B) 
through C 3 , which removes the d.c. component of the waveform and converts it into a 
fully a.c. wave having a mean value of zero. V^B) then rectifies this a.c. wave, so 
producing a truly negative d.c. level just as did the grid and cathode of the sync 
separator in the illustration on page 3.4. 

The resultant negative half-cycles are first smoothed by C 5 and the resistive attenu- 
ator formed by R 5 and R 6 , and then additionally smoothed by C 6 and R 8 . A negative 
AGC potential is then applied to the first vision i.f. amplifier via R 7 ; C 4 is an r.f. 
decoupling capacitor for the grid circuit of this valve. 



§23] 



3.101 



Mean-level AGC— A Typical Circuit (continued) 

The potential developed across C e is also applied, via R 3 , Q and C 2 , to the tuner, 
where it is used as a gain-controlling voltage in the cascode amplifier and mixer 
stages. R 3 , Q and C 2 act as additional smoothing and r.f. decoupling components, 
with Cj (a low-inductance capacitor) offering better r.f. filtration than C 2 . 

The circuit shown is of the double-delay type, the first delay being formed by 
varying the reverse bias applied to the rectifying "diode" V^B). This is done by 
varying the positive potential applied to the cathode of the "diode", the control which 
does this (RVO also acting as a contrast control. As the slider of this control is moved 
further and further up towards the HT(+) line, so fewer and fewer of the positive 
half-cycles of the video waveform are rectified, and so the mean value of the wave 
applied to R 5 , R 6 and C 5 becomes more positive (=te negative). The valves con- 
trolled by this waveform in the vision i.f. amplifier and the tuner are thus allowed to 
pass more current, and a greater amplitude of video waveform is applied to the picture 
tube. 

The second delay is formed by the semiconductor diode D x and its anode load R 1( 
with Rx being switched into the circuit (by the retailer, not the viewer) if signal 
strength in the reception area where the receiver is to be used is below average. 



HT (+) 




q b Weak 

Strong 
Signals 



Negative AGC to Cascode Amplifier 
and Mixer Stage of the Tuner 



Negative AGC to 



First Vision IF Amplifier Valve 



4 ?***/ MEAN-LEVEL AGC Circuit 



3.102 [§23 

Mean-level AGC — Blocking 

A serious drawback of mean-level AGC is its susceptibility to a phenomenon called 
blocking. Blocking occurs when the AGC voltage is unable to respond quickly enough 
to counteract the sudden appearance of a very strong signal (such as might occur when 
the viewer switches channels). It is the suddenness of the signal's appearance which 
causes the trouble, for if an equivalent change in signal strength were to be applied 
over a longer period of time, the AGC would have no difficulty in following the 
change and reducing the gain of the receiver appropriately. 

Blocking occurs in the following way. Imagine that a very strong signal is suddenly 
applied to a receiver which has been operating at high gain from a much smaller 
signal. Because of the long integration time of the AGC circuit (typically, 200 ms), 
it will be unable immediately to deal with the new signal, and a large-amplitude signal 
will be applied to the grid of the final vision i.f. amplifier. This valve promptly over- 
loads, and its anode current rises to near-saturation. Anode potential "bottoms", 
with the result that a signal of near-constant amplitude is supplied to the vision 
detector. The detector, in turn, produces an abnormally large positive-going signal 
of near-constant amplitude, and applies this to the video amplifier — which promptly 
overloads also and stays in this condition for the duration of its input signal. 

The output from the VA during its overloaded condition is, of course, at near- 
bottoming potential and contains little video information. This means that a 
meaningless picture appears on the picture tube and, even more important, that a 
near-d.c. waveform is applied to the anode of the "diode" Vi(B) in the illustration on 
the last page. This means that there is no signal to be rectified (for a coupling 
capacitor will not pass d.c), and therefore that no negative AGC voltage will be 
created to counteract the large signal which is causing the trouble. Nothing can 
alter this state of affairs until the receiver is switched OFF. 

In practice, blocking can be prevented without much trouble, and with the use of 
very few extra components. A popular method is to connect a parallel combination 
of resistance and capacitance in the grid circuit of the final i.f. amplifier. This is 
shown in the illustration opposite, which is taken from the full circuit diagram of the 
Dual-Standard I.F. Amplifier A appearing on page 2.99, the same component 
numbering being used in both diagrams. 

As long as the gain of the receiver is under the control of the vision AGC circuitry, 
the anti-blocking components Ci 8 , Ci 9 and R 9 have little effect. At i.f. frequencies 
Cis is appreciably less inductive than is the much larger-capacity C 19 , and therefore 
presents less impedance to the i.f. signal. To this signal it therefore acts as an r.f. 
decoupling component, effectively short-circuiting R 8 and thereby returning the lower 
end of L 8 and L 9 to chassis. 

When a very large signal is suddenly received, an abnormally large signal appears 
also at the grid of the final i.f. amplifier. The peaks of this signal cause grid current 
to flow, which in turn causes a negative potential to be built up across C 19 and R 9 . 
This biases the valve further back into the negative V g region — with the result that less 
anode current flows and the valve is prevented from overloading. 



§23] 



3,103 



Mean-level AGC — Blocking (continued) 

With overloading prevented in this way, a less-distorted waveform will be applied to 
the detector, and a nearly-normal video waveform will reach the video amplifier. 
This near-normal waveform is passed on to the grid of the sync separator; a negative 
AGC potential is allowed to build up; the gain of the receiver is reduced, and fully 
normal operation is restored. 

The action of C 19 and R 9 is, of course, simply that of leaky-grid biasing— the time 
constant of C 19 -R 9 being made quite long (100 ms in the illustration) so as to give the 
AGC circuit ample time to get itself into a condition in which it can handle the larger 
incoming signal. 



HT(+) 




I.F. Signal 
from V1 



Anti- Blocking 
Components >v /' 

I = 
\ 
\ 



Final I.F. Amplifier 



ANTI-BLOCKING 



Seeking a Better AGC System 

Despite its simplicity and relative cheapness, it has to be accepted that mean-level 
AGC is very much a "second-best" method of controlling the tonal contrast of the 
reproduced picture. At its best, the effect it has on the picture is rather like that which 
results when the video signal is a.c.-coupled to the picture tube— i.e., when no true 
black-level reference is present at all. 

The great weakness of mean-level AGC is that it depends entirely on the picture 
signal content of the video waveform. It is not possible to prevent the AGC potential, 
and hence the gain of the receiver, from varying every time the average brightness level 
of the scene changes, so giving rise to a picture having an almost permanently un- 
natural contrast. Many ingenious circuits have been designed over the years to over- 
come this particular failing of mean-level AGC, most of them making use of the line 
sync pulse to indicate the true mean level of the received signal. 



3.104. [§23 

The Gated AGC System 

Most vision AGC systems which rely on using the mean amplitude of the line sync 
pulses for successful operation are grouped under the heading of gated AGC systems. 
The principle on which they work will be easily understood from the illustration below, 
which shows three lines (not necessarily consecutive) of a 405-line video signal in- 
spected at three different moments of time—?!, t 2 and t 3 . Each line has a different 
level of mean amplitude, and a widely different picture-signal content. 



White (100%) 



Mean Level of 
Composite Waveform 

Black (30%) 

Mean Level of Sync Pulses — *• 

— 



W\rv/Vs iv^WVM ina/VV^ 



v*~ 



-U 



ti 



White (100%) 

Mean Level of 
Composite Waveform^ 

Black (30%) 
Mean Level of Sync Pulses 



White (100%) 

Mean Level of 
Composite Waveform 

Black (30%) 
Mean Level of Sync Pulses 



-H 



* •- - w ary. 



| ^^ ! ^--ff^*' N ^\ -J^^W "^**!- -J"^^ 




Measuring the MCAH SYNC Wise Level 

A fact immediately obvious from the waveforms is that, although the mean level of 
the complete video waveform is indeterminate (because of the variations in picture 
signal content), the mean amplitude of the sync pulses is in every case clearly defined, 
and is completely independent of the picture signal. 

Look at the three lines representing time interval t 2 . The overall amplitude of the 
video signal has increased since t u as is shown by the mean level of the sync pulses. 
Yet the low picture-signal content has actually made the mean level of the complete 
waveform less than that of the three-line waveform at r a . Clearly, any AGC system 
working on the mean-level of the picture signal would derive a larger AGC control 
voltage from the waveform at t t than it would from the waveform at t 2 . But this 
would be the very opposite of what is wanted, for the overall (peak-to-peak) amplitude 
of the t x waveform is already lower than is that of the waveform at t 2 , and so needs 
boosting rather than lowering. 

This example shows that the amplitude of the vision signal carrier is much better 
determined from the amplitude of the line sync pulses than it is from the mean level of 
the picture signal itself; and you will now be looking at AGC systems working on this 
principle. (Note that the line sync pulses are used in preference to the field sync 
pulses because the much greater frequency with which the line pulses occur enable the 
amplitude-measuring circuits to respond more quickly to variations in amplitude.) 



§23] 



3.105 



The Gated AGC System (continued) 

In the 405-line system, video modulation of the vision carrier is positive-going 
(carrier amplitude increasing with picture brightness), and the line and field sync 
pulses start from a black level equivalent to about 30% modulation and extend down- 
wards from this level until they reduce carrier amplitude to near-zero. 

To measure the amplitude of the line sync pulses, it is necessary for the measurement 
to be made during one or other of the blanking periods which immediately precede 
and immediately follow every individual sync pulse — that is to say, during either the 
front or the back porches. Almost all sync pulse measuring circuits use the back 
porch period because it is considerably the longer of the two — having, you will recall, a 
typical duration of some 7-5 fxs against the 1-75 ;j.s of the front porch. 

It is during the duration of the back porch, therefore, that the measuring circuit 
required must be gated, or made active. 

The illustration below shows the essential features of the first part of a typical gated 
AGC circuit. (This one, as it happens, was designed by the Mullard Research 
Laboratories.) 



Video Signal to 
Picture Tube 



HT(+) 



WAVEFORMS 




Sync Pulse Waveform at V1 (A) Grid 



FIRST PART of a 

6HTED-A6G 

CIRCUIT 



(+)- 





Effective Waveform at V1 (A) Grid 



3.106 [§23 

The Gated AGC System (continued) 

Valve Vi(A) in the circuit on the last page is connected to operate as a cathode 
follower. It forms one half of a double triode (whose other half you will meet quite 
soon); and it has two separate input signals applied to it — one from the anode of the 
video amplifier and the other from that of the sync separator. 

The signal from the VA is obviously the same as that which is delivered to the 
picture tube and to the grid of the sync separator. It is coupled to the grid of \\(A) 
by the potential divider formed by R x and R 2 — these resistors reducing the amplitude 
of the signal (it is about 70 V at this point) to a level within the grid base of the valve, 
and also providing circuit isolation. 

The input signal from the anode of the sync separator consists of negative-going 
pulses of large amplitude, coupled to V^A) through C 2 and R 3 , and occurring at 
exactly the same moments as the positive-going sync pulses in the video waveform. 
But the amplitude of the negative sync pulses is so much greater than the maximum 
amplitude attained by the positive-going sync pulses of the video waveform that these 
latter are effectively cancelled. The resultant waveform at Vi(A) grid is similar to that 
shown. The important thing about this waveform is that its most positive excursion 
corresponds to the black level of the video waveform — whence the name of black-level 
detector commonly applied to all that part of the circuit lying to the right of the vertical 
dotted line in the illustration. 

The waveform appearing across R 4 , the cathode load resistor of the black-level 
detector, would under the normal operating conditions of a cathode-follower be almost 
identical in size, shape and polarity to the signal present at the grid. But normal 
operating conditions are here modified by the presence of a 0-01 jxF capacitor, C 3 , 
connected in parallel with R 4 . This capacitor, in conjunction with R 4 and the output 
impedance of the valve, forms an integrating circuit. As long as signal amplitude at 
the grid remains steady, an almost steady level of d.c. is built up across C 3 . The 
amplitude of this d.c. level is proportional to the maximum level reached by the signal 
at the grid — which, as you know, occurs during the periods of the front and back 
porches. 

But the voltage across C 3 , which is required eventually to form the steady AGC 
potential it is desired to obtain, soon begins to leak away through R 4 between the 
arrival of successive sync pulses to charge it up again, and would quickly become the 
reverse of steady if nothing were done about it. This is where the capacitor Q 
(shown in dotted outline in the illustration on the last page) comes in. In conjunction 
with the parallel resistance of Rj and R 2 , Q forms a differentiating circuit at the grid 
of V X (A) which produces a positive overshoot at the end of every sync pulse (i.e., 
during the period of the back porch). C 3 — an integrating capacitor — is made to 
charge up to the peak value of this overshoot, whose value is carefully chosen to make 
up for the charge leaking away during the subsequent picture signal period. Thus the 
voltage across C 3 is not allowed to fall below that represented by the black level of the 
input waveform, and an accurate AGC voltage level is maintained. 

Whenever the strength of the received signal increases, a larger positive-going signal 
is applied to the grid of the VA, and a larger negative-going waveform appears at its 
anode and at the grid of V X (A). But the reference level of this waveform (represented, 
of course, by the tips of the sync pulses) remains constant however great its amplitude 
becomes ; so that the only effect of a stronger received signal is to cause the average 
value of the waveform at V X (A) grid to move in a more negative direction. 

The result is a reduction in the voltage across C 3 ; and the average value of the resul- 
ting AGC voltage is lowered just when such a reduction is needed. 



§M] 



3.107 



The Gated AGC System (continued) 

The signal developed across C 3 by the black-level detector is not yet suitable for 
feeding direct into the AGC line. It still requires amplification, in order to give the 
AGC circuitry rapid response and good gain-controlling efficiency; and it still requires 
a measure of d.c. restoration to ensure that its polarity is always negative with respect 
to chassis. At the moment, although moving in the desired sense (becoming less 
positive as vision signal amplitude increases), it is at all times positive (above zero 
volts) and so unsuitable for its destined AGC role. 

Before seeing how this is done, however, you should first break off and consider 
the nature of the 625-line video signal, whose characteristics will inevitably affect the 
nature of any corrective circuitry through which the two signals may need to be put in 
a Dual-Standard Receiver. 

The illustration below compares the way in which the output signal from the VA 
varies with changes in the amplitude of the received signal, in the two line systems. 
The point to note is the way in which the sync level of the 625-line signal actually 
becomes more positive when signal amplitude rises. In other words, this signal, in 
addition to being at all times positive with respect to chassis (which is not desired) 
does not even move consistently in the right sense. 




■ HT (+) - 
Sync Level 

Black 



White 



Average Value of 
Sync Pulses 



0V 




100% Increase in Amplitude 



405UN6 



HT (+ 




Sync Level 
,=s Black 

White 



Average Value of 
Sync Pulses 




100% Increase in Amplitude 



62S-UM 



3.108 [§23 

The Gated AGC System (continued) 

You know that the reason why the 625-line signal expands in both senses as signal 
amplitude increases is that the tips of the sync pulses represent 100% modulation of 
the vision carrier, and that peak white is only reached at 12% modulation. The 
system contains no zero-signal reference level at all. 

The trouble is that, as the amplitude of the sync pulse tips rises with increasing 
signal amplitude, so must the average value of the sync pulses viewed as a whole. 
With the mean level of the sync pulses thus increasing with signal amplitude, there will 
be a corresponding increase in the black-level signal present at the grid of Vj(A), and a 
most undesirable increase in the integrated signal appearing across C 3 . An AGC 
voltage of such a nature is worse than useless, and some form of switching is required 
if the same amplifier is to be used to handle the AGC signal on both line systems. 

A Full Gated-AGC Circuit 

A full gated-AGC circuit suitable for use in a Dual-Standard TV receiver is shown 
opposite. The amplifier (V^B)) forms the second half of a double-triode valve, of 
which the first. half forms the black-level detector. 

The input circuit of the amplifier is switched by two sections of the Standard 
Selection switch in such a way that, when the receiver is set for 405-line operation, the 
negative-going signal from the black-level detector is applied to the cathode of the 
amplifier — the grid being returned to chassis through the preset contrast control RV 2 . 
The input signal is thus amplified without being inverted, and an amplified AGC 
voltage of the desired polarity is obtained. 

With the receiver set for 625-line operation, the positive-going signal from the 
black-level detector is applied to the grid of the amplifier, the cathode being returned 
to earth through the 625-line preset contrast control RV^ With the amplifier con- 
nected in this way, its input signal is not only amplified but also inverted; and the 
desired negative-going signal again appears at the anode. 

The effect of the preset contrast controls is to vary the operating bias of the amplifier, 
and so the threshold level which has to be exceeded by the input signal before AGC 
action can start. In the circuit shown, RV X and RV 2 are capable of being individually 
adjusted, and a single Contrast control on the outside of the receiver can be operated 
by the viewer to adjust to his liking the contrast of the displayed picture whichever the 
line standard on which his set is receiving. 

The next step is to see how the AGC amplifier introduces a new d.c. level into the 
wholly-positive signals produced by the black-level detector V^A), so that they 
become wholly negative — and so usable for AGC purposes. 

Observe an odd feature about V^B). It lacks the usual anode load resistor, and its 
anode is not returned to a positive d.c. voltage at all. It is connected instead to the 
separate AGC winding on the line output transformer which you read about in 
Section 20. Connected in this way, the valve is only capable of passing anode 
current at all when a positive-going pulse (of some 300 V amplitude) is applied to its 
anode through the capacitor C 4 . Being derived from the extra winding on the line- 
output transformer, these pulses recur at the repetition frequency of the line-scan 
generator. 

Whenever such a pulse is applied to C 4 , the anode of the amplifier is momentarily 
driven positive, and the valve conducts anode current — but only if the AGC delay 
potential set by the Contrast control has been overcome by the input signal from the 
black-level detector. If this delay potential has not been overcome — i.e. if the vision 
signal itself has not exceeded a certain amplitude — V^B) anode does not conduct, 
and no AGC voltage is developed at all. 



§23] 

A Full Gated-AGC Circuit (continued) 



3.109 




3.110 [§23 

A Full Gated-AGC Circuit (continued) 

The pulse arriving from the line output transformer charges C 4> and at the end of 
the pulse the capacitor has its upper electrode at a high positive voltage and its lower 
electrode at a much lower level — a little above the level of the cathode (or of the 
grid, whichever of them is "earthy" at the time). When the pulse ceases, the applied 
voltage falls to zero and the upper electrode of C 4 falls with it. C 4 cannot change its 
charge instantaneously, so the lower electrode also drops by the same amount (300 V). 
The anode of the AGC amplifier at once becomes heavily reverse-biased, and no anode 
current can flow. 

Q, however, begins to discharge slowly through the large-value resistor R 7 . The 
reverse bias on the anode becomes less, and when it has dropped just enough, anode 
current again begins to flow — but only momentarily until another pulse from the line 
output transformer recharges C 4 and reverse-biases the anode once again. In this 
way, the anode of V^B) is only allowed to go positive at all during the few micro- 
seconds in which C 4 is recharging through the low resistance of the valve. Virtually 
the whole of the output waveform is thus of negative polarity. 

This is, of course, nothing but the familiar action of d.c. restoration. The outcome 
is a negative potential developed at the anode, almost steady save for the brief upward 
excursions while the valve briefly conducts during the period of the anode pulses. 
These upward fluctuations are attenuated by the potential divider formed by R 7 -R 8 , 
and smoothed by the integration circuit R 8 -C 5 . A very steady negative AGC potential 
is thus developed across C 6 , fully suitable for controlling the gains of the tuner and 
vision i.f. stages. 

This AGC voltage, of course, varies in amplitude in proportion to the amount of 
anode current caused to flow, during the short duration of the anode pulses, by the 
signal applied to its grid (625-line system) or to its cathode (405-line) respectively. 
When a sharp increase in signal amplitude occurs, the AGC voltage is driven even 
more negative. It continues at that polarity for exactly as long as the signal remains 
abnormally strong. Thus the stronger the signal, the larger the negative AGC control 
voltage applied as grid bias to the earlier stages of the receiver, and the smaller 
become their respective gains — until the adverse effects of the suddenly increased value 
of the received signal have been wholly overcome. 

Advantages of the gated type of AGC circuit include freedom from the "blocking" 
paralysis to which the type of circuit working on mean signal level is prone. This 
immunity is due to the d.c. coupling between the anode of the VA and the grid of the 
black-level detector, and to the absence of long time-constants. 

Another advantage is flexibility — in that, though the positive pulses applied 
through the capacitor C 4 to the anode of the AGC amplifier are usually in synchronism 
with the sync pulses of the received signal, they do not have to be; and their repetition 
rate is in practice immaterial. 

Lastly, the system is very efficient, because the degree of amplification applied to the 
signal from the black-level detector is sufficient to allow quite large changes in the 
amplitude of the received vision signal to occur without causing more than minor 
variations in the amplitude of the resulting video signal. 

AGC delay diodes are used in gated AGC systems in the same way, and for the 
same purposes, as they are in mean-level ones. 



§23] 3.111 

REVIEW of Automatic Gain Control 

The purpose of AGC in a TV receiver is to maintain the sound and video signals 
constant at a predetermined level set by the Volume and Contrast controls on the outside 
of the chassis, whatever variations may occur in the strengths of either of the received 
r.f. carriers. 

Most TV receivers have separate sound and vision AGC controls, one operating from 
the sound i.f. carrier, the other from the vision i.f. carrier. It is usual to arrange for the 
vision AGC to control the gain of any common sound and vision amplifying stages there 
may be (as, for instance, in the tuner) and for the sound AGC to control the gain of the 
sound i.f. amplifying stages only. 

AGC for the Sound Carrier is required only in the 405-line system, because in the 
625-line system the sound carrier is frequency modulated and is deliberately limited to a 
constant amplitude before it is detected. AGC for the 405-line sound carrier is derived 
from the d.c. level produced by the sound detector, and is applied to control the gain of the 
sound i.f. amplifying stages. 

Mean-level AGC is the simplest, cheapest and most widely used AGC system for the 
vision carrier, but not the best. In it, the amplitude of the vision carrier is estimated by 
measuring the mean amplitude of its video modulation. This is done by taking the 
fluctuating d.c. level produced by leaky-grid action at the grid of the sync separator valve, 
and smoothing it by means of a simple C-R circuit. 

The reason why this method is inaccurate is because a strong signal having little 
picture-signal content (and therefore representing a dark scene) will produce a weaker 
AGC voltage than will a much less strong signal having a large picture-signal content; 
and vice versa. The AGC voltage is therefore dependent on the tonal composition of the 
transmitted scene rather than on the actual strength of the received signal. 

Gated AGC systems determine the amplitude of the vision signal carrier from the 
amplitude of the line sync pulses, rather than from the mean level of the picture signal 
itself. Though more complicated and expensive, this method of determining the level 
of signal at which AGC needs to be applied is much more accurate, and produces a 
steadier control of contrast in the picture appearing on the screen. 

Delayed AGC is used to prevent any control of gain being applied until the received 
signal is strong enough to overcome a preset threshold. The result is to stop any AGC 
being applied when weak signals are being received. The AGC threshold which causes 
this delay is produced by a voltage developed across a forward-biased diode. 

Double-delayed A GC employs two separate delay diodes, one operating after the other. 
The second delay allows the early stages of the receiver to operate at fairly high gain to 
ensure a high signal-to-noise ratio, but restricts the gain of these stages whenever signal 
strength becomes excessive. 



3.112 



§24: POWER SUPPLIES 



All valve-type TV receivers operated from the mains have two requirements for 
primary power supplies: (a) a high-voltage (HT(+)) supply, typically of around 200 V 
d.c, for the anodes of the various valves in the circuitry; and (b) a low- voltage (LT) 
supply, typically of 6*3 V at 0-3 A, a.c, for the heaters of the valves and of the picture 
tube. 

The main secondary voltages required are the EHT and boost voltages needed for the 
electrodes of the picture tube. You have already seen in Section 20 how these two 
voltages are generated in the line scan output stage, which is itself powered in the first 
place from the HT( + ) and LT supplies. This type of derived voltage will therefore 
not be described again, though they are shown schematically in the block diagram 
below. 



A.C. Mains 
Input 

O— — 



RECTIFICATION AND 

AMPLITUDE 

CONVERSION 



The 

POWER SUPPLIES 

for a TV Receiver 



LINE OUTPUT 
STAGE 



200 V d.c. 

HT (+) for Valves 

6-3V(S>0-3Aa.c. 
for Valve and 
Picture Tube Heaters 



E.H.T. 



Picture 
\ Tube 
Boost Electrodes 

Voltage J 



Other operating voltages, such as those required for screen grid electrodes and for 
grid bias and reference potentials, are usually derived from potential divider networks 
connected across the HT(+) supply, or from valve-operating currents flowing through 
small-value resistances. 

The power consumed by a typical 19" TV receiver from a 240 V mains supply is 
of the order of 180 W — which is not much more than the power consumed by the 
electric lamp which you would normally be using to light the sitting room. 



The Mains Supply 

Although thousands of receivers are still in use which were designed to work on the 
earlier mains voltages ranging from 200 V to 240 V, the domestic a.c. main supply in 
Britain is now standardised at 240 V, 50 Hz. It is conveyed on a three-wire system. 
Two of these wires, designated Live and Neutral, carry the 240 V between them. The 
third is a protective Earth wire which is usually connected to ground where the supply 
enters the house. The Neutral wire, though not directly earthed, is normally at near- 
earth potential, whereas the Live wire carries the full 240 V. 

Until a few years ago, the standard colour coding for British mains wiring was Red 
for the Live wire, Black (or Blue) for the Neutral wire, and Green for the Earth wire. 
But recent international agreement has resulted in a new standard colour code which is 
Brown for the Live wire, Blue for the Neutral wire and Green and Yellow for the Earth 
wire. All modern TV receivers are being wired in accordance with this convention, 
although only the Live and Neutral wires will probably be used (you will see why this 
is so shortly). 



§24] 3.113 

The HT Supply 

The 200 V required for the HT(+) supply, being quite close in value to the value of 
the a.c. mains supply itself, is usually obtained through a simple half-wave rectifier 
circuit connected directly to the mains, very similar to the circuit described in Basic 
Electronics, page 1.18. The rectifier may be either of the metal type or it may be a 
diode valve. In the more modern TV receivers, it will more likely be a semiconductor 
diode. 

The big advantage of this method of producing the HT voltage is circuit simplicity 
and low cost. The big disadvantage is that the Neutral lead from the mains has to be 
connected directly to the receiver chassis — which, in turn, means that the chassis must 
be isolated from the viewer. The reason for this is a bit involved. Since the chassis 
cannot be directly connected to earth, there is no need to equip it with a third (Earth) 
wire, and it is therefore usual to connect it to the mains by means of a simple (and 
cheap) two-pin connector. Unfortunately, most connectors of this type are capable 
of being plugged into the wall socket either way round — which means that it is possible 
for the chassis of the receiver itself to become "live". 

This is of no particular significance as far as the receiver itself is concerned, but it 
could be a potentially serious hazard for the viewer if he should accidentally touch 
a metallic part of the chassis. It is as a partial safeguard against this risk that control 
knobs are usually of the push-on type, instead of being secured by metal screws, 
and that the loudspeaker grille is frequently made of non-conducting fabric or plastic. 
Other precautions are taken to ensure that all accessible metal parts are well insulated 
from the chassis. 

The use of a 3-pin mains connector, even though its Earth pin remained unused, 
would prevent the accidental reversal of the mains supply to the receiver, and is for this 
reason to be recommended. But even this method of connection relies on the wiring 
within the mains plug being correct — something which the viewer should never take 
for granted. 



Rectifier 




200V 

HT(+) 



Chassis 



Half -Wave Rectification of the HT Supply 

The illustration shows a typical half-wave rectifier circuit, with LC smoothing, such 
as might be found in a 19" receiver. The three resistors connected in series with the 
diode enable the receiver to operate from mains supply voltages over the full range 
200-240 V. When a 240 V supply is applied, all the resistors are used and the output 
voltage from the diode is reduced to the desired 200 V. When a 200 V supply is 
applied, none of the resistors is used. Intermediate supply voltages call for the use of 
intermediate values of resistance. 

These resistors are called dropping resistors and are usually constructed as a single 
high-wattage resistor tapped at intervals along the length of the wire from which it is 
made to provide the individual resistance values required. The same resistance unit 
is used to provide similar, but separate, dropping resistors for the valve heater supplies, 
as you will see in a moment. 



3.114 [§24 

The LT Supply 

Most of the valves used in any TV receiver require operating voltages of some 6 or 
7 volts for their heaters, but several require somewhat higher voltages than this. A 
theoretically practical way of providing such voltages would be to use a small heater 
transformer producing the desired heater voltage and operated directly from the 
mains supply. Such an arrangement, however, presupposes that every valve used in 
the receiver operates at the same heater voltage, and that a.c. mains supply is always 
available. Since this is not always the case, something more flexible is required. 

A simple and cheap method of producing the required heater power is to connect 
all the heaters in series and then to connect the whole chain across the mains supply 
through any additional resistance which may be required. This method is used in all 
British receivers. Clearly, since all the heaters are in series, they must all be supplied 
with the.same operating current, which means that valves operated in this way must be 
designed to operate at a standardised heater current. This standardised heater current 
is 300 mA (0-3 A). It remains constant even for valves whose operating voltage 
requirements are different. 

Consider, for example, a receiver containing 17 valves, ten of which require heater 
voltages of 6-3 V, four of 7-6 V and three of 27 V. All operate from a heater current 
of 0-3 A. The total voltage which needs to be connected across such a series chain is: 
(10 x 6-3 V) + (4 x 7-6 V) + (3 x 27 V) = 174-4 V. With the chain connected across a 
mains supply of 240 V, an additional resistance must be added of a value such that, 
when the chain is passing 0-3 A current, it will "drop" the excess voltage. This excess 
voltage is obviously (240 V— 174-4 V=) 65-6 V; and the value of the resistance needed 
to drop such a voltage with this current is determined by Ohm's law. R = 65-6 V-f- 
0-3 A, which works out at about 219 ohms. 

The illustration below shows how the heaters in the example quoted are connected 
to the mains supply. Verify, by Ohm's law, that the resistor values shown as being 
encountered by the electrons entering the circuit from the mains are those required to 
drop the excess voltage if the mains voltage is 240 V, 220 V, 210 V or 200 V respectively. 




I ' i \ Valves 
I ! I i 

(6-3V 1 , J6-3V' 27V 7-6V 6-3V 7-6V 6-3V 6-3V 27V 7-6V 

VI V2 V3 V4 V5 V6 V7 V8 V9 V10 

27V 6-3V 6-3V 7-6V 6-3V 6-3V 6-3V 

^rLiiiirLTLrLru 

V17 V16 V15 VU V13 V12 Vll 



Live 



Mains Supply 



Chassis 



p/ALVEiHEATERSl 

Iff 
MifttMM 



Neutral 



§M] 



3.115 



The LT Supply (continued) 

The heater element of a valve is very like the filament of an ordinary electric lamp. 
Its resistance when cold is low; but by the time it has reached normal operating 
temperature (which is red hot), this resistance has become comparatively high. So 
if a series-connected chain of cold heater elements were suddenly switched across the 
mains supply, the low initial resistance of the chain would allow an enormous surge of 
current to flow — more than enough to "blow" the fuse in the receiver. This is pre- 
vented in practice by connecting in series with the chain a temperature-dependent 
resistor called a thermistor (a name derived from the words "thermal resistor"), such 
as you have already met in Section 20. 

A typical thermistor may have a resistance when cold of nearly 4,000 ohms; but 
this resistance, when the thermistor is hot and passing 0-3 A, may drop to as low as 
50 ohms. With such a resistance connected in series with the heater chain, the 
"switch-on" heater current is limited to a safe value of under 60 mA, given a mains 
voltage of 240 V. As the thermistor warms up, its resistance becomes progressively 
less, and more heater current is allowed to flow; the process continues until both 
thermistor and valve heaters reach an equilibrium temperature at which the desired 
heater current of 0-3 A flows. Allowing the heaters to warm up gradually in this way 
does much to prolong their useful life. 

The physical appearance of a typical thermistor is shown below. It is some 20 mm 
long, with a diameter of about 10 mm. Note, however, that there exist a great variety 
of thermistors of all shapes and sizes, their dimensions depending on their intended use 
and the design preference of their manufacturers. 



Theoretical Symbol Low 

Resistance 



n» rmmmm 




High Resistance 



The HT and LT Supplies— Circuit Diagram 

The complete circuit of an HT/LT power supply for a typical valve-type TV receiver 
(it is in fact Baird Model M.620, and its circuit diagram is given by courtesy of Radio 
Rentals Ltd) is shown in the illustration overleaf. The mains voltage adjustment 
resistors are, as you know, manufactured as a single unit, and enable the receiver to 
operate over the voltage range 200 V to 250 V. HT adjustment is made (for ease of 
identification) through the two Red leads, and LT adjustment through the two Brown 
leads. 

The several capacitors shown are there for decoupling purposes. They also help 
to prevent interference from the many different waveforms and frequencies present in 
the receiver from being conveyed to other parts of the circuit by way of the heater 
chain. 

The UHF tuner, of course, presents a special case because of the very high frequen- 
cies involved; and small r.f. chokes are included in its heater circuit with the object 
of suppressing any interference which may arise at this point. 

The danger of creating interference is also a reason for positioning the heater of the 
line oscillator stage (see circuit diagram) at the remote, or earthy, end of the chain, 
whence it is harder for the line oscillator waveform to reach the more sensitive stages 
of the receiver. 



3.116 



[§24 



The LT and HT Supplies — Circuit Diagram {continued) 

Another reason for situating certain heaters at the earthy end of the chain is to 
keep the heater-to-cathode potentials of certain valves as low as possible, so minimising 
the possibility of heater-to-cathode breakdown. 

This is the reason why the picture-tube heater itself is placed almost at the end of 
the chain. You will recall that the cathode of this tube is connected to the anode of 
the video amplifier, and is therefore normally held at a high positive potential. If the 
heater of the picture tube were to be connected in the high-voltage end of the chain, a 
very high potential difference between the heater and cathode would be created every 
time the mains voltage passed through its negative half-cycle. 

Such a potential could easily cause the eventual breakdown of the heater-to-cathode 
insulation, resulting in a highly undesirable short-circuit being created between these 
two electrodes of a very expensive component of the TV receiver. 

Red, ^Red 



200V 




HT(+) 



4r X 4 

820p 820p OOlja OOljt 002>i 002>i -002^ 0047^ 0047ji 0047^ 



ti» HT»hi LT POWER SUPPLIES 



§25: FAULT-FINDING IN THE TV RECEIVER 



3.117 



Efficient fault-finding in a TV receiver is a matter of applied logic, just as it is in the 
case of a motor-car or any other complex piece of equipment which relies for its 
successful operation on the correct functioning of several separate stages. It is true 
that an experienced TV serviceman or garage mechanic, long familiar with the make of 
receiver or car brought in to him for attention, can often put his finger on the cause of 
the trouble without going through any recognisably logical mental process at all. It 
is simply that he has seen or heard that particular symptom or collection of symptoms 
before, and so can make an experienced guess as to which particular component in 
which particular stage of the mechanism is causing the trouble — and he can often 
do this without having much in the way of theoretical understanding of how the 
mechanism works. 

But the key word in that sentence is "experienced", and the experienced TV service- 
man is not likely to learn much from reading this particular Section of Basic Television 
anyway. For the trainee repairman, however, and for the "do-it-yourself" enthusiast 
wishing to learn how to maintain his own set, there is no substitute for a more dis- 
ciplined approach. This approach must begin with an exact definition of the symptoms 
displayed by the faulty equipment, followed by logical analysis based on the TV 
theory covered in this series, until the particular stage in the equipment in which the 
fault is located has been identified by a process amounting to a rational elimination of 
the impossible. 

The first rule in fault-finding, therefore, is to define the symptoms. Even the home- 
repairman will find the rule helpful in getting himself started off straight in the process 
of rational deduction which must follow; but for the serviceman confronted with a 
strange set which he is simply told "won't work", it is an essential beginning. Start 
by asking yourself or your customer whether it's the sound or the vision that's giving 
trouble, and say you get the answer, "There's no sound". "Ah! But the picture's all 
right, on all channels?" you ask. "Oh yes, that's fine." "Good. Is the sound missing on 

all channels?" "No, only on the 405-line ones " Already you've got a long way, 

but ram it home with, "I see. So there's nothing wrong on the 625-line channels, but 
the sound is missing from all the 405-line channels even though the picture on these 
channels is OK?" "Yes, that's right." 

What you have done is to define the fault so that you know exactly what your cust- 
omer is complaining about. The fact that (as you may well find out later on) the 
picture is wishy-washy and the tube ought to have been replaced years ago has nothing 
to do with the fault you are looking for. Your customer is quite happy with the 
picture he is getting, bad as it is. All he wants is the sound to go with it. Anything 
else you can do to improve the performance of his set will be merely a bonus — hope- 
fully to both of you. 

The next rule is simple. Switch on and verify that what the customer thinks has 
gone wrong is in fact what has gone wrong. In the process you may well spot other 
symptoms which will help you in the coming diagnosis. 

Now start thinking, and as a first step eliminate the impossible. In the case in 
question, there can be nothing wrong with the loudspeaker, or with the amplifier 
feeding it; for otherwise it would not have worked when the receiver was set for 625 
lines. And since the fault persists on all the 405 channels but does not affect the 
corresponding pictures, it is unlikely to lie in the tuner or in any of the common sound 
and vision i.f. stages. At once you know you are looking for a stage which is common 
to all 405-line channels, but which affects the sound signal only and which forms no 
part of the audio-frequency amplifying stages. . . . 



3.118 [§25 

Logical Analysis of the Symptoms 

Well, it shouldn't take you long now to deduce that the trouble must lie somewhere 
between the sound i.f. stage and the sound detector, and that is where you start looking. 
The fault itself may be in the detector diode or in one of the components associated 
with it, or it may be a faulty section of the Standard Selection switch. These are 
things which you will have to determine by a process of trial-and-error. But you will 
have enormously cut down this process by careful identification and verification of the 
exact nature of the symptoms displayed, and by a couple of minutes' rigorous analysis 
of what could possibly have caused them. 

Logical analysis of the symptoms displayed is thus the basis of all successful fault 
location, just as it is of any good medical diagnosis. The next few pages will accor- 
dingly be devoted to a general approach to the problem on these lines, followed by 
a brief look at a dozen of the more common symptoms and the best way of rectifying 
them. 

Fault-location Charts 

Featured on the next few pages are three fault-location charts, to which the block 
diagram below acts as a key. Answer the questions posed on this key, and you will 
be directed to the particular chart you need. Note that, in the key and in the charts 
alike, whenever the answer to a particular question is Yes, you are directed down the 
page to the next question; whenever it is No, you are directed across the page either to 
the left or to the right. 



THE FAULT 



I 



ARE BOTH 

PICTURE AND SOUND 

FAULTY? 



-►(no) 



IS SOUND 

ONLY 
FAULTY? 



-*(no) 



IS VISION 

ONLY 
FAULTY? 



SEE 
CHART No. 3 



SEE 
CHART No. I 




All the charts have obviously had to be simplified, in that they cannot hope to cover 
every one of the wide variety of faults and symptoms likely to be encountered in 
practice. But they provide a very useful introduction to correct fault-finding pro- 
cedure, and in most cases will guide you pretty directly to the "suspect" stage of the 
receiver at which a more detailed examination should begin. 



§25] 

Faulty Sound 



3.119 



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3.120 [§25 

Faulty Sound 

Fault Location Chart No. 1 is based on the fact that in the Dual-Standard Receiver 
faulty sound can occur either on one standard or on the other, or on both of them 
together. Alternatively it may occur on one only of the available 405 channels. 

Take the case where sound is missing on both standards, and assume that a faint 
sound (it is probably a 50 Hz mains hum) can be heard when the ear is placed close to 
the speaker. Assume, further, that the volume of this faint sound is unaltered when 
the Volume Control is operated. 

The chart tells you that the fault probably lies within the circuitry associated with 
the sound output stage, and that the sound output valve needs checking. This can 
be done either by connecting it to a valve-tester, if you have got one, or by temporary 
replacement by another suitable valve known to be operational. 

You are also advised to check the operating potentials of this sound output valve, 
and the value of the HT supply to it. While you are doing this, it is good practice to 
look around for signs of overheated or burnt-out resistors. Overheating can occur if 
a grid coupling capacitor goes short-circuit or develops a low internal resistance, 
thereby causing the valve to pass an enormous overload current. (This is always 
potentially liable to happen, you will recall, because of the positive voltage applied 
to the grid of the valve from the anode of the preceding stage.) 

Lastly, you are advised to check the operation of the Volume Control itself, lest the 
valuable evidence provided by the fact that it doesn't cure or alleviate the fault when 
it is operated should itself prove unreliable. If operation of the Volume Control 
does mitigate the symptoms, it is obvious that the fault must lie somewhere in the 
circuitry earlier than the point at which the Volume Control exerts its effect. 

Faulty Vision 

The number of things which can go wrong with what appears on the screen of a TV 
receiver is formidably large — whence the size and apparent complexity of Fault 
Location Chart No. 3 {see pages 3.122-3.123). It is devoted entirely to locating 
faults affecting the visual presentation of the received signal — whether the symptoms 
be loss of picture signal without impairment of the raster, or distortion of the picture 
signal or of the raster or of both, or even a total absence of anything appearing on the 
screen at all. 

Consider, as an instance, the case where a dark horizontal band appears across the 
screen, but everything else about the picture is all right. On the chart you answer Yes 
seven times to the questions concerning size, linearity, brilliance, contrast and syn- 
chronisation; but the question "Is the picture free from interference?" forces the 
answer No. You are now directed to more questions to the left, where you eventually 
answer Yes to "Is there a dark horizontal band across the picture ?" You are then told 
to check the presence of mains hum in the video circuit, because a dark bar of this 
type is almost always caused by a 50 Hz mains signal getting into the video feed to the 
picture tube. It usually does so by way of heater-to-cathode leakage (or even break- 
down) in one of the valves associated with the vision signal, or by faulty decoupling in 
these stages. 

If the hum bar, as it is called, is accompanied by a 50 Hz sound from the speaker, the 
fault probably lies in one of the valves common to both sound and vision signals. If 
there is no such sound accompaniment, the fault must lie in a stage which handles the 
vision or video signals only. 

Another possible cause of the hum bar type of fault is faulty decoupling of the 
vision AGC line, for any 50 Hz ripple not removed from this line will be fed back to 
every valve in the vision stages which is controlled by the AGC potential. 





SCREEN BLANK: 
NO SOUND 






1 






ARE VALVE 

HEATERS 

GLOWING? 


— »Aio) — 




/ 


YPS1 














Check 

Mains supply. 
Mains plug fuse. 
Receiver fuse. 
Continuity of 
picture tube and other 




Check 

HT voltage. 
If absent, check HT 
fuse, rectifier diode, 






dropping resistor 
connections and 
continuity. 
Check for short- 




valve heaters. 

Mains dropping 
resistor and its 
connections. 




circuits on HT line. 






Thermistor. 



§25] 3 - 121 

Screen Blank: No Sound 

This fairly common fault can fortunately be located and rectified with little difficulty. 
As you will see from Fault Location Chart No. 2, it is usually traceable to either the 
a.c. or the d.c. side of the power supply circuits. 



nun 

LOCATION 

CHART 

Ho.2 



Fault-finding Hints 

When you are tracing a simple fault of this kind, never be tempted to overlook the 
obvious. Even experienced service engineers will probably not deny that they have 
on occasion spent time trying to find out why no mains supply is reaching a receiver, 
only to discover some minutes later that it is the wiring to the mains socket on their 
own workbench which is faulty. The quick plugging-in of a bench-light known to be 
operational would have revealed this fault at once. 

Another point to watch out for is the possibility that an apparently faulty valve may 
merely be loose in its holder. When looking for the cause of "no heaters glowing", 
first make sure that every valve is fitting firmly in its holder before starting out on the 
lengthy job of finding the one which has an open-circuit heater. 

For a lengthy job it can certainly be if there are a large number of valves in the set, 
and every one of them has to be taken out in turn and checked individually for heater 
continuity. 

A simpler, though potentially more risky, way of doing this is to connect a pair of 
short insulated wires to a 68-ohm wire-wound resistor rated at about 6 watts, and to 
connect the other ends of the wires to a pair of prods which are fully insulated apart 
from their tips. Then touch these prods for a short period across the heater connec- 
tions of every valve in turn. 

As soon as the valve with a faulty heater is reached, every other valve heater in the 
chain will light up — for the reason that continuity of the whole chain will have been 
restored by the 68-ohm resistor. The value of this resistor was purposely chosen as 
being representative of the "hot" resistance of a typical valve heater. 

Since this test must be made with the set switched ON, great care must be taken when 
carrying it out. If you are unsure of your ability to do it safely, substitute for the 
prods a pair of crocodile clips, and connect them across each valve in turn with the set 
switched OFF. Switch it on only when the connection is securely made, and when you 
and your hands are well clear of the danger area. 



3.122 
Faulty Vision 



[§25 



FAULTY 


i' 




Is there a 
raster? 


H^)" 




Check 

Line-output and line- 
scan oscillator voltages 
and components. 

Boost voltage and 
Width controls. 

Line-scan coils. 

Boost diode. 



Check 

Field output stage 
operating voltages 
and components. 

Ditto of field-scan 
oscillator. 

Vertical Linearity 
and Height controls. 



Is it linear? 



>(^y 



Check 

Brilliance Control 
and decoupling 
capacitor. 

Picture tube 
potentials (not EHT). 



(NoW Is its brilliance controllable? 



Any picture? 



■/noV 



Is there a dark 
horizontal band across 
picture? 



5 



Check 

Presence of mains hum in 
video and vision circuits. 
HT decoupling capacitors. 



Is there a thin band 
of vertical dots at 
screen centre? 



-©*— 



Check 

Contrast control 
and connections to 
picture tube. 

Video amplifier 
operating voltages 
and components. 

Standard Selection 
switch sections. 



Is there sound 
on vision? 



Check 

Tuning of sound 
rejector circuits. 



Is picture covered 
with random dot 
pattern? 



Check for corona 
discharge during 
line flyback. 



Check EHT supply for 
corona discharge. 



(Yes) 



Is sync OK? 



Is picture free 
of interference? 



«/noV 



WHAT ARE YOU 
COMPLAINING 
ABOUT THEN? 



§25] 



3.123 



VISION 



Is there a 
vertical line? 



— *®- 



Check 

Line-scan coils 
and width 
control. 



Is there a 
horizontal line? 



-HNo 



Check 

Field-scan coils. 
Field output stage 
and transformer. 
Field-scan oscillator. 



Is vertical linearity ^ fTN 

poor? v^/ 



Check 

Vertical Linearity 
controls and components. 

Field-scan oscillator 
valve and its 
operating voltages. 

Field output stage. 



Is horizontal linearity 
poor? 



Check 

S-correction capacitors. 
Position of linearity 
loops. 

Line-scan coils. 
Boost voltage and 
boost diode. 




-HNo 



Check 

Line-output 
valve. 

Line-output 
transformer. 

EHT rectifier 
diode. 



625 and 405 

missing? 



-WNo 



405 missing? W No 



625 missing? 



Check 

Video amplifier 
connections to 
picture tube. 

Standard Selection 
switch sections to 
video amplifier and 
from VHF and UHF 
tuners. 



Check 

405 video detector 

diode and components. 

Standard Selection 

switch sections and 
connections to video 
amplifier and to 
vision IF's. 



Check 

625 video detector 
diodes and compo- 
nents. 

Standard Selection 
switch sections and 
connections to video 
amplifier and to 
vision IF's. 



Is sync poor for both 
vertical and horizontal? 



Is vertical 
sync poor? 



-HNo 



Is horizontal 

sync poor? 



Check 

Sync separator 
valve, its operating 
voltages and 
connections to V.A. 
and to line and field 
scan oscillators. 



Check 

Field sync pulse 
integration components 
and connections. 

Sync pulse limiter. 

Field scan oscillator. 

Vertical Hold control. 



Check 

Line sync pulse differentiating 
components and connections. 

Flywheel sync circuit and 
components. 

Line-scan oscillator and 
Horizontal Hold control. 



m iMKrm cum m.t 



3.124 

Some Typical Fault Symptoms 

The next few pages of this Section describe and illustrate a round dozen fault 
symptoms which could affect the picture displayed on the screen of any TV receiver. 
The most probable places in which to look for the cause of the fault symptoms are 
discussed. 

■ 1 Reduced Picture Height 

The symptoms are that the picture is reduced to a few centimetres in height, though 
its width remains normal The vertical linearity of the picture, despite its reduced 
height, remains normal also and is still controllable by the Height, or Vertical 
Linearity, control. 



REDUCED 
PICTURE 
HEIGHT 




If the height, small as it is, does remain controllable by the Height control, the fault 
is due either to incorrect bias voltage to the cathode of the field output valve (though 
this would admittedly tend to cause some loss of linearity) or, more probably, to an 
open-circuit in the cathode bias decoupling capacitor. This large-value capacitor 
(it has typically a capacitance of 50 to 200 wF) is prone to failure of this sort. When 
it goes open-circuit, there is no decoupling for the cathode bias resistor and a large 
amount of negative feedback is applied to the field output valve. This severely re- 
duces its overall amplification— whence the loss of height— but also brings about 
significant improvement in vertical linearity thanks to the power of negative feedback 
to reduce distortion. 

If the height of the picture is not controllable by the Vertical Linearity control, the 
fault is almost certainly to be looked for in the circuitry of the control itself. 

V4 Line Tearing 

The symptoms of this fault are a series of meaningless lines disrupting several areas 
of the picture. A possible cause is weak signal strength, and it is especially common 
in fringe areas of reception. 

A temporary cure can often be effected by re-setting the Line Hold control; but the 
setting of this control is very critical, and the cure tends to be short-lived. 

Possible seats of the fault itself include damage to the aerial; but more likely ones 
are the Line Hold control itself or components in the sync separator stage, especially 
the capacitors coupling the video amplifier to the sync separator or those coupling the 
line sync pulses to the line oscillator. 



§25] 



3.125 



Line Tearing (continued) 

Line tearing is particularly liable to occur in receivers not employing flywheel 
synchronisation — in which case it is the edges of the picture which take on a ragged 
appearance. 



LINE 
TEARING 




The origin of the fault is generally sharp bursts of noise pulses causing erratic 
triggering of the line scan generator — which they are often apt to do in areas of weak 
signal strength. 

^} Poor Vertical Hold 

In this type of fault, the picture appears to be slipping down from top to bottom of 
the screen rather as if it were a series of pictures mounted on an unwinding roller-blind. 



POOR 

VERTICAL 

HOLD 




It is sometimes possible to lock the picture by careful adjustment of the Vertical 
Hold control; but the setting of this control also is usually critical, and here again the 
cure tends to be short-lived. 

The cause of the fault is often a faulty component preventing the field sync pulses 
from reaching the field scan generator; or it may be trouble in the Vertical Hold 
control itself, or in an internal preset hold control. 



3,126 [§25 

^£ Complete Loss of Synchronisation 

The raster is completely unsynchronised in both horizontal and vertical dimensions. 
The picture, if it can be called one at all, consists of a meaningless pattern of lines and 
shapes drifting aimlessly up or down the screen from top to bottom. 



COMPLETE 
LOSS OF 
SYNC 




Since the synchronisation of both line and field scans is affected, it is clear that the 
fault must lie in a section of the receiver which influences the synchronisation of both 
line and field scan generators. The most likely cause of the trouble is a faulty sync 
separator valve, or damage to a component connected to one of its electrodes. 
Alternatively, it could be a fault in the capacitor coupling the sync separator to the 
video amplifier. 

Another possible cause could be something wrong with the VA itself or with one of 
its associated components. More rare would be misalignment of the vision i.f. stages, 
causing distortion of the sync pulses before they reach the sync separator stage. 

Q Collapse of Field 

The only thing visible on the screen is a narrow band of meaningless lines and 
patterns. The Vertical Hold is ineffective, and it is impossible to obtain a stable 
picture. 



COLLAPSE 
OF FIELD 




Such symptoms are usually caused by a fault in the field output valve or in one of the 
components connected to it, or by damage in the field output transformer. 



§25] 



3.127 



■9 Picture "Wavy" and Laterally Displaced 

With. this type of fault, the vertical (field) stability of the picture is quite good but 
parts of the picture tend to drift off to right or to left of their proper positions. The 
picture develops a wavy pattern, and needs constant adjustment of the Line Hold to 
keep it steady. 



Picture 
WAVY, with 
Lateral 
Displacement 




When the horizontal (line) synchronisation is as poor as this, the fault is usually 
to be found in that part of the receiver circuitry which lies between the sync separator 
and the line scan generator. The sync separator itself is probably all right because 
the vertical synchronisation of the picture is not affected. Suspect components are 
therefore those forming part of the CR differentiating stages in the circuit producing 
the line sync pulses, or any component whose failure could upset the flywheel sync 
circuit. Faulty operation of the line scan generator itself is another possibility. 

mh Vertical Foldover 

The lower region of the picture appears to be lifted and folded back on to the part 
of the picture immediately above it. The bottom of the screen is often left blank. 



FOLDOVER 
at Foot of 
Picture 




A fault of this kind is usually caused by malfunction of the field scan generator, 
or of the field output valve. Likely causes of the trouble are leaking coupling capa- 
citors, incorrect valve bias, anode or screen resistors of the wrong value, too low an 
HT supply, or valves with poor emission from a "poisoned" cathode. 



3.128 



[§25 



Hum Bar on Picture 



The symptoms are a thick dark band lying horizontally across a large area of the 
picture. It is usually accompanied by a low hum from the loudspeaker, deriving from 
the 50 Hz mains supply; but occasionally the bar appears across the picture with no 
hum present. 



HUM BAR 

across 

Picture 




When the mains hum is present in the loudspeaker, the trouble is usually only a 
faulty smoothing capacitor in the HT power supply circuit. Things are more serious 
when the bar is present but the hum is not. The first thing to examine then is the 
picture tube, looking in particular for a short-circuit between the cathode and its 
heater. The remedy could be to connect a separate heater transformer for the picture 
tube, with the aim of isolating the heater circuit from the rest of the receiver. The only 
alternative is to replace the picture tube itself. 

KB Trapezium Distortion 

The symptom is a picture of the correct depth on the left-hand side of the screen 
narrowing gradually at both top and bottom until it is only some 70% of the correct 
depth on the right-hand side. 



Trapezium 
Distortion 




A fault like this is almost always caused by an internal short-circuit within the field 
coil assembly, usually between a single pair of adjacent turns on one field coil. The 
only remedy is to replace the coil assembly — which usually means replacing the line 
coils as well. 



§25] 



3.129 



III Severe Vertical Distortion of Picture 

The comic picture below of a gentleman with a very bad smell under his nose is an 
example of sharp vertical non-linearity caused by a fault in either the field scan 
generator or the field output stage. 



Vertical 

Non-Linearity 




Possible sources of the trouble are: (a) faulty valves in either of these stages; (b) 
incorrect bias voltage on one or more of these valves; (c) an open-circuit in the vertical 
linearity control or in one of the components associated with it; {d) leaky coupling 
capacitors ; (e) too low a voltage on the screen of the field output valve. 



Q 



Grainy Picture 



A grey, spotty picture like the one below, with very poor overall definition, would 
be infuriating if you had had a bet on the race about to start! 



Picture 
GRAINY 




Symptoms such as these are often caused by a noisy valve, or by a fault in the tuner 
or in the vision i.f. stages. For a noisy valve there is no cure save replacement; but 
it is worth looking around for something equally capable of causing the trouble but 
which is easier to cure. This "something" is often poor contact between the pins of 
a valve and its valveholder, or between a pair of switch contacts. 

A thorough cleaning using a rag or clean brush impregnated with a patent solution 
of carbon tetrachloride called "Thawpit" will often remedy the condition. 



3.130 



[§25 



Negative Picture 



A picture like this resembles the negative of a family snapshot plus movement, and 
plus also a number of prominent white lines running horizontally across the picture. 



Picture 
NEGATIVE 




Though it looks at first sight as if a faulty picture tube is the obvious source of the 
trouble, it is in fact those prominent flyback lines which give the clue to a more pro- 
bable cause. For they generally indicate that the fault is caused by abnormal working 
in the video output stage. 

Points to check include the operating voltages of the valves in the output stage, the 
values of the several resistors in the stage, and the continuity of the frequency- 
compensation inductors. The coupling capacitors in the circuit should also be checked 
for an open circuit. 

Test Equipment— The CRO 

Once the fault has been located to a particular section of the receiver, it is usually 
necessary to carry out a number of follow-up measurements of voltage, current and/or 
resistance to pin-point the fault to an actual component or connection. This process 
is considerably assisted if a cathode-ray oscilloscope (CRO), such as that shown, is 
available for inspecting the various waveforms at different points in the receiver. 

The usefulness of a CRO lies in the fact 
that it can often enable a faulty component 
to be quickly located by showing up on its 
screen either the absence of, or the degree 
of distortion present in, the waveform 
which ought to be present at a particular 
point in the equipment circuitry. 

It is not easy, for instance, to verify the 
presence of an open-circuit in a low-value 
capacitor without resorting either to direct 
substitution "on spec" or to the use of a 
rarely-used capacitance-measuring instru- 
ment. But the CRO will quickly show 
that all is not well in that part of the 
circuit by the abnormality of the waveform 
it displays on its screen. 




§25] 3.131 

Test Equipment— The CRO {continued) 

The TV waveforms which can most easily be inspected on a CRO are those of the 
video signal, the line and field sync pulses both before and after separation, and the 
demodulated sound (audio) signal. 

The selection of a suitable CRO for television fault-finding is, as usual, governed by 
the conditions which the instrument will be likely to meet. Since you are going to 
be examining waveforms most of which have very sharp edges (and therefore a high 
frequency content), the frequency response of the internal "Y" amplifier of your CRO 
must obviously be wide enough to reproduce these edges without itself introducing 
distortion in the waveform being inspected. Thus the frequency bandwidth of the 
internal "Y" amplifier of a suitable CRO should be of the order of 3 MHz or better. 
Its sensitivity (the sensitivity of a CRO is expressed in millivolts per centimetre) should 
be 100 mV/cm or better; and it should preferably have a calibrated input-signal 
attenuator. The timebase sweep rates should extend from about 1 us/cm to 5 ms/cm, 
and the triggering facility should be positive and stable. 



The Multimeter 

Often, however, it will be neither possible nor necessary to use a CRO, and fault- 
finding procedure will depend on simple measurements of voltage, current or resis- 
tance. For this purpose a multi-range testmeter (commonly abbreviated to "multi- 
meter") capable of measuring resistance and a wide range of d.c. voltages and currents 
is an essential piece of equipment. 

The range of voltages which a multi- 
meter suitable for use in a TV receiver 
should be able to measure would extend 
from about 0-5 V for measuring such 
things as grid bias and AGC voltages to 
about 700 V for the boost voltage and the 
potential on the first anode. As regards 
current, the ability to measure from half-a- 
milliampere to some 300 mA would be 
adequate. 

Typical resistance values which might be 
encountered range from a few ohms in the 
scanning coils to several megohms in the 
grid bias resistors. For this formidable 
dynamic range, however, a "multi" 
possessing a measurement range extending 
from about I k to 1 M would be a reasonable compromise. 

The internal resistance (and hence sensitivity) of the multimeter should be as high 
as possible so that its presence does not appreciably affect the operation of the circuit 
when voltages developed across high-value resistors such as the anode and screen load 
resistors are being measured. The sensitivity of a testmeter is expressed in ohms per 
volt, and a good-quality multimeter should have a sensitivity of at least 20,000 ft/V. 
Certainly anything less than 10,000 O/V would be of little value. 

Note that it is very seldom necessary to measure a.c. voltage when fault-finding in a 
TV receiver. There is usually only one source of a.c. voltage, the mains supply; and 
the presence or absence of this is easily detected with the aid of an electric light bulb 
and two short pieces of wire. 




3.132 m 

The AVOmeter 

If you are a prosperous sort of chap with a growing repair business, a desirable 
possession is a type of multimeter called the "AVO Model 8", This reliable and 
rugged instrument, with 20,000 Q/'V sensitivity and a large number of ranges, is highly 
regarded in "the trade"; but the more humble operator or repairman will find that 
some of the many 20,000 D/V multimeters advertised in the popular technical maga- 
zines offer good value for an outlay of a few pounds. 

One final word of warning to be borne in mind when you are using a multimeter of 
any kind. Always check that you have it set to the correct range before you make any 
sort of measurement with it. You would be surprised how many "pro's" with long 
years of experience behind them still forget this simple rule— and how many rather 
expensive testmeters are ruined in consequence. 

The Signal Generator 

A piece of equipment of essential value to the professional service engineer is an 
r.f. signal generator of the type pictured below. With such an instrument he can 
improvise a steady signal at the aerial socket of any TV receiver he is testing, setting it 
to the frequency of any particular channel or to any sound or vision carrier frequency 
within that channel. This makes him independent of all fixed-time test transmissions 
and enables him to work on a receiver at any time which suits him or his client. 




4 tyfifcat TV 

SIGNAL 
GENERATOR 



The principal use to which a professional repairman puts a TV signal generator is to 
help him in the re-alignment of the r.f, stages in the tuner. Alternatively, he can reset 
the instrument to the much lower i.f. frequencies and use it to help him align the sound 
or vision i.f. stages. In addition, most signal generators provide means for intro- 
ducing a fixed amount of amplitude modulation into the signal they generate, so that 
an "audio output" appears at the loudspeaker during alignment. This can be used 
either as an aid to correcting alignment or as a means of checking the efficiency of the 
demodulation process and all the subsequent audio circuits. 

The do-it-yourself fault-finder has much less need of a signal generator; for the 
need to re-align an individual TV receiver comes but rarely, and the process of align- 
ment is in any case not the sort of task which should be lightly undertaken — certainly 
not without full alignment instructions from the manufacturer of the receiver. 

As you would expect, the signal generators suitable for TV servicing which possess 
the most comprehensive range are also the most expensive. Such generators can 
produce signals over the entire frequency spectrum from Band 1 to Band 5. Less 
expensive models cover the frequencies in Bands 1 to 3; while cheaper ones still are 
limited to the sound and vision i.f. frequencies only. 



§26: TREHDS IN TV RECEIVER DESIGN 



3.133 



The reasons why this Series on Basic Television has been almost wholly explained 
in terms of the thermionic valve, despite the progressive replacement of the valve in 
many modern receivers by semiconductor devices of various kinds, have been briefly 
mentioned in the Preface. The second of the reasons there given was basically 
instructional convenience and efficiency, and no more will be said of it here; but the 
first reason, which postulated that the valve will continue to find significant use in TV 
receivers for some years to come, calls for a brief outline of some of the technical and 
commercial considerations which affect manufacturing and marketing decisions by 
TV manufacturers in Britain in the latter half of the year 1971. 

Among the most relevant of these considerations are the following: 

1. Though the semiconductor device (commonly, though not always accurately, spoken 
of generically as "the transistor") has many advantages over the valve — princi- 
pally in its small size, low power consumption and greater robustness, but in- 
creasingly also in lower manufacturing cost — it can seldom be used in straight 
substitution for a valve without considerable re-design of associated components. 

2. Re-design of an existing chassis, and tooling-up to mass-produce it thereafter, is an 
expensive business — not lightly to be undertaken save at relatively long intervals 
during a period of swift technological change like the present. 

3. Though the 405-line system in Great Britain will certainly be replaced sometime, 
nobody yet knows when that "sometime" will be. As long as VHF transmissions 
on 405 lines continue, receivers will be needed to pick them up. But it is unrealistic 
to expect such receivers, with their limited life expectation however well they are 
made, to be continually re-designed to keep up with technical advances. 

For these reasons, there are grounds for thinking that the thermionic valve— efficient, 
well tried-and-tested in a wide range of operating conditions, and not yet priced out of 
the market for many TV requirements — will remain in significant use for a considerable 
time after it has become technologically obsolescent. 

It is not a waste of time learning how the many valves of various kinds used in the 
British Dual-Standard Receiver do their job, for they will probably still be there for 
some years to come. 



The Semiconductor in TV Receiver Design 

The first valve-type component to be replaced by a semiconductor equivalent was 
the diode, for the S/C diode is much smaller and more reliable, and can be substituted 
for the valve diode without creating any considerable difficulty in the design of associated 
circuitry. The change took place many years ago, as soon as the semiconductor 
diode became competitive in price. 

Stages of the receiver into which the S/C diode was introduced included those 
handling signal demodulation, AGC and the limitation of noise. In these applications 
diodes of suitable rating were designed which occupied only a fraction of the space 
demanded by their valve-type counterparts and which consumed virtually no power 
(because they require no heater supply). They had thus the further advantage of not 
contributing to the heat-dissipation problem in the receiver. 



3.134 



[§2« 



The Semiconductor in TV Receiver Design (continued) 

Other important stages of the receiver into which the S/C diode was early intro- 
duced were those handling the rectification of the alternating mains current, and 
generation of the HT( + ) supply. In the early receivers, large metal rectifiers of the 
copper-oxide or selenium type such as those illustrated in Basic Electricity, page 3.21, 
had been exclusively used. They were heavy and large — often more than 100 mm in 
length and 50 mm in effective diameter — but reliable and robust under conditions of 
severe usage. 

These rectifiers were later superseded in many models by smaller and lighter valve- 
type diodes, two of which were often connected in parallel to provide the large current 
flows required by receivers containing many valves. These valve-type diodes, of 
course, ran at high temperatures, so contributing to the overall heat generated within 
the cabinet, and they suffered from the additional disadvantage of being much less 
reliable than the metal rectifiers. 

Nowadays, however, this sort of job is almost always done by a single semiconductor 
diode less than half the size of a thimble. It is not only smaller, lighter and cheaper 
than the heavy rectifier and the valve diode, but by reason of its much lower forward 
voltage drop is far more efficient electrically than either of them. 




An old-style 
METAL RECTIFIER 



A 



A modern 

SEMICONDUCTOR 

DIODE 



The Transistor in TV Receiver Design 

The replacement of the principal triode and pentode, etc. operating valves in the TV 
receiver has been a much slower process, and even now is far from complete. Granted 
that some of the latest colour receivers contain no valves at all, many of the latest 
monochrome receivers, even those of single-standard 625-line type, are still no more 
than 50% transistorised. 

The reasons for this are those of commercial logic. Look inside several of the 
many different makes of receivers which are marketed today under various brand 
names, and you will see that they differ principally in external appearance. Inside, 
you will find that no more than three or four different types of chassis are used, all 
designed and assembled by one or other of the small number of large component 
manufacturers who make for others but are not directly concerned with selling a 
finished product. When a well-designed chassis is selling well and when (as is often 
the case) it uses valves produced by the chassis manufacturer himself at a cost often 
significantly lower to him than would be that of a transistor counterpart, the incentive 
to produce a new design based entirely on the use of transistors and other semicon- 
ductor devices is limited. 



§26] 3135 

The Transistor in TV Receiver Design (continued) 

This is especially true of receivers capable of receiving programmes transmitted at 
VHF on the 405-line system. It is known that this system will one day be phased out 
completely, but the decision when this will actually happen is a political one — and not 
by any means an easy one either. There exist in Britain today many tens of thousands 
of TV receivers capable of receiving 405-line programmes only. Their owners are not 
unhappy with the programme variety they are getting for their money at present and 
will thank no one for a decision whose effect (as they may well see it) is to make their 
present receiver useless in the interests of a further advance in technical progress whose 
benefits they do not want at the price. Since a high proportion of these folk are 
likely, in the nature of things, to belong to the less affluent sections of the electorate 
(old-age pensioners and the like), the difficulties of the man who must one day take the 
fateful decision are obvious. 

The only sets currently being manufactured which are capable of receiving pro- 
grammes broadcast at VHF on the 405-line system are Dual-Standard Receivers of the 
type described in this Series. For the reasons given above, it seems probable that this 
Receiver may have a good long life still ahead of it, and that valves will continue to be 
used in its circuitry for as long as it continues to be made. 

Two main difficulties face manufacturers wishing to replace, e.g., an amplifying valve 
with a transistor equivalent. The first is that it is not so simple a job as it was with 
the diode. The operating voltages are completely different, and so are the amplifying 
parameters and the input and output impedances. A good deal of re-design is thus 
necessary in any case — and since a transistor amplifier is so small by comparison with 
the valve it replaces, the second difficulty is to know where to stop. Why not take 
advantage of very small component size to reduce the overall dimensions of the 
cabinet itself? — and at once you are faced with the need for a new chassis, new cabinet 
styling, heavy tooling-up expenses and all the other manufacturing and promotional 
costs of launching a completely new model. 

In the illustration below, a typical TV amplifier valve shown (both actual size) 
alongside a transistor device capable of replacing it entirely demonstrates the extent 
of the opportunity, and of the challenge, presented. 




A typical 
AMPLIFIER PENTODE 



and its 

TRANSISTOR EQUIVALENT 



3.136 [§26 

The Transistor in TV Receiver Design (continued) 

Historically, the first section of the receiver to be transistorised was the UHF 
Tuner stage — principally because it was a new item in the overall circuit which had not 
been needed at all before the coming of UHF transmission. There were therefore no 
reasons of commercial or other logic why it should not be designed around the most 
efficient and up-to-date components from the start. 

Later came the transistorisation of separate, discrete sections of the receiver cir- 
cuitry such as the sound and vision IF stages; but the more complex sections such as 
the line and field output stages, together with the ones which have to handle high 
voltages like the EHT power supply, still largely depend on the use of valves. Indeed, 
it is only quite recently that transistors capable of performing all the functions of the 
valve in a TV receiver have become available at an economic price. 

Looking to the Future 

A significant aspect of recent design philosophy is the concept of modular construc- 
tion, in which entire stages or parts of stages in the TV receiver are manufactured as a 
complete unit which can be replaced by plugging in another unit of the right kind if 
anything should go wrong with the circuitry of the first. 

Plug-in units of this kind can be wired in the normal way, but more often they are 
designed to take advantage of the greater neatness and compactness afforded by 
another development of recent years called printed circuit wiring. In PCW, the 
wiring of the components making up whole sections of the circuitry of a TV receiver is 
formed by etching away all unwanted areas of the thin metal backing of the insulated 
board on which the components are mounted. A completed circuit card, as it is called, 
is thus a maze-like arrangement of flat metal strips mounted side by side — each strip 
no more than a fraction of a millimetre thick and each having a different wiring 
pattern etched into its surface. Complete circuit cards are then plugged into appro- 
priate sockets on the chassis, with the result that the initial assembly, testing and 
alignment are greatly simplified. Later, the all-important servicing becomes merely 
a matter of tracing the fault to a particular card and replacing it with a new one. The 
old card is then sent for repair by an experienced specialist in the workshop. 

More advanced circuit designs still are making use of integrated circuits for some 
sections of the receiver. These are devices, often no larger than a postage stamp, 
which carry in the semiconductor-type material of which they are made whole circuits 
otherwise requiring scores of transistors, diodes, resistors and capacitors. A fre- 
quency-modulated sound IF circuit, for example, complete with ratio detector, can 
be bought today in integrated circuit form contained in a "package" like the one 
shown more than twice actual size in the illustration below. 



A COMPLETE FREQUENCY- 
MODULATED SOUND IF 

circuit in Integrated 
Circuit form 




SMI 



3.137 



Looking to the Future (continued) 

This is undoubtedly the direction in which design is moving; and eventually all TV 
receivers will be of the integrated-circuit type. Before this comes about, however, the 
picture tube of today, with its electron-beam scanning and enormous HT voltage 
requirement, will need to be replaced by a matrix of light-emitting diodes scanned by a 
system of electron switching. TV receivers operating on this advanced technique 
should become commercially available before the 1970's are out. 



INDEX TO PART 3 



Aluminizing 


3.81 


Focusing (of Electron Beam) 


3.69 et seq. 


Automatic Gain Control (AGC) 


3.93 et seq. 


— , Electromagnetic 


3.71 


Auto-Transformer 


3.57 


— , Electrostatic 


3.69 et seq. 


AVOmeter 


3.132 


Frequency Comparison 


3.26 et seq. 


Black Spotter 


3.90 et seq. 


Gated AGC 


3.104 et seq. 


Blocking (of Mean-Level AGC) 


3.102 et seq. 


Ghosts (TV) 


3.84 


Blocking Oscillator 


lAZetseq. 


Grainy Picture 


3.129 


Boost Diode 


3.55 et seq. 






Brightness Control 


3.16 


HT Supply 


3.113 


Cathode Ray Oscilloscope (CRO) 


3.130 et seq. 


Hum Bar, on Picture 


3.128 


Circuit Cards 


3.136 






Clipping Level (of Limiter) 


3.14 


Integrated Circuits 


3.136 


Coincidence Detector 


3.27 et seq. 


Ion Burn 


3.80 et seq. 


Collapse of Picture Field 


3.126 


Ion-Trap Magnet 


3.80 


Colour Coding (of Wiring) 
Contrast Control 


3.112 
3.99 et seq. 


Limiter (of Field Sync Pulse) 


3.14 


Corner Cutting 
Corona Discharge 
Cross-Talk 


3.76 
3.63 
3.46 


Line Hold 

— Output Stage 

— Pairing 


3.19 

3.17, 3.48 et seq. 

3.12 






— Scan Generator 


3.17, 3.18 et seq. 


Deflection of Scanning Beam 


3.72 et seq. 


— Scanning Coils 


3.16, 3.49 et seq. 


Delayed AGC 


3.97 


— Sync Pulse 


3.2 


Direct- View Picture Tube 


3.82 


Separation 


3.6 et seq. 


Double-Delayed AGC 


3.98 


— Tearing 


3.24, 3.124 et seq. 






Linearity Correction 


3.79 


Efficiency Diode 


3.53 


Loss of Sync 


3.126 


EHT Smoothing 


3.82 


LT Supply 


3.114 et seq. 


— Voltage Generation 


3.59 et seq. 






Electron Gun Assembly 


3.67 et seq. 


Magnetostrictive Effect 


3.58 


Equalizing Pulses 


3.13 


Mean-Level AGC 


3.96 et seq. 






Modular Construction 


3.136 


Fault-Finding 


3.117 et seq. 


Multimeter 


3.131 


Fau